Auotmatic Switching for Nationwide Telephone Service
A. B. CLARK AND H. S. OSBORNE 823

Fundamental Plans for Toll Telephone Plant J. J. PILLIOD 832
Nationwide Numbering Plan W. H. NUNN 85]
Automatic Toll Switching Systems F. F. SHIPLEY 860

Mathematical Theory of Laminated Transmission Lines—Part I
SAMUEL P. MORGAN, JR. 883

Impedance Bridges for the Megacycle Range H. T. WILHELM 999
Abstracts of Bell System Papers Not Published in this Journal 1013

THE BELL SYSTEM TECHNICAL JOURNAL is published six times
a year by the American Telephone and Telegraph Company, 195 Broadway,
New York 7, N. Y. Cleo F. Craig, President; Carroll O. Bickelhaupt, Secretary;
Donald R. Belcher, Treasurer. Subscriptions are accepted at $3.00 per year.
Single copies are 75 cents each. The foreign postage is 65 cents per year or 11

cess of the No. 4 installation at Philadelphia, led to studies of the
service and operating results which might be expected from a nationwide
extension of automatic switching. The conclusion was reached that this
would be a desirable objective of the Bell System companies and would
result in a very substantial further improvement in the speed and ac-
curacy of handling of long distance messages. Accordingly, during the
next few years, a national plan was prepared and was adopted by the
telephone companies.

The features of this nationwide plan and the present status of its
application form the subject of the three technical papers which accom-
pany this introductory paper.” “° The basic requirements to be met*
in the development of this plan included the following:

1. It should be suitable for the nationwide extension of automatic
switching both by originating toll operators and by the customers direct.

When this work was commenced it was clear that a program leading
toward general nationwide operator dialing was desirable. Subsequent
developments have confirmed the wisdom of making the basic plan
consistent with general nationwide customer dialing as well since it now
appears that a very wide extension of this form of service will become
desirable.

2. The plan must provide for satisfactory overall service between
any two telephones in this country and Canada.

Under manual operation satisfactory overall service was provided
for by the general toll switching plan in use since about 1930. This plan
is modified to recognize the far greater speed and accuracy of automatic.
switching compared with manual switching. This involves also modifi-
cations of transmission design standards so that the overall connections
will continue to be satisfactory.

3. The system must be designed for instantaneous service, so that
delays due to lack of circuits or equipment would be very infrequent.
This is necessary, both from the standpoints of service and the avoidance
of tieups, particularly of the automatic switching machinery.

A trunking system must therefore be devised which will most economi-
cally meet this requirement, considering overall costs of lines, switch-
ing equipment and operation.

4. Machines must be designed for use at strategic points in the net-
work, called “control switching points’, to perform automatically the
various tasks required to make the overall plan operative and economical.

type of local central office equipment, called No. 5 crossbar, which was
designed with this in view.’ For older types of equipment, the job is
more difficult.

2. The switching equipment must be provided with automatic means
for recording all of the information necessary for charging the call. In
the case of operator dialing this is now done manually by the operator.

Great advances have been made in recent years in the development of
automatic message recording equipment. In 1944 there was placed in
service in California the first installation in this country of automatic
ticketing equipment.” This equipment is associated with step-by-step
local switching equipment and automatically prints for each call a
ticket similar to that prepared by the operator with manual operation.
In 1948 there was installed in Media, near Philadelphia, a greatly im-
proved type of message recording equipment in which the information
appears in the form of punched holes in a tape.’ This equipment is
much more economical than the earher system and also lends itself to
the automatic preparation of toll statements or bills.

The present forms of equipment have been designed to be associated
with local central offices. A careful study has been made of their field
of application and cf the basic plan necessary to provide for a general
nationwide extension of customer dialing. This indicates that there will
be a large field for automatic message accounting equipment associated
with the toll network and arranged to receive orders for toll messages
from a number of local dial offices. This centralized AMA equipment,
as it is called, is under development and an initial installation will be
made next year in Washington, D. C. In this installation the range of
customer dialing will be limited and certain service features will be
lacking, which it is planned to add later.

Recognizing that the best way to develop these questions is a trial,
arrangements were made to open such a trial last fall at Englewood,
N. J. This office is equipped with a No. 5 crossbar system so that arrange-
ments for such a trial could readily be made there. The Englewood
customers are able to dial directly any of about eleven million telephones
in ten metropolitan areas scattered throughout the country, including

All of these plans depend upon the successful development of striking
innovations in toll switching and automatic message accounting equip-
ments. The plans in turn react upon the features to be incorporated
in such equipments and upon the schedule of their development. Mr.
Shipley’s paper, pages 860 to 882, tells about the more important fea-
tures of these equipments and the problems which are involved in their
development.

Experience with operator toll dialing shows clearly that it provides a
marked improvement in toll service. This improvement will increase as
progress is made toward the full application of the nationwide automatic
switching plan. j

The development of long distance dialing by customers is at an early
stage. The results of recent trials, however, indicate that nationwide
customer dialing has service advantages and will generally be received
with enthusiasm by telephone users. It is anticipated, therefore, that
customer dialing will rapidly expand both on a regional and on a nation-
wide basis.

Switching plans providing for the systematic routing of toll telephone
traffic have been employed by the communication industry for many
years. These plans have contributed directly to the high quality of long
distance telephone service enjoyed by the public in the United States
and Canada. This generally excellent service is the result of the coopera-
tive work of many organizations including the Bell Operating Companies,
many independent connecting Companies and others in the United
States as well as in adjoining countries. The techniques employed today
reflect a great amount of research and engineering and improvements in
manufacturing skill and in construction, maintenance and operating
methods developed over a period of many years.

Throughout the United States and Canada there are approximately
20,000 different places — cities, towns, and villages — that serve as toll

In order to illustrate the problem a specific example may be useful.
Fig. 1 is a map of Wisconsin and Minnesota on which nearly 1200 circles
indicate points at which exchange facilities may’ be connected to the
toll network. The extent of the coverage in this area is typical of that
found throughout the country.

More than 1,000 smaller circles on the map represent “tributaries” —
that is, towns where little or no toll operating is done. Toll connections
to and from these points are completed at the toll centers which in gen-
eral do the toll operating required.

In the United States and Canada as a whole, there are approximately
2,600 toll centers. The remainder of the toll connecting points—about
17,500—are tributaries.

Fig. 2 gives an idea of the variety and complexity of the network of
circuit groups required to interconnect the toll centers in one area. Here
each line represents a group of circuits, known as “intertoll trunks,”
between two toll centers. Each group may contain anywhere from one
to several dozen trunks. The location of the lines on the map is unrelated
to the geographical routing of the trunks, and only a part of the circuit
groups are shown. To get a complete picture one should visualize that a
cluster of relatively short circuit groups radiates from each toll center
to its tributaries, of which there may be up to 15 or more.

Physically, the plant consists of a network of open wire lines, cables
and radio systems. On these, voice frequency or carrier operation is
employed in each section as required to provide the necessary intertoll
trunks. The routes of the lines in Minnesota and Wisconsin are shown
by Fig. 3. In this area there are no radio routes carrying telephone cir-
cuits, but a radio system between Chicago and Minneapolis is in the
planning stage.

Areas like Wisconsin and Minnesota must, of course, be connected to-
gether, and Fig. 4 shows the major Bell System toll routes that accom-
plish this. On a map of this kind it is not possible to include anything
like the detail shown in Fig. 3. One must visualize, therefore, that each
state contains a network of routes generally comparable to those shown
for Wisconsin and Minnesota.

This then represents the interconnection problem to be met by an
orderly switching plan that will provide efficient, reliable and fast toll
telephone service between any two points.

extensive use of carrier than would have been practicable with a slower
rate of growth.

- Consideration of these factors which offer an opportunity to improve
service has led to the gradual reorientation of the fundamental plans
for the intertoll trunk plant which is now under way.

in turn one segment of the Chicago “region” which serves a somewhat
larger area than shown by Fig. 5.

Under this arrangement, toll calls between two tributaries in the Hib-
bing toll center area can be completed by switching at the toll center.
In a similar manner, any two points within the Duluth tandem outlet
area can be served by switching at Duluth. The same treatment also
applies for connections between any two points in the same sectional
center area or in the same regional center area. For example, a connection
from Hibbing to any point within the Chicago region (which involves
more than six states as shown in Fig. 7) requires no more intertoll links
than Hibbing to Duluth, Duluth to Minneapolis and Minneapolis to
Chicago, and a corresponding number of links on through to another
sectional center, and primary or tandem outlet to the toll center desti-
nation. Circuits between the toll center and tributaries are not referred

Fig. 5—Intertoll trunks between Davenport, Iowa and Hibbing, Minnesota,
showing alternate routing possibilities.

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tically all overflows from the high usage groups during the heavy traffic
periods. The “high usage’’ and “final” groups which could be used for
routing calls between Hibbing, Minnesota and Davenport, Iowa are
shown by Fig. 5.

THe expanded general toll switching plan for nationwide dialing con-
templates a degree of alternate routing far in excess of that used with
the former switching plan designed for manual operation. This change
along with the reduction in toll centers will have a marked effect on the
normal flow of many traffic items through the intertoll network. As a
result the arrangement of the present intertoll trunks will be significantly
modified both in number, routing and terminating points. It is necessary
to take these facts into account in engineering toll plant additions so
that they will lead toward an advantageous layout for future nationwide
dialing as well as meet the needs of the more immediate future. Fortun-
ately, the effect is in the direction of greater concentration of circuits
in main routes so that with the new cable and radio facilities available,
over-all economy and better service should result.

Fig. 8—Terminal equipment of type-N1 cable carrier system. Provides twelve
message channels with self contained signaling equipment over two pairs of cable
conductors in same sheath.

Fic. 10—Microwave radio relay tower at Cotoctin Mountain, Maryland, on a
New York-Washington radio route. There are 300 message circuits in service with
more planned.

on the type of carrier system (Fig. 8). Coaxial cable transmission systems
currently provide up to 600 telephone channels per pair of coaxials
(Fig. 9). A new coaxial system, under development, is expected to pro-
duce about 1,800 telephone channels per pair of coaxials.

Most of the applications of radio for toll telephone service now contem-
plated, involve the use of point-to-point microwave systems. By employ-

Fig. 11—Growth in Bell System intertoll trunk mileage showing trend toward
more extensive use of carrier type facilities.

vantages. They are inherently of the ‘four-wire” type which minimizes
the number of possible singing and echo paths on a circuit. Also, the
speeds of propagation over carrier systems are generally higher than over
voice frequency systems thereby further minimizing the echo problem.
These features are of great advantage in reducing limitations on circuit
design and layouts of the general toll switching plan.

In addition to the ability to carry messages, intertoll trunks must be
provided with suitable signaling facilities.” ’ These must provide a means
of: first, attracting the attention of the distant point, either an operator
or automatic equipment, to the fact that a connection is to be established ;
and second, in the case of dial operation, transmitting coded information
in the form of pulses for establishing the connection; and third, trans-
mitting a general class of supervisory signals including connect and
disconnect signals, on and off switch hook signals, recall signals and

With dial operation, the number of trunks in tandem in a given toll
connection may vary on successive calls. To avoid undesirable trans-
mission contrasts and other adverse effects, it is important that every
trunk be designed to operate as closely as possible to the theoretically
correct transmission loss. The problem is complicated by the fact that
the extent to which the echo, noise and crosstalk will limit the perform-
ance of an individual link is not directly proportional to the length of the
circuit. In fact, the minimum loss at which a particular circuit used
singly or in various built-up combinations can theoretically be operated
depends on the number, length and characteristics of the other circuits
connected in tandem with it. Arrangements for precisely adjusting the
loss in the individual trunks for each call would be complicated. Adequate
performance can be achieved however by compromise methods which
provide for automatic adjustments in the loss of each trunk in accordance
with the following:

1. When a trunk is switched to other intertoll trunks at both ends it
is operated at the minimum loss practicable. This loss is known as “‘via
net loss.’”? (VNL)

2. When the trunk is switched to another intertoll trunk at one end
only, the loss is increased two db.

end a further loss of two db is added. This loss which is four db greater
than the via net loss is known as “terminal net loss.”? (TNL)

The data and methods used in the derivation of the via net loss are
rather complex and not within the scope of this paper.

The definite routing patterns established for the toll machine switching
operation impose more severe transmission conditions on certain classes

of circuits than on others. For example, a trunk in a “final” group be-
~ tween a TC and a PO can become involved in an eight-link connection,
whereas a trunk in a “high usage” group, say, between a PO and another
PO will not be involved in more than a three-link connection.

This creates a need and provides an opportunity for allocation of the
available facilities among the various trunk groups in a way that will
provide the best overall service. For example, to the extent practicable
it is desirable to assign carrier grade facilities to trunks in “‘final’’ groups
that may be involved in connections with the maximum number of links.
Facilities with less favorable transmission characteristics may then be
reserved for trunks in groups that are used for connections involving
fewer links.

1. H. 8. Osborne, ‘‘The General Switching Plan for Telephone Toll Service,”’
A.I.E.E. Transactions, 49, pp. 1549-1557, 1930.

. C. A. Dahlbom, A. W. Horton, Jr. and D. L. Moody, “Application of Multi-
frequency Pulsing in Switching,” A.J.Z.E. Transactions, 68, pp. 392-396,
1949.

7. N. A. Newell and A. Weaver, ‘‘Single-frequency Signaling System for Super-
vision and Dialing over Long Distance Telephone Trunks,” A.J.E.E. Trans-
actions, 70, (7 pages), 1951.

8. Articles prepared by American Telephone and Telegraph Company for infor-
mation of Dial Interexchange Committee of the United States Independent
Telephone Association. Published in Telephony on dates indicated.

In telephone language a numbering, plan gives each telephone in a city,
a town, or a geographical area an identity or designation different from
that given any other telephone in the same area. There is a wide variation
in the types of numbering arrangements in use today in the Bell System,
and this paper gives the reasons for this diversity, and examples of the
various numbering plans now in use. With the introduction of modern toll
switching facilities and the extension of toll dialing to nationwide scope,
it was realized that an improvement in the method of dialing toll calls to
distant cities was essential in order to realize the maximum speed and
accuracy inherent in toll dialing. A nationwide numbering plan covering the
United States and Canada has been designed. Each of the more than 20,000
central offices in the two countries are to be given a distinctive designation
which identifies that particular office. This designation is to consist of a
regional or area code and a central office code The new switching equipment
for the key points in the toll network is being designed so that any toll opera-
tor, wherever located, will use the same designation or code for reaching a
given office. The combination involved in laying out these areas and the
composition of the area codes are presented. A total of 152 codes are available
of which approximately 90 are assigned to the present numbering plan areas.
Ultimately each central office will be given a type of number consisting
of an office name and five numerical digits, such as LOcust 4-5678, in
which the first two letters of the office name become the two letters of the
central office code. The entire program will take a considerable number of
years to realize, but 1s one which must be accomplished in order to achieve
the best results in operator toll dialing and the ultimate goal of nationwide
customer toll dialing.

In telephone language a numbering plan is exactly what the name im-
plies, a plan or system of giving each telephone in a city, a town or any
geographical area an identity or designation which is different from that
given every other telephone in this same area. This designation is the

In many places local service areas were changed so that customers
could call into contiguous exchanges at local rates. To enable customers
to dial into these neary-by places the original numbering plans required
expansion to include this increased number of offices. In addition, with
the advance in the telephone art many cities introduced equipment for
automatic charging on multi-unit and short haul toll calls so that cus-
tomers could dial such calls directly instead of placing them with an
operator for completion. In order to enable customers to dial these calls,
it was necessary to expand the original city numbering plans to encom-
pass wider and wider geographical areas.

In expanding the various types of numbering plans to serve a larger
number of central offices than were originally anticipated, various ex-
pedients were resorted to. In the largest cities having three-letter office
codes a numeral was substituted for the third letter thus very materially
increasing the code capacity from about 325 to about 500 and making it
possible to form a number of codes using the same office name. The name
CANal for example, instead of serving but one office may serve a number
of offices, CAnal 2, CAnal 3, CAnal 4, etc. In the medium size cities
having two-letter codes, expansion meant adding a digit to the code to
all or in some cases to only a part of the offices in the city.

The five-digit places were usually expanded by adding a digit to
some of the numbers so that some of the telephones had five digits and
others six digits in their numbers.

As a result of choosing originally a numbering plan which at the time
seemed adequate and most suitable for the city involved and in many
cases being forced to expand to meet changing needs, we now have in the
Bell System a considerable variety of different numbering plans. These
are given in Table I. The numbering plans given are all adequate to
serve the present local dialing needs for the cities in which they appear.

Having reviewed the numbering plan situation as it exists today in
the various cities and towns, let us turn to the problem of handling toll
calls. Under ringdown operation there is an operator at the outward toll
center where the call originates and another operator at the terminating
or inward toll center. On built-up toll connections there are additional
operators at each intermediate toll switching point. The inward toll
operators, who are familiar with the numbering plans in the offices served
by their particular toll center, can be relied upon to connect to the de-
sired station even though there is uncertainty on the part of the calling
customer or the outward toll operator regarding the precise pronuncia-
tion or spelling of the name of the called office or the particular form of
numbering system used at the called city.

Place Directory Listing Customer Dials Ordinarily Referred to as
Philadelphia, Pa. LOcust 4-5678 LO 4-5678 Two-five
Los Angeles, Cal. PArkway 2345 and | PA 2345 and | Combined two-four
REpublic 2-3456 RE 2-3456 and two-five
Indianapolis, Ind. MArket 6789 MA 6789 Two-four
El Paso, Texas PRospect 2-3456 | PR 2-3456 and | Combined two-five
and 5-5678 5-5678 and five digit
San Diego, Cal. Franklin 9-2345 F 9-2345 One letter, four and
Franklin 6789 F 6789 five digit
Des Moines, Iowa 4-1234 and 62-2345 | 4-1234 and 62- | Combined five and
2345 six digit
Binghamton, N. Y. | 2-5678 2-5678 Five digit
Manchester, Conn. | 5678 and 2-2345 5678 and 2-2345| Combined four and
five digit
Winchester, Va. 3456 3456 Four digit
Ayer, Mass. 629 and 2345 629 and 2345 Combined three and
four digit

but the Z is never used in a central office code), hence any office code
will always avoid a 1 or a 0 in the first two places. The digits 1 and 0
can therefore be used in area codes to distinguish these from office codes.
It is not practical to use them as initial digits of area codes since custo-
mers dial 0 to reach operators and the local dial equipment is arranged to
ignore an initial 1 for technical reasons. A 1 or 0 in the second place,
however, can be employed in an area code without conflicting with any
central office codes or interfering with any existing practices. Accord-
ingly the area codes will consist of three digits with either a 1 or a 0 as
the middle digit, 516, 201, etc. A few codes of this type are now in use,
leaving a practical total of 152 of these area codes available as compared
to approximately 90 assigned to our present numbering plan areas. This
will provide a comfortable spare for additional future numbering plan
areas or possibly for reaching overseas points which may later be in-
corporated into the toll dialing network.

As shown in Fig. 1, states and provinces such as Montana or Alberta
which are contained in a single numbering plan area will have area
codes with a 0 as the middle digit to distinguish them from areas in
divided states such as Texas where the middle digit will be a 1. This is
to enable toll operators to differentiate between the two classifications
of areas. On calls to single area states the operators will always know that
every call to the state in question uses the one area code, whereas on
calls to subdivided states additional information will be required to de-
termine which of the several area codes should be employed to reach the
particular destination. It is proposed to show on the operator position
bulletin the codes of all single area states and the codes of all frequently
called cities in multi-area states. The area codes of the less frequently
called places in the multi-area will be obtained from a routing operator.

Within each numbering plan area each of the 590 or fewer offices
are to be given a three-digit office code which will be different from that
of any other office code in that same area. Ultimately each central office
will be given a 2-5 type of number consisting of an office name and five
numerical digits, such as LOcust 4-5678, illustrated for Philadelphia.
In the larger cities customers will dial seven digits, LO 4—5678, on local
calls to numbers in the same exchange. In many of the smaller places
the customers on local calls will dial only the numerical digits, the office
name being employed for toll dialing purposes only.

Considering the thousands of central offices which now have numbers
other than the 2-5 type and the fact that to change existing numbering
systems is a difficult and often costly procedure, it will be a number of
years before this ultimate objective is realized. As a practical measure,

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digit local dialing and for offices in the larger places served by certain
types of dial equipment, as they are arranged today, it will be necessary
to prefix the dialing of toll calls by a transfer or directing code to permit
the customer getting from the local office into the toll network.

Independent of the advantages of a universal 2-5 numbering plan
for nationwide operator and customer toll dialing, the Bell System has
made considerable progress in this direction over the past several years.
New York and Northern New Jersey adopted 2-5 numbering in 1930
in order to take advantage of the flexibility of office code assignments
and the large code capacity which this type of local numbering provides.
Since World War II many cities and their environs such as Chicago,
Boston, Philadelphia, San Francisco, Oakland, Pittsburgh, Milwaukee,
Providence and a number of smaller cities have followed suit. Presently
about 12 million telephones are in areas which have 2—5 numbering
exclusively in addition to perhaps two million telephones with 2-5
numbers in mixed 2-4 and 2-5 areas. Another five million telephones
are already planned for conversion to 2-5 numbers within the next
several years.

The entire program will take many years to realize but it is one which
must be accomplished in order to achieve the best results in operator
toll dialing and make it possible for a customer at any telephone in the
United States and Canada to reach a telephone anywhere in the two
countries by dialing without the assistance of an operator.

switching points is based on the Toll Crossbar System? now in service
and has been constructed by the addition of the necessary CSP features
to the basic structure of that system.

The necessity of providing this feature in CSP switching systems
arises from the nature of the numbering and switching plans. The num-
bering plan* is constructed with the objective of using a minimum
number of digits to give each telephone user in the country a distinctive
number.

Numbers delivered to the CSP equipment are in the form ABX-
XXXX if the called place is in the same numbering area as the CSP.
AB represents the first two letters of any office name and X represents
any numeral. If the called place is in another numbering area this set of
digits will be preceded. by XOX or X1X. XOX or X1X is the area code,
ABX the local office code, and these are the digits used for routing
purposes. Regardless of the number of switches required to complete the
call, these two sets of code digits are all that will be supplied. They are
universal codes in that they identify specific destinations — any place

digits forward by MF and leaves the connection to accept another call.
This information is relayed from the decoder to the sender by way of
the marker. The work time of the decoder has been in the order of a half
second.

The marker determines the identity of the frames on which the in-
coming and outgoing circuits are located, finds an idle path between the
two circuits and sets up the connection. After checking the path through
the switches to be sure that there are no troubles it notifies the sender
that its task has been completed and then leaves the connection. Its
work time has also been in the order of a half second.

In the meantime other digits have been coming in to the sender but
it does not wait for all of them to arrive before advancing the call. When
the marker selected the circuit to New York a signal was immediately
sent forward to summon a sender in the New York switching system.
The process of attaching the sender in New York was carried on con-
currently with the establishment of the connection through the switches
in Atlanta.

When the New York sender is attached a signal is sent to the Atlanta
sender to advise it that pulsing may proceed. It immediately sends the
area code 207 to New York by MF pulsing and follows it with the
remaining digits of the called number, AC4—2345, as they are received
from the operator, ending with a start pulse, and then leaves the connec-
tion. All common control equipment in Atlanta is now free.

In New York, as soon as the Maine area code is received it is submitted
to the decoder. Upon examination of the code the decoder finds that it is
insufficient for routing purposes. New York has a direct circuit group
to Portland over which traffic to some offices in Maine is routed, but
other offices are reached through Bangor by way of Boston. In order to
determine which route to take the decoder must know what office is
desired. It, therefore, gives the sender a signal saying ‘“‘come again when
you have six digits’? and leaves the connection. When the sixth digit
arrives the sender again calls for a decoder and gives it the complete
code 207-AC4.

The decoder again translates the area code, which now directs it to the
foreign area translator which serves the Maine area, and submits the
complete code to that translator. From the ensuing translation it learns
that the route.is by way of Boston and that all digits should be sent
forward by MF. It then calls for a marker and releases the foreign area
translator.

both codes are again translated since Boston also has a choice of routes
to Maine, and the route to Bangor is selected. The translating equipment
in Boston knows that Bangor is in the Maine area and that the area
code will, therefore, not be needed. However, since Bangor is a TO
having no senders, the Boston sender must pulse forward all of the
digits needed to complete the call through switches in Bangor, Houlton
and Monticello. It is assumed that Houlton is arranged to route the call
to Monticello on receipt of the digits AC4. Numerical digits 2345 will
route the call through the Monticello switches to the called customer’s
line. These digits are all registered in the Boston sender but the digits
required to switch the call through Bangor are not and must be supplied.
An arbitrary set of digits beginning with ‘‘1”’ can be used for this purpose
since no office code begins with “1” and there will, therefore, be no
conflict.

In setting up this call all of the characteristic CSP features were em-
ployed, automatic alternate routing in Atlanta, six-digit translation in
New York and Boston, digit storing and variable spilling at all CSP’s
with substitution of arbitrary digits for the area code at Boston.

* All talking connections through the CSP system are made on a four-
wire basis, that is, separate pairs of conductors are provided for trans-
mission in the two directions. This is done in order to simplify the
problem of maintaining satisfactory balance so that the loss introduced
by extra links in a connection can be held to a minimum value. The
importance of this feature is emphasized by the fact that the switching
plan permits as many as eight intertoll trunks to be connected in tandem
for the completion of a call.

The advantages of four-wire switching were fully explained in the
paper’ on the toll crossbar system now in service.

Two separate groups of incoming senders are provided, one to receive
DP and the other MF pulsing. Whether the system is installed in a
step-by-step or a panel-crossbar area both groups of senders will always
be needed. MF will be received from senders in other CSP’s and from
switchboard positions. DP will be received from switchboard positions

pattern of the CSP network for the country as seen from the CSP
concerned and are interconnected in a pattern of progression correspond-
ing to the fixed order of alternate route selection. Group busy leads from
the toll line groups are connected to the contacts of the relays in such a
manner that if a group is busy the relay corresponding to the next choice
route in the chain will be operated. In this way the lowest choice route
having an idle circuit will be speedily selected without testing individual
trunks of separate groups. The decoder learns from the translator which
relay in the array to operate first and the choice of the best route avail-
able follows automatically. The principle will be readily understood by
reference to the simplified sketch in Fig. 3. Contacts not shown on the
rclays cause the translator to select the route corresponding to the last
relay operated in the chain.

The magnitude of the translating job for nationwide dialing led to the
decision to develop a new translator operating on a principle radically
different from that employed in other crossbar systems. In previous
systems translation is done by relays. The code digits — never more than
three — operate a group of relays which cause a single terminal corre-
sponding to the code to be selected. A cross-connection is made between

HIGH-USAGE ROUTES FINAL ROUTES
OISTANT DISTANT DISTANT HOME ALL
PRIMARY SECTIONAL REGIONAL NATIONAL REGIONAL TRUNKS
OUTLETS CENTERS CENTERS CENTER CENTER BUSY

1-INDICATION FROM TRANSLATION OF THE FIRST ALTERNATE ROUTE
2-INDICATION THAT ALL TRUNKS IN THE GROUP ARE BUSY

the code point and a route relay associated with the trunk group to be
selected. The route relay has a number of contacts:which are cross-
connected to supply the information required for proper routing of the
call. When changes in routing or equipment location of trunks within
the office are made it is necessary to change cross-connections.

With the nationwide dialing plan in operation routing changes or
opening of new offices in one part of the country will necessitate trans-
lator changes in many offices, some of them far removed from the scene
of the event that forces them to be made. The changes in any one CSP
will, therefore, be frequent and to make them by running cross-connec-
tions would be cumbersome and expensive. The new translator uses
punched cards instead of relays, making it possible to effect changes by
the simple process of removing old cards and inserting new ones in a
machine. This can be done in a very short time and not only saves labor
but requires less out-of-service time for the equipment. Fig. 4 is a photo-
graph of the machine.

A metal card about 5 by 102 inches is provided for each area code
and also one for each office code that must be translated in a particu-

TRANSLATOR BOX NUMBER
LGe SAR ORE oA ODIkG. "RIS Gein Ublicn.
INO! HB BTO BT1 BUO BU! BU2 Bu4 BU7

ROUTING INSTRUCTIONS CONT & DIGIT CONTROL
RIO RI1 RI2 RI4 RI7° “COCO CcDdC1 CDC2 CDC4

$
i SOURCE OF RECORD” Sao
OM C OT MT Tv CT TRI “re TRI “Tre Q ! 2 6 7 8 9 10 uZ are mer <4 ! 16 7
TSE OF RECORD MARRERSTT OR COE :
s? FIF MFT CEROSTIMSTIATRETST SOT __oGt] 9 1 2 93 4 1 8 9/0 1 29. ari 6 or)
SOR, FA TENS SOR FR UNITS
3.4 5 6 7 8 910 2.3.4 5 6 ‘ 5 0. ii -
Soaucrten bustets See SST Ca
Wms oe see eile ee a he
DECODER Pur COOL
Miao 1 2 4 a7 90.) 2 4 07 coy 2 4 C? 00 1 2 4 9 £0 1 2 4 FO) 2 477
CODE BARS
AO 1? 4 a7 60 1 24 7 co 1 204 ¢7 poy 2 4 om oy 2 4 ee OP a
OCCODER InpuT

S21 3p 6D 6DA vO _NVO NRO CKICFM PF _TSA TSB TSC SBD} LI U2 U3 1a
DECODER ROUTE ADVANCE
RA RAI RAZ RA GSO GS! _GS2 653. G54 GS5 GO _GI_G2_G3 GB RLS MB_RO_AOIT ROTC
Tar

sie
TRANSMITTEO-MMRKER TO SENDER
R31 a _DC_MF_SxD LPD XOD.XSG DLC SXR20C___ODG 4DG 5DG_NSK SK3 SK6

59
DECODER CAOSS Ix} GECODER Time-OUT MARKER TIME -OUT
CAREC IK CRK ODT TRA CF TRE ORL ALT WT_FT) MO reol "Tie Twi TH2 Tus 38s SOFFICE
MaanER CROSSIN=) OATE
x15 IPs 38 St MS OK OT JP WS OLS JS _SM_SMISMO TL RCK TKS TR TRL STRMRL TIF TOF

2: ie G.  Khinhend A. J. Busch and F. F. Shipley, ‘‘Crossbar Toll Switching
System,” A.J.H.H. Transactions, 68, June Section, pp. 302-309, 1944.

3. C. A. Dahlbom, A. W. Horton, Jr. and D. L. Moody, “Application of Multi-
frequency Pulsing in Switching,” AI.E.E. Transactions, 68, June Section,
pp. 505-510, June 1949.

F. J. Scudder and J. N. Reynolds, ‘‘Crossbar Dial Telephone Switching Sys-
tem,” A.I.E.E. Transactions, 58, May Section, pp. 179-192, 1939.

N. A. Newell and A. Weaver, ‘‘Single Frequency Signaling for Telephone
Trunks,’’ Presented at Winter General Meeting of A.I.E.l., Jan. 31, 1951.

F. A. Korn and J. G. Ferguson, ‘‘The Number 5 Crossbar Dial Telephone
paverne System,” A.J.H.E. Tranasctions, 69, First Section, pp. 233-254,

. J. Meszar, ‘‘Fundamentals of the Automatic Telephone Message Accounting
System, » Presented at the Winter General Meeting of A.I.E.E., Jan. 31,
1951.

A mathematical analysis is given of the low-loss, broad-band, laminated
transmission lines proposed by A. M. Clogston, including both idealized
parallel-plane lines and coaxial cables. Part I deals with ‘“Clogston 1”
lines, which have laminated conductors with a dielectric, chosen to provide
the proper phase velocity for waves on the line, filling the space between the
conductors. Part II will treat lines having an arbitrary fraction of their
total volume filled with laminations and the rest with dielectric, and will be
concerned in particular with “Clogston 2” lines, in which the entire propaga-
tion space is occupied by laminated material.

The electromagnetic problem is first formulated in general terms, and then
specialized to yield detailed results. The major theoretical questions treated
include the determination of the propagation constants and the fields of the
principal mode and the higher modes in laminated transmission lines, the
choice of optimum proportions for these lines, the calculation of the fre-
quency dependence of attenuation due to the finite thickness of the laminae,
the increase in loss caused by improper phase velocity (dielectric mismatch)
in Clogston 1 lines and by nonuniformity of the laminated material in
Clogston 2 lines, and the effects of dielectric and magnetic dissipation.

Ty, VODROMUCHION sin fire ot enact Gxwh dkeaad enced pam band chek daypebek 884
II. Wave Propagation Between Plane and Cylindrical Impedance Sheets. 887
III. Surface Impedance of a Laminated Boundary..................... 896

IV. Principal Mode in Clogston 1 Lines with Infinitesimally Thin Laminae 908
V. Effect of Finite Lamina Thickness. Frequency Dependence of Attenu-

VI. Effect of Dielectric Mismatch. .......... 0.0... c cece teens 931
VII. Dielectric and Magnetic Losses in Clogston 1 Lines................ 940
Appendix I: Bessel’ Function Expansions...................000005 944
Wabble-of Symi bls 22.55: c sees oes asa yt de eeaneasie antenatal lathe yak at ae kedes 946

A recent theoretical paper! by A. M. Clogston presents the very
interesting discovery that under certain conditions skin effect losses in
the conductors of a transmission line at elevated frequencies can be
much reduced by laminating the conducting surfaces, parallel to the
direction of current flow, with alternate thin layers of conducting and
insulating material. The requirements are that the thickness of each
conducting layer must be considerably smaller than the skin depth in
the conductor, and the phase velocity of waves on the transmission line
must be held very close to a certain critical value, which depends on the
relative thicknesses and the electrical properties of the conducting and
insulating layers: Under these conditions the ‘‘effective skin depth” of
the laminated surface is greatly increased; in other words, the eddy cur-
rents induced by a high-frequency alternating field will penetrate much
farther into such a laminated structure than into a solid conductor, with
consequent marked reduction of ohmic losses in the metal. The metal
losses can also be made to vary much less with frequency, over a fixed
band, than the ordinary skin effect losses, which are known to be very
nearly proportional to the square root of frequency.

Clogston goes on to show that a laminated material composed of
alternate thin conducting and insulating layers may itself be regarded
as a transmission medium. Tor example, if the space in a coaxial cable
which is ordinarily occupied by air or other dielectric be filled with a
large number of coaxial cylindrical tubes which are alternately conduct-
ing and insulating, the cable will propagate various transmission modes,
and under the proper circumstances some of these modes will exhibit
lower attenuation constants than the transmission mode in a conven-
tional coaxial cable of the same size at the same frequency.

Experimental verification of Clogston’s theory of laminated conductors
has been obtained? at the Bell Telephone Laboratories, and the trans-
mission properties of a line filled with laminated material have also been
measured at these Laboratories and found in reasonable agreement with
theory. However experiments with structures as complex as those pro-
posed by Clogston are by no means simple, and the experimental work
on laminated conductors is still in an early, exploratory stage. Inasmuch
as the experiments are necessarily time-consuming, it has been thought

1A.M. Clogston, Proc. Inst. Radio Engrs., 39, 767 (1951), and Bell System Tech.
J., 30, 491 (1951). References will be to the Bell System Technical Journal article,
although ocr for equation numbers the two papers are identical.
2H ack, C. O. Mallinckrodt, and 8. P. Morgan, Jr., Proc. Inst. Radio
Engrs., 40, p. 362 (1952).

we discuss losses due to imperfect dielectrics and lossy magnetic ma-
terials.

Part II will be largely devoted to transmission lines of the so-called
“Clogston 2” type, in which the entire propagation space is filled with
the laminated medium, though to a lesser extent we shall also consider
transmission lines having an arbitrary fraction of their total volume
filled with laminations and the rest with dielectric. We shall first derive
expressions for the propagation constant and the fields of the lowest
Clogston 2 mode assuming infinitesimally thin laminae, so that the
attenuation constant is essentially independent of frequency, and then
go on to investigate the transition of the lowest Clogston 1 mode into
the lowest Clogston 2 mode as the space occupied by the main dielectric
is gradually filled with laminations. We shall also discuss the higher
modes which can exist in Clogston 1 and Clogston 2 lines with infinitesi-
mally thin laminae. Next the effect of finite lamina thickness on the
variation of attenuation with frequency in a Clogston 2 will be investi-
gated, and then the important question of the influence of nonuni-
formity of the laminated medium on the transmission properties of the
line. We shall conclude with a short section on dielectric and magnetic
losses.

Insofar as possible, plane and coaxial lines will be treated together
throughout the paper. Since however Bessel functions are not so easy
to manipulate as hyperbolic functions, there will be a few cases where
explicit formulas are not yet available for the cylindrical geometry. In
these cases the formulas derived for the parallel-plane geometry usually
provide reasonably good approximations, or if greater accuracy is desired
specific examples may be worked out numerically from the fundamental
equations in cylindrical coordinates.

The purpose of the present paper is to set up a general mathematical
framework for the analysis of laminated transmission lines, and to treat
the major theoretical questions which arise in connection with these
lines. In view of the length of the mathematical analysis, we have not
devoted much space to numerical examples, although a large number of
specific formulas are given which may be used to calculate the theoretical
performance of almost any Clogston-type line that happens to be of
interest. A considerable part of our work is directed toward evaluating
the effects of deviations from the ideal Clogston structure. Both theoreti-
cal and experimental results suggest that the limitations on the ultimate
applications of the Clogston cable are likely to be imposed by practical
problems of manufacture. These limitations, however, depend upon
engineering questions which we shall not consider here.

The quantity o is called the mtrinsic propagation constant and 7 the
intrinsic impedance of the medium.

We begin by considering structures bounded by infinite planes parallel
to the x-z coordinate plane, and we confine our attention to transverse
magnetic waves propagating in the z-direction. We assume that the
only non-vanishing component of magnetic field is H; , and that all the
fields are independent of x. Then the non-zero field components, written
to indicate their dependence on the spatial coordinates, are H,(y, 2),
E,(y, 2) and E,(y, z), the time dependence e“’‘ being understood through-
out. The field components are shown in Fig. 1. .

where g is the intrinsic propagation constant defined above. It is easy
to see that (4) is satisfied by a wave function of exponential form, say

ponents corresponding to any particular H, are easily obtained from
equations (2). i

A concept important in what follows is that of wave impedances’ at
a point. For a wave whose field components are H,, EF, , E,, the wave
impedances looking in the positive and negative y- and z-directions at a

For waves of the type that we consider, Z; and Z, are functions of
y only, so that if two media having different electrical properties are
separated by the plane y = yo, the continuity of the tangential compo-

is quite important. The sheets are located at y = +4), as shown in Fig.
1, and the space between them is filled with a medium whose electrical
constants are €0 , Ho , Jo (Or oo , 0 , if we wish to use the derived constants).
From the symmetry of the boundary conditions it is evident that for
any particular mode H, must be either an even function or an odd func-
tion of y about the plane y = 0. Taking the even case first, we have

If we replace go + iwes by oo/m and xo by (a9 — 7)', the boundary con-
dition at y=3b, namely

2no
The transcendental equations (11) and (18) are satisfied by the propa-
gation constants of the various even and odd modes; presumably each
has an infinite number of roots, which we could find, at least in principle,
if we knew the explicit form of the function Z(y). We shall confine our-
selves here to deriving an approximate expression for the propagation

constant of the principal mode (lowest even mode) when the walls are
very good conductors.

If the walls were perfectly conducting we should have Z(y) = 0,
and the lowest root yo of (11) would be given by

The principal mode between perfectly conducting sheets is just an un-
disturbed slice of the plane TEM wave which could propagate in an -
unbounded medium. If Z(yo) is not rigorously zero, but still so small that

and if Z(y) does not vary rapidly with y in the neighborhood of yo,
then the lowest root of (11) is given approximately by

where the second term is the first-order change in y due to the finite
impedance of the walls. If we formally set go = 0 (this does not actually
restrict us to perfect dielectrics since we could still assume eo or po to be
complex), we have

As an example of the use of (20) and (21), suppose that the impedance
sheets in Fig. 1 are electrically thick metal walls of permeability uw and
(high) conductivity gi. Then to a very good approximation at all en-
gineering frequencies and for all ordinary dielectrics between the walls,
the surface impedance is

is the skin depth in the metal. We thus obtain from (20) and (21) the
familiar formulas

and
OE ,/dp — dF ,/dz = iwpnH, , (28)
from which we can eliminate Hp and EF, to obtain
Hd _ 10H, Hz , @Hs 2
EEA ATOR ak cel ee = 2
ae ag ge ge eo

If we assume a wave traveling in the positive z-direction with propaga-
tion constant y and write

where «x is given by (6) as before. But (31) is just the equation satisfied
by modified Bessel functions of order one and argument xp, so

where A and B are arbitrary constants. The other field components can
be obtained from H, using (27); the results are

For cylindrical fields of the type that we are considering, the wave-
impedances looking in the positive and negative p- and z-directions at a.
typical point are defined to be, respectively,

We shall now discuss the propagation of circular transverse magnetic
waves in a homogeneous region of space whose electrical constants are
€) , Mo, Jo (OF oo, m0), and which is bounded by coaxial cylinders of radii
p1 and p2, where p2 > p1, aS Shown in Fig. 2. We suppose that the radial
impedances looking from the main dielectric into the inner and outer
cylinders are, respectively,

If equations (36) are to be satisfied by values of A and B which are not
both zero, it is easily shown that a necessary and sufficient condition is

nop o(Kop1) ae Z1(y) Kx (kop) a nopL& o(Kop2) = Za(y) Ki (Kops) (38)
noolo(Kopr) — Zi(y)L1(Kopi) nopLo(Kop2) + Ze(y)L1(Kop2)

and (88) is a transcendental equation for the determination of the propa-
gation constants of all the circular magnetic modes in the coaxial line.

As in the discussion of the parallel-plane line, we shall confine our
attention to the principal mode and shall assume forthwith that the wall
losses are small.’ Since for the principal mode we expect that y will be
nearly equal to oo , we may write yo for oo and evaluate Z; and Z:2 at yo ;
and we may replace the modified Bessel functions in (88) by their ap-

5 J. A. Stratton, Electromagnetic Theory, McGraw-Hill, New York, 1941, pp.
551-554, gives a similar treatment of the principal mode in an ordinary coaxial
cable with solid metal walls.

proximate values for small argument. From the series given in Dwight*
813.1, 813.2, 815.1, and 815.2, we have

for |«| < 1, where log represents the natural logarithm. If we put
these approximations into (88) and if we suppose that the wal] im-
pedances are so small that

2 2 2
= _ = : 41
ieee cae no log (p2/p1) oe
Now further assuming that
1 | Zi(yo)/p1 + Ze(yo)/pe |
= | —— | <1 42
8 | ~ cum log (on/o1) 7 ae
we get by the binomial theorem
Z
eee (yo) /pi + Zelyo)/pe (43)

If we formally set go = 0, we find that the attenuation and phase con-
stants of the principal mode in a coaxial line with low-loss walls and no
dissipation in the main dielectric are

As before, these approximations for a and 8 will ultimately break down
as the frequency approaches zero, but they will certainly be valid over
the frequency range in which we are interested in the present paper.

tion, Macmillan, New York, 1947. We shall refer to Dwight for a number of stand-
ard series expansions.

The magnetic field lines of the principal mode will of course be circles
and the electric field will be largely radial, but with a small longitudinal
component unless the wall impedances are rigorously zero. The general
expressions (83) for the fields may be reduced to simple approximate
formulas if we use the fact that xo is given by (41) and xp is small com-
pared to unity. The ratio A/B may be obtained from either of equations
(36). Introducing the approximations (39) for the Bessel functions and
carrying out a little algebra, we get the following approximate expres-
sions for the fields:

where the amplitude factor J is equal to the total current flowing in the
inner cylinder. Incidentally we note that the above results might have
been derived from more elementary arguments if we had started with the
fields in a coaxial line with perfectly conducting walls and treated the
effect of finite wall impedance as a small perturbation.

If we consider an ordinary coaxial cable with solid metal walls at a
frequency high enough so that there is a well-developed skin effect on
both conductors, then to a good approximation

If necessary we may take account of dissipation in the main dielectric
of either a plane or a coaxial transmission line by assigning complex
7
values’ to « and po, say

7 See, for example, C. G. Montgomery, Principles of Microwave Circuits, M. I. T.
Rad. Lab. Series, 8, McGraw-Hill, New York, 1948, pp. 365-369 and 382-385.

tenuation due to dielectric and magnetic losses,
ag = Reo = Re toW nei(1 — 7 tan do) (1 — 7 tan $0)
Lieya/ ule, (tan go + tan £0),
provided that tan ¢ and tan {o are both smal] compared to unity, as they
will always be in practice. We shall neglect second-order effects and so

regard the dielectric losses, the magnetic losses, and the wall losses as
additive.

in a layer of homogeneous, isotropic material whose electrical constants
are ¢, wu, g (or o, 7), and which is bounded by planes perpendicular to
the y-axis. Henceforth we shall always assume that the z-dependence of
every field component is given by the factor e ”*, where the complex
quantity y, whose value may or may not be known a priori, is the propa-
gation constant of the wave in the z-direction. Then the first of Maxwell’s
equations (2) yields

Ey = —ly/Qg + twe)|Hz , (52)
and on eliminating EH, from the other Maxwell equations, we get
0H,/dy = —(g + wwe)E.,

Now if we formally identify H, with “current” and FE, with
“voltage”, equations (53) are just the equations of a uniform one-dimen-
sional transmission line extending in the y-direction, with series im-
pedance «’/(g ++ iwe) per unit length and shunt admittance (g + iwe)
per unit length; in other words a transmission line whose propagation
constant is x and whose characteristic impedance is 7, , where

Hence we can apply the whole theory of one-dimensional transmission
lines with the assurance that in so doing we shall not violate the field
equations. For example, if £(0), H(O) and E(d), H(t) represent the
tangential field components E,, Hz at two planes separated by a dis-

tance t, these fields are related by the general circuit parameter matrix
of a uniform line, namely

We are now in a position to determine the surface impedance normal
to a laminated plane structure composed of layers of which every other
one has thickness 4, and electrical constants o,, m, while the inter-
vening layers each have thickness ¢, and electrical constants o2, m2.
Fig. 3 shows the cross section of such a stack in which the total number
of double layers is n (2n single layers), while Fig. 4 represents the corre-
sponding coaxial stack. Ultimately we shall assume the layers of thickness
i: to be good conductors and those of thickness f to be good insulators,
but these assumptions need not be brought in immediately.

If the fields in the plane stack all vary with z according to e ”, then
when we look in the direction of increasing y each double layer may be
regarded as a four-terminal network formed by two sections of uniform
transmission line of lengths 4; and &, the propagation constants and
characteristic impedances of the two sections being given respectively by

The matrix of the double layer is the product of the matrices of the two
single layers in the proper order. Thus if the tangential field components
are Ey , Hy at the lower surface of the first layer and EH, , H; at the upper
surface of the second layer, we have

The stack of double layers may be regarded as a chain of iterated four-
poles; such chains have an extensive literature.’ The relation between
the tangential fields Z, , H, at the upper surface of the nth double layer
and Hy, Ho at the lower surface of the first double layer is

where M is the @@CD-matrix appearing in equation (57). However there
is a simple expression” for the nth power of a square matrix of order
two, namely

I, (60)
where I is the unit matrix of order two, I is the propagation constant
per section of the chain of four-poles, defined by

The determinant of the matrix whose elements are given by (58) is unity,
as may easily be verified; but this may not be the case for all the matrices
which occur in our study of cylindrical structures. M will therefore be
carried explicitly in the following equations.

Ky, is the impedance seen when we look into a semi-infinite stack of
double layers if the first layer is of type 1, while K» is the impedance seen
if the first layer is of type 2. In calculations relating to Clogston 1 lines ©
with dissipative walls, the real parts of Ki and Ke will both be positive.
By a straightforward procedure we may express the matrix elements
@, ®, ©, D in terms of Ki, Ke, T, and M, and then transform equation

10 F. Abelés, Comptes Rendus, 226, 1872 (1948). This result was called to the
author’s attention by Mr. J. G. Kreer.

Finally we obtain from (59) and (64) an expression for the impedance
Zo looking into a plane stack of n double layers when the nth layer is
backed by a surface whose impedance is Z,, , namely
gz, - Eo 17,(Kye™ + Kee ™") + KiKe sh nI 65)
° Ay Z, sh nt + 4(Kie * + Ke")

For the cylindrical geometry, matters are a good deal more compli-
cated. If we consider waves having field components H,, E,, Zz in a
homogeneous, isotropic shell bounded by coaxial cylindrical surfaces,
and assume a propagation factor e ”, Maxwell’s equations (27) and (28)
may be written .

If desired, we might identify /, with ‘‘voltage” and — pH, with ‘“‘current”’
and regard equations (67) as describing a nonuniform radial transmis-
sion line, having series impedance «’/(g + iwe)p per unit length and shunt
admittance (g + twe)p per unit length. Since, however, in equations (34)
we have already defined the radial wave impedance to be a field ratio
without the extra factor of p, we shall carry out the analysis of the
present paper directly in terms of the field components H, and —H,.

From the general expressions (83) for the fields in cylindrical co-
ordinates, we can show that the matrix relation between the tangential
field components HL, , —H, at two radii p: and pe is given by

It may be verified that the determinant M of the square matrix ap-
pearing in (68) is simply

where terms of the order of ¢/p; represent the first-order curvature cor-
rections. If we use the same value of p; , say p, for both parts of a double
layer, then up to first order the elements of the matrix of the double
layer become

As in the analogous equations (58) for a plane double layer, the sub-
scripts 1 and 2 refer to the first and second layers respectively.

If we have a stack of double layers in which all the layers of the same
kind have the same thickness and same electrical constants, then the
only term in (73) which varies from one double layer to the next is the
mean radius p. Depending on the circumstances, we may wish to use a
single value of p for the whole stack, or a few different values, or even,
if high-speed computing machinery is available to carry out the matrix
multiplications, a different value of j for each double layer. The matrix
of the whole stack then becomes a product of powers of as many different
matrices as we have chosen values of p. Obviously this method is better
adapted to the numerical analysis of special cases than to the general
theoretical treatment of a stack whose ratio of outer radius to inner
_ radius is unspecified.

In principle we are now able to compute the normal surface impedance
of any laminated plane or coaxial stack at a given frequency provided
that we know the electrical constants and the thickness of each layer,
the number of layers, the propagation constant y in the z-direction, and
the normal impedance Z,, of the material behind the last layer. Since the
general formulas even for plane stacks are quite complicated, however,
we shall introduce at this point some very good approximations which
will be valid for all of the following work.

Henceforth we shall take the layers of thickness é, to be such good
conductors that the ratio we:/g: of displacement current to conduction
current is negligible in comparison with unity. For metals like copper
this is an excellent approximation at even ‘the highest engineering fre-
quencies. Then on introducing the characteristic skin thickness 6; , we
have for the conducting layers,

a= Viowg = (1+ 1)/4, i
m= Vion/g = (1+ )/qd,
where
b= V2/omgn - (75)
For pure copper the permeability and conductivity are

where fye is the frequency in Mc-sec”. Referring to equations (56)
and (69) and bearing in mind the above numerical values, we see that
for the conducting layers we have

to a very good approximation, since in our applications the quantity y
will always be of the order of 27i/d, , where the vacuum wavelength
dy is at least a few meters, while the skin thickness 6; will be at most a
small fraction of a centimeter.

For the insulating layers of thickness f we shall set the conductivity
gz equal to zero, so that

We denote the relative dielectric constant and permeability by e, and
ber respectively; dissipation in the insulating layers may be included

if necessary by making e, and/or 2, complex. In MKS units we have
€2 = €2r€y M2 = Herby y (81)
where the electrical constants of vacuum are

= a (82)
My = 1.257 X 10 ~ henrys-meter ~.
It follows that
Qrt €or 2WUf ye WV par €oy us
02 = CoV pore = a Sr ee Vee f Sone. “r meters (93)

where as usual the subscript v refers to vacuum. It is clear that unless
we deal with ferromagnetics, the quantities c2 and 7 will be of roughly
the same order of magnitude as o, and yn, . From (56) and (69) we have

where since o2 and y are both of the same order of magnitude as 2z7/), ,
in general no further approximations can be made.

In all of what follows we shall assume that the thickness t of each
insulating layer is very small compared to the vacuum wavelength at
the highest operating frequency; in practice ft, will be at most a few mils
and X, at least a few meters. Then the quantity | kete |, which is of the
order of 27t2/d, , will be so small that to an excellent approximation we
may set sh kote = kolo and ch kof, = 1. Using this simplification, together
with the fact that m, <« m2, for all frequencies which may conceivably
be of interest, it is not difficult to show from (58) that the matrix ele-
ments of the plane double layer reduce to

DOD = Maytals sh Kyly + ch Kil .
My
The determinant of the matrix is unity, and from (61) the propagation
constant per section is defined by

Ky = = pMaykale + -V (Snoykote)? + miynoykele coth «ti + iy » (87)
Ko = + omaykate + V (Sneykete)? + mameykete coth mtr + ni, -

If we make the same simplifications in the approximate expressions
(73) for the matrix elements of a coaxial double layer, we obtain

In the preceding equations no restrictions have been laid on the
thicknesses t; and ¢: except the trivial requirement that t, shall be small
compared to a wavelength. We shall now consider the limiting case in
which both ¢; and ¢é, are infinitesimally small. When we make this last
and most drastic approximation we do not expect that the idealized
structure thus obtained will show all of the features which are of interest
in a physical transmission line with finite layers; but the results of the
simplified analysis will be useful in some cases nevertheless. It need
scarcely be pointed out that we are dealing here only with a mathematical
limiting process, in which we assume that each layer, no matter how
thin, always exhibits the same electrical properties as the bulk material.
If this assumption be regarded as unrealistic, it may be observed that
the quantity which we actually allow to tend to zero is the ratio of layer
thickness to skin depth. The skin depth may be made as large as desired
by lowering the frequency, so that the formulas which we derive by

letting t, and t, approach zero at a finite frequency will also hold for finite
thicknesses if the frequency is sufficiently low.

We shall let .@ denote the fraction of the stack which is occupied by
conducting material, so that

where at present t; and ¢; are both infinitesimal. Then the stack may be
regarded as a homogeneous, anisotropic medium, characterized by an
average dielectric constant € perpendicular to the layers, an average
permeability @ parallel to the layers, and an average conductivity @
parallel to the layers. Sakurai” has treated such an artificial anisotropic
medium, and from his formulas we find that when the layers are al-
ternately conductors and insulators, the average electrical constants are,
to a very good approximation,

Sakurai has also shown that the average values of the electrical con-
stants may be used in Maxwell’s equations for the average (macroscopic)
fields, due regard being paid to the orientations of the field vectors with
respect to the laminae.

where s is the thickness of the stack. The propagation constant TI, per
unit distance normal to the stack and the characteristic impedance K
of the stack are given by

I, and K may also be derived from equations (86) and (87) by limiting
processes; we have

It should perhaps be noted that terms of the order of we/g: and weo/gi
compared to unity were omitted in the expressions (90) for @ and g,
and in the derivations of IT, and K. Since, however, under all practical
circumstances the omitted terms appear to be insignificant, we shall
not take space to write out the formally more complicated results which
would be obtained by keeping them.”

In a cylindrical stack of infinitesimal layers, the average fields satisfy

0H,/dz2 = —iwéE, ,
O(pH5)/dp = GoL. , (97)
dE./dp — dE, /dz = iwpH, .
The relation between the tangential field components L,, —H, at two

12 Tn Reference 1, equations (II-17) through (II-26) give examples of equations
in which these small terms have been retained.

IV. PRINCIPAL MODE IN CLOGSTON | LINES WITH INFINITESIMALLY THIN
LAMINAE

An idealized parallel-plane Clogston 1 transmission line is shown
schematically in Fig. 5. It consists of a slab of dielectric of thickness b,
with electrical constants yo , € , bounded above and below by laminated
stacks each of thickness s. Outside each stack there may be an insulating
or a conducting sheath, of which nothing more will be assumed at present
than that its normal surface impedance Z,(y) is known. The total dis-
tance between the sheaths will be denoted by a, where a = b + 2s.

The corresponding Clogston 1 coaxial line is shown in Fig. 6. We de-
note the thickness of the inner and outer stacks by s; and s2 respectively,
while a is the radius of the inner core (if any), and b is the inner radius
of the sheath around the outer stack. The inner and outer radii of the
main dielectric are p; = a + s, and p. = b — 82, respectively. In practice
the core may be a dielectric rod and the sheath may be a conducting
shield, but in the present theoretical analysis we shall merely assume that
the radial impedances Z,(y) and Z,(y) looking into the core and the
sheath are known.

In Part I of this paper we shall deal with “extreme” Clogston 1 lines,
in which the space occupied by the stacks is small compared to the
space occupied by the main dielectric. We may then regard the laminated
boundaries as impedance sheets guiding waves whose phase velocity is

determined by the properties of the main dielectric, as discussed in Sec-
tion II, and we may use the intrinsic propagation constant of the main
dielectric in calculating the surface impedance of the boundaries. This
approximation simplifies the analysis of Clogston 1 lines a great deal.
We shall treat the general case, in which an arbitrary fraction of the
total space is filled with laminations, in Section IX of Part II, as a part
of our study of Clogston 2 lines.

In this section we shall assume that the laminae are infinitesimally
thin, so that the stacks may be completely characterized by their average
properties é, 7, and g. The case of finite laminae will be taken up in the
next section. We shall also assume throughout that dielectric and mag-
netic dissipation may be neglected except, as in Section VII, where the
contrary is explicitly stated.

In general the current density and the other field quantities in a plane
stack of infinitesimally thin layers will be linear combinations of the
functions sh T',y and ch Ty, where y is distance measured into the stack,
and I, is the propagation constant per unit distance, as given by (98).
The qualitative behavior of the fields in a cylindrical stack will be similar.
In particular, if the stack is thick enough the current density and the
fields will fall off as ¢ ", and we can define an “effective skin depth” A by

ohmic losses in a stack carrying a fixed total current the current density
should be uniform across the stack, and that we can achieve uniform cur-
rent density by adjusting the poe product of the main dielectric so as to
make I, equal to zero. If in equation (93) we set

Equation (102) will be referred to henceforth as Clogston’s condition’
If the permeabilities of the various materials are all equal, the condition
reduces to

When Clogston’s condition is satisfied, Tz = 0 and the effective skin
depth of the stack is infinite; that is, the current density is uniform in
any stack of finite total thickness. The quantities [, and K vanish
simultaneously, but the limiting value of their ratio is finite; and the
matrix of the plane stack, as given by (92), takes the form

which is, as might have been expected, just the impedance between
opposite edges of.a unit square of material of conductivity g and thick-
ness s through which the current density is uniform, in parallel with the
sheath impedance Z,(yo). It follows from equations (20) and (21) of
Section II that the attenuation and phase constants of the principal mode
in a plane Clogston 1 line with infinitesimally thin laminae, Clogston’s
condition being satisfied exactly, are

3 This statement is certainly accurate enough for all practical purposes, al-
though an exact calculation which takes into account the small terms that were
neglected in the approximate formula (93) for fT, shows that the effective skin
depth is \o/276, where Xo is the length of a free wave in the main dielectric. The
exact result is derived by Clogston in Reference 1, equation (II-26). In practice,
finite lamina thickness will restrict us to effective skin depths much smaller than
this theoretical limit.

whereas 6s will usually be several times the skin thickness 6; in the fre-
quency range of interest. If the sheath is free space, its impedance is a
fortiori much greater than 1/gs, since then it may be shown that

where 7,» = 376.7 ohms is the intrinsic impedance of free space, and
Mor and €, are the relative permeability and relative dielectric constant
of the main dielectric. Under most circumstances, therefore, we may
neglect 1/Zn(yo) in comparison with gs, and obtain the very simple
results,

To this approximation the line exhibits neither amplitude nor phase
distortion.

For a coaxial stack of infinitesimally thin layers with Clogston’s con-
dition satisfied, the stack matrix given in (98) reduces to

where po and p, denote the inner and outer radii of the stack. It follows
from (112) that

where Z;(yo) and Z2(yo) are the radial impedances looking into the stacks
at pi and pe respectively, and Z,(yo) and Zs(yo) are the radial impedances
looking into the core and the outer sheath. From equations (44) and
(45) of Section II, the attenuation and phase constants of a coaxial Clog-
ston 1 cable with infinitesimally thin layers, Clogston’s condition being
satisfied exactly, are

Z1(yo)/p1 + Zo(-yo)/p2
I 115
pea tla teal ca
where Z1(yo)/pi and Ze(yo)/p2 are given by (118).
The impedances Z,(yo) and Z,(yo) may be computed if we know ihe
structure of the core and the sheath. For a solid, homogeneous core and
a homogeneous sheath of effectively infinite thickness, we have

but of course the intrinsic propagation constant o and the intrinsic im-
pedance 7 need not be the same for the core and the sheath. If the sheath
is of finite electrical thickness or has a laminated structure (alternate
layers of copper and iron, for example, to provide effective shielding),
its surface impedance may be calculated by a straightforward but longer
procedure. We shall not go into this matter here, but shall merely observe
that in many cases of interest Za(yo) and Z(yo) are so large that we
may neglect the terms containing their reciprocals in (118). This means
that we neglect the total conduction and displacement currents flowing
in the core and the sheath, compared to the conduction currents in the
stacks. Then the expressions for the attenuation and phase constants
become

and again to this approximation there is neither amplitude nor phase
distortion.
The formulas which have just been derived on the assumption of

for —3b S y S 4b, where H is an arbitrary amplitude factor and
Zo(yo) is given by (105). In the stacks the fields are
He & Hill + gZo(vo)(3b F y)le™,

for 36 S$ |y| S $a, where in cases of ambiguous sign the upper sign
refers to the upper stack (y > 0) and the lower sign to the lower stack
(y <0). It should be noted that whereas the tangential field components |
H, and E, are continuous through the stack, the normal field component
E, is discontinuous at layer boundaries. From equation (52) we have, in
the conducting layers,

To our approximation, therefore, the only contributions to the average
field #, come from the insulating layers.
The average current density J, in either stack is uniform, being

The total current per unit width carried by the stack is just J.s, where
s is the thickness of the stack; there will also be small currents in the
sheaths unless we assume the sheath impedance to be infinite. The
potential difference between any two points y: and 72 in the same trans-
verse plane may easily be found from

For a Clogston 1 line of the proportions which we have been considering,
the potential difference across the stacks will be small compared to the
potential difference across the main dielectric.

In a coaxial Clogston 1 with infinitesimally thin laminae, the fields in
the main dielectric are given to a good approximation by equations
(46) of Section II, namely

where J is an arbitrary amplitude factor and Z,(yo) and Zo(yo) are ex-
pressed by (113). In the inner stack we have

The average current density in either stack is uniform and is given by
J, = gE. , (129)

though in general the current density will not be the same in the two
stacks because of the difference in cross-sectional areas. The potential
difference between the surface of the inner core and any other point in
the same transverse plane is

If the stacks are thin compared to the thickness of the main dielectric,
as we are assuming throughout Part I, then the potential difference
across the stacks will be small compared to the potential difference
across the main dielectric, and the characteristic impedance Z; of the
Clogston 1 cable will be approximately the same as the characteristic
impedance of an ideal coaxial cable with perfect conductors of radii
pi and p2 and the same main dielectric, nome,

Z, = 60 ~ log = = ohms. (131)
Or
We shall defer making any field plots i Clogston-type transmission
lines until Section IX of Part II, when we shall discuss the transition
from Clogston 1 to Clogston 2 as the space originally occupied by the
main dielectric is gradually filled with laminations. Our present results
will then appear as the limiting case in which the thickness of the stacks
is small compared to the thickness of the main dielectric.
In conclusion we shall mention briefly the question of how to dispose
a given amount of laminated material in a Clogston 1 coaxial cable so as
to achieve the minimum attenuation constant. The whole problem of
optimum proportions for Clogston cables is a complicated one of which
an adequate treatment would require a separate paper in itself, with
the results depending to a great extent on engineering considerations
which limit the ranges of the parameters that we can vary in any practical
case. Here we shall discuss only the following rather highly idealized
problem:

Given a coaxial Clogston 1 with infinitesimally thin laminae, having a
high-impedance core and a high-impedance sheath of fixed radius b,
and in which the total thickness s; + se of both stacks is a fixed constant
2s. Assuming that 2s is small compared to b, what should be the radius
a of the core, and how should the total stack thickness be divided between
the outer and inner stacks so as to minimize the attenuation constant of
the line? Finally, what should be the fraction @ of conducting material
in the stacks to minimize the attenuation constant, if the electrical
constants of the conducting and insulating layers are fixed, but the
properties of the main dielectric are at our disposal?

and vary s; and s2 in accordance with this relation while holding a and b
constant, it is easy to show that the expression on the right side of
(133) is a minimum when

Va + Vo’ Vat Vb
These equations tell us the most efficient way to divide the stacks in a
Clogston 1 when the radii of the core and the outer sheath are a and b re-
spectively, still assuming of course that the thickness of each stack is
small compared to its mean radius.
If we introduce the optimum values of s; and s2 into (183), we get

If b is fixed, the last expression is a minimum, considered as a function
of a, when

attenuation constant of a Clogston 1 cable with infinitesimally thin
laminae and high-impedance boundaries may be written in the form

where the first factor depends on the electrical constants of the com-
ponents of the cable, while f(a, b, s: , s2) isa function only of the geometry.
By (110) the attenuation constant of a plane Clogston 1 has the same
form, only with a different dependence on the geometrical factors. Now
assume that the geometrical proportions of the line are fixed, and that
the electrical constants 41, 91, we, and e of the conducting and insula-
ting layers are given, but that the constants yo, « of the main di-
electric and the fraction of space @ occupied by conducting layers in
the stacks are at our disposal. The woe product of the main dielectric
is to be codetermined with 6 so that Clogston’s condition (102) is always
satisfied. Solving (102) for 6 gives

Ho€o — [2€2
=. 142
Hoéo +- (yu = M2) €2
Hence the first factor in the expression (141) for a may be written

If we minimize the right side of (143) with respect to 9 , all other quanti-
ties being held constant, by equating to zero the derivative with respect

G= (145)
Sui i (uj ae Surya)?’
and the corresponding attenuation constant is proportional to
(€o/ U0)" = (€o/ uo)? Bur + (ui + Sprue)? (146)

It will be observed that so far we have determined only the optimum
value of the product poe) , and so we are still free to alter the ratio of
Mo to € while holding the product of these two quantities constant. For
given values of 4 and pe , we obtain the lowest attenuation constant by
making ¢€ as small as possible and yo as large as possible, subject of course
to the practical restriction that « cannot be lower than the dielectric
constant of free space. However if we permit ye and po to be simul-
taneously increased, as by magnetic loading of both the insulating layers
and the main dielectric, we find from (146) that on paper it is possible
to decrease the attenuation constant without any definite limit. This
observation is in accord with the fact that the attenuation constant of
an ordinary coaxial cable may be decreased indefinitely, with a corre-
sponding decrease in the velocity of propagation along the cable, if we
are willing to assume an unlimited amount of lossless magnetic loading.

When po = pe, corresponding to no magnetic loading, we must take
€é) = 3e,, and (148) reduces to

magnetic material in the stacks (u. = we = p,), and the optimum propor-
tions given by (189) and (147), we have
4.857

impedance of the main dielectric, which cannot be air in a Clogston cable.

attenuation constant a, of an air-filled standard coaxial of the same size,
made of the same conducting material, is

where por and ¢€; refer to the main dielectric of the Clogston cable.
Since the attenuation constant of a standard coaxial cable is propor-
tional to the square root of frequency in the range we are considering,
while the attenuation constant of the ideal Clogston cable is independent
of frequency in this range, there will be a crossover frequency above
which the Clogston cable has a lower attenuation constant than a con-
ventional coaxial cable of the same size. If we are dealing with copper
conductors and if frequencies are measured in Me-sec and linear
dimensions in mils, then from equations (78) and (153) we find that the
crossover frequency is given approximately by |

me, 49.50 (€0,/ Mor)
°™ (st + 82)inits
For example, let us take an ideal Clogston 1 cable of outer diameter

0.375 inches, excluding the sheath, with no magnetic loading, and assume
the following values:

This cable has a lower attenuation constant than a standard air-filled
coaxial of the same size at frequencies above about 1 Me-sec ’, the ap-
proximate formula (154) yielding 0.955 Me-sec™ for the crossover
frequency and the exact equation (118), taken in conjunction with (151),
yielding 1.251 Mc-sec™.

The reader is cautioned that the comparison given by (153) between
Clogston and conventional cables is based upon certain highly idealized
assumptions. In the first place we have neglected the finite thickness of
the laminae, which will in fact cause the attenuation constant of a
_ physical Clogston cable to increase with increasing frequency, and
ultimately to cross over again and become higher than the attenuation
constant of a conventional air-filled coaxial. We have also neglected
dielectric and magnetic losses, which are likely to be directly propor-
tional to frequency and by no means negligible at the upper end of the

frequency band. In practice, too, the poco product of the main dielectric
must be held very close to the Clogston value or the benefit of the large
effective skin depth is lost; and the stacks must be extremely uniform or
again the depth of penetration is greatly reduced. We shall take up all
these matters in later sections, and shall see that while the results just
given represent ultimate limits of performance, the practical improve-
ments which can be achieved over conventional cables depend upon the
degree to which one can solve the manufacturing problems that tend to
make every actual Clogston cable differ more or less from the ideal struc-
ture considered above.

V. EFFECT OF FINITE LAMINA THICKNESS. FREQUENCY DEPENDENCE OF
ATTENUATION IN CLOGSTON 1 LINES

The principal effect of finite lamina thickness in a Clogston cable is to
introduce a frequency dependence into the propagation constant, and
in particular to cause the attenuation constant to increase, with increas-
ing frequency, above the value which we have found for infinitesimally
thin laminae (or for finite laminae at low frequencies). The increased
losses are associated with the fact that the penetration depth in a lami-
nated stack decreases with increasing frequency, even when Clogston’s
condition is exactly satisfied, if the laminae are of finite thickness. We
shall now obtain expressions for the surface impedance of a plane lami-
nated stack of n double layers, such as is shown in Fig. 3, when Clogston’s
condition is satisfied but the individual layers are of finite thickness.

where in the last step we have used the fact that in the conducting
layers m1y is equal to 7 and x, is equal to o; to a very good approximation.
We now introduce the dimensionless parameter

which may be regarded as a measure of the electrical thickness of the
individual conducting layers. From (86) and (156) we have, for the prop-
agation constant per double layer,

Ky = a [— 20 + 6? — © coth © + 1)4,
since My = Ki/Ga = O/giti
If the thickness ¢, of each conducting layer is moderately small com-
pared to the skin depth 6, at the highest frequency of interest, the quanti-
ties T, Ky, and Ky may conveniently be expanded in powers of ©. The
identity

after we expand sh © and ch 9 by Dwight 657.1 and 657.2 and collect
terms. Taking the square root by the binomial theorem gives

the negative sign being introduced because from (157) ©” is a positive
imaginary number and we want Re T > 0. Then

er ae 170! >
Paes | - zal?’ +4 c+:
i fe. & Q° |
--3.|5+% + met aa
provided that we expand the sh’ function by Dwight 706. From (159)
we get
a. ee. @? + iVv3 @! iv’3 ef + |,

where we have expanded coth © by Dwight 657.5 and chosen the sign
of the square root to make Re K, and Re K; both positive.

Our first observation is that when the lamina thickness is finite the
effective skin depth of the stack is also finite. We have, from (157) and

(163),
1 E it} Sti |
=—~|/54+—4-—57-"'' (165)
| V3 Le 158; 52561 :
and the average propagation constant per unit distance into the stack is

Re I, ti mug: fte tt
a result also given by Clogston.” The number N of double layers in one
effective skin depth is

T, is essentially the thickness of conducting material in each stack which
is effectively carrying current; it is evident that for small values of 4/6
this effective thickness is inversely proportional to the frequency f and
to the thickness t; of the individual conducting layers, but independent
of the thekness 2 of the insulating layers, provided that tf is very small
compared to the length of a free wave in the insulating material.

In the general case, still assuming of course that Clogston’s condition
is satisfied, the surface impedance Zo(yo) of a plane Clogston stack is
given by equation (65) of Section III, namely

924 THE BELL SYSTEM TECHNICAL JOURNAL, SEPTEMBER 1952
where Z,,(yo) is the impedance of the surface behind the stack. If © = 0,
(170) reduces to (105) of Section IV, that is,

1 1
Gs + 1/Zn(yo.) = Ti + 1/Zn(y0)’
where 7; is the total thickness of conducting material in the stack. If
Zn(Yo) is infinite, then for all values of © and n we have

and if Re nT is large, corresponding to a stack many effective skin depths
thick, then for any Zn(yo) we have

Once Zo(yo) has been computed for a particular frequency, the at-
tenuation and phase constants of the plane Clogston 1 line at that fre-
quency are given, as in Section II, by

Explicit expressions for the surface impedance of a coaxial stack of finite
layers have not been derived. However, if in a coaxial Clogston 1 the
thickness of each stack is small compared to its mean radius, or if the
depth of penetration given by (167) is small compared to the radius of
the surface near which the currents flow, then the parallel-plane formula
(170) may be used for the stack impedances Zi(yo) and Ze(yo) which are
to be substituted into the equations of Section II for the attenuation
and phase constants, namely

WV Moe + Im Ona log palo) . (177)
If the plane approximations are regarded as insufficiently accurate, one
can compute the surface impedance of a cylindrical stack by repeated
multiplication of matrices similar to the one given by equations (88) of
Section III. This procedure would obviously involve considerable numeri-
cal computation, but we can hardly expect that it would reveal anything
qualitatively new for Clogston cables of the proportions considered in
Part I.

It will be instructive to compare the impedance of a laminated plane
stack with the impedance of a solid metal plate over the full frequency
range from zero to very high frequencies.” If the stack contains n con-
ducting layers, each of thickness ¢, , and the metal plate is of thickness
T, = nt, , the impedances of the plate and of the stack will be equal at
zero frequency, and also at very high frequencies where the first layer
of the stack is already many skin depths thick. For simplicity we assume
that both the plate and the stack are backed by infinite-impedance
surfaces at all frequencies.

To orient ourselves we shall define three critical frequencies, for which
respectively the thickness of the solid plate is equal to one skin depth in
the metal, the thickness of the stack is equal to one “effective skin depth”’,
and the thickness of a single conducting layer is equal to ~/3 skin depths
in the metal. These frequencies are

The approximate forms of the surface impedance functions of the plate
and the stack in the various frequency ranges are then quite simple.

In the range 0 S f S fi, the surface impedance of the solid plate is
approximately constant and given by

while in the range f = f; we see approximately the surface impedance of
an infinite plate,

which is proportional to +/f. The surface impedance of the stack is
approximately constant in the range 0 S f S fe, where

while in the range fo S f S f; it is approximately equal to the impedance
K, of an infinitely deep stack of moderately thin layers as given by the
first of equations (164), namely

18 In this connection see also Reference 1, Fig. 2, p. 494. Clogston compares
a laminated stack with a solid plate of the same total thickness as the stack, hence
a plate which contains more conducting material than the stack.

which is directly proportional to frequency (and independent of con-
ductivity). For f 2 f3 the stack acts much like an infinitely thick solid
plate, for which

By substituting the above series into equation (170), we can obtain
the variation of the stack impedance with frequency so long as ;/6; is
sufficiently small. Although in principle there would be no difficulty in
taking into account an arbitrary sheath impedance Z,(yo), for brevity
we shall restrict ourselves here to the case in which the sheath impedance
is so high that at all frequencies of interest the current in the sheath may
be neglected. Then we have equation (191) (see next page).

Fig. 7—Surface resistance R of solid plates and laminated stacks versus
feoqueney f on log-log scale.

= V3/(amgitiTs), (196)
at which frequency the approximate formulas yield
AR/Ro = 1/3, AX/Ry = V3. (197)

We may now answer the question: What must be the thickness ¢, of
the individual conducting layers in a plane stack which contains a given
total thickness 7, of conducting material, if at a specified top frequency
fm the resistance of the stack is not to have increased by more than a
specified small fraction of its de value? We find that the permissible

and we note that this value of ¢, is inversely proportional both to fm and
to 7, . If we measure ¢; and 7, in mils and f,, in Me-sec , then on putting

For a plane Clogston 1 with stacks of equal thickness, the attenuation
constant is given by (174), and the fractional change in attenuation
with frequency is equal to the fractional change in resistance of either
stack, as calculated from (193). For a coaxial Clogston 1 with stacks
thin enough so that the plane approximation is valid we may also use
(193), but the fractional changes in resistance will be different for the
two stacks if these are of different thicknesses, and the fractional change
in the attenuation constant must be calculated from equation (176).
Tf Rio and Rs are the de resistances “per square” of the two stacks, and
AR, and AR; their increments as obtained from (193), then the fractional
increase in attenuation is given approximately by

For either plane or cylindrical geometry we find that if we scale up a
particular Clogston line by multiplying the thicknesses of the stacks and
the main dielectric by the same factor, then the low-frequency at-
tenuation constant will be divided by the square of the scale factor.
However, the permissible thickness of the individual conducting layers,
if we are to have the attenuation flat to a specified degree up to a fixed
frequency, is inversely proportional to the scale factor. Thus if we double
the overall dimensions of the line and double the amount of conducting
material in the stacks, we shall divide the low-frequency attenuation
constant by four, but we shall have to make the individual layers half as
thick in order to maintain the same relative increase in attenuation
constant at the same top frequency f,, . In addition it is clear that if we
double the top frequency while maintaining the same requirement on
Aa/a for a line of given dimensions, we shall also have to cut the thick-
ness of the individual layers in half.

As a numerical example, let us return to the cable whose specifications
were given by (155) at the end of Section IV. For this cable we have:

If the conducting layers are copper, we find that equation (200) for the
fractional increase in attenuation becomes, numerically,

If for example the copper layers are 0.1 mil thick and the polyethylene
layers 0.05 mil thick, since we are assuming 6 = 2/3, then the attenuation
constant has increased by 10 per cent of its “flat”’ value at a frequency of
about 9.1 Mc-sec™.

We may also ask for the upper crossover frequency, above which the
Clogston cable will have a higher attenuation constant than a standard
air-filled coaxial of the same size. Such a crossover frequency must exist
because the dielectric loading of the Clogston cable (in our case @, =
6.78) introduces a factor ~/e, into the asymptotic expression for the
attenuation constant at extremely high frequencies when the stacks look
like solid metal walls; in addition there will be slight differences due to
the fact that the geometric proportions of the conventional and Clogston
cables are not exactly the same.

We assume, subject to a posteriori verification, that the upper cross-
over frequency lies between the critical frequencies f2 and f; , defined by
(178), for each stack. Then we have in effect infinitely deep stacks of
moderately thin laminae, whose surface resistances are equal and are
given by (182) to be

The attenuation constants of the conventional and Clogston cables are
obtained from (151) and (176) respectively, where for the conventional
coaxial we set nm = 7,. After a little arithmetic we find for the upper
crossover frequency in this particular case,

Thus if the copper layers are 0.1 mil thick, the upper crossover frequency
is about 280 Mc-sec™’, which turns out to lie well inside the interval
between the critical frequencies fz and f; for both stacks.

Comparing this result with the result at the end of Section IV, we see
that a 0.375-inch Clogston 1 cable with 0.1-mil copper conductors and’
the other specifications given by (155) is nominally better than a con-
ventional air-filled coaxial cable of the same size in the frequency range
from about 1 Mc-sec™ to 280 Me-sec’. We are still neglecting the effect
of failure to satisfy Clogston’s condition exactly, the effect of stack non-
uniformity, and dielectric losses. All of these factors will be present to a
greater or less degree in any physical embodiment of a Clogston cable,

In terms of k, the general expressions for T, K; , and Ke in a plane stack
of finite layers take a relatively simple form. We have

after a little rearrangement, where the only approximation that has
been made so far is to set my % m and m  o . Substituting (207) into
(86) and (87) gives

If k = 0, equations (208) and (209) evidently reduce to (158) and (159)
of the preceding section. For a stack of infinitesimally thin layers, the
constants IT, and K are given by equations (93) and (94) of Section
IIT, namely

Up to this point we have set no restrictions on the magnitude of k,
and we have not even assumed that k is necessarily real. Throughout
the rest of this section, however, we shall assume that k is a positive or
negative real number, as it must be if there is no dielectric or magnetic
dissipation.

In practice both the lamina thickness and the amount of dielectric
mismatch will be as small as it is feasible to make them. It will be useful,
therefore, to obtain approximate expressions for I, K,, and Ke under
the assumptions

If |k| « 1 we can neglect 2k compared to unity in the coefficient of
©*, but since we have made no assumptions as to the relative magni-
tudes of |@| and |k|, we cannot drop either the term in kO* or the
term in ©*. If we replace sh I by 3 in (214), we get

where we have taken the square root of the complex quantity by Dwight
58.2, and

The effective skin depth of a stack of moderately thin layers in the
presence of slight dielectric mismatch is, from (215),

An equation essentially equivalent to this was given by Clogston, in
somewhat different notation.” It is clear from (211) or (218) that if the
layers are infinitesimally thin, we have
A = 6/0|k i, (219)
and the effective skin depth in the stack is proportional to the skin
17 Reference 1, equation (III-42).

depth 6, in the conducting material at the operating frequency, although
if the mismatch parameter k is small, the proportionality constant mul-
tiplying 6; will be large. In the general case, the number of double layers
in one effective skin depth is

It is instructive to plot the effective skin depth of a given stack at a
fixed frequency as a function of dielectric mismatch. If

Ao = V(t + &)51/ti (222)
denotes the effective skin thickness when there is no mismatch, then the
relative skin thickness when the mismatch parameter is k is just

This ratio is plotted against k in Fig. 8, a universal curve being obtained
by measuring k in units of (4/6:)”. It is worth noting that when k =
(:/6:), the effective skin thickness is only 53 per cent of the skin
thickness with perfect dielectric match.

The surface impedance Zo(yo) of a laminated plane stack at any fre-
quency and with any amount of dielectric mismatch is given by equa-
tion (65),

Za(yo) sh mE + 3(Kie""? + Kee"?) ©
For a stack with infinitesimally thin layers and total thickness s, the
equation becomes

Saye) a Z(yo) sh Tes + K ch Tus’ (225)
where I, and K are given by (211) and (212). At zero frequency,
| 1
Di ae (226)

Zo(yo) = K coth Tys. (228)
If the stack is many effective skin depths thick, we have
Z(yo) = Ki, (229)
while if the individual layers are infinitesimally thin, |
Z(vo) = K, (230)

where K, and K are given by (209) and (211), respectively.
When Z(yo) is known, the attenuation and phase constants of the
parallel-plane Clogston 1 are given as usual by

' but the impedances of the cylindrical stacks are easy to compute only if
we can employ the parallel-plane approximation for each stack. To take

Fig. 8—Relative skin depth A/A» in a stack of finite layers versus dielectric
mismatch parameter k, measured in units of (t1/61)?.

curvature effects into account would require a considerable amount of
numerical calculation. Equation (98) of Section III provides an explicit
expression for the surface impedance of a cylindrical stack of infini-
tesimally thin layers in the presence of dielectric mismatch, in terms of
Bessel functions of complex argument; but if the layers are of finite
thickness we can at present do nothing better than multiply out the
matrices of the individual layers step by step.

The variation of the surface impedance of a laminated stack with
frequency over the full frequency range is not quite so simple in the
presence of dielectric mismatch as when Clogston’s condition is exactly
satisfied, but a somewhat analogous discussion may be given. As in the
preceding section, we consider a plane stack of n conducting layers
each of thickness ¢; , where nt; = 7',, and backed by an infinite-imped-
ance surface. When the mismatch parameter is k, the three critical
frequencies are:

In the range 0 S f S fe, the surface impedance of the stack is ap-
proximately constant, being given by

where Ky is given by (217) provided that k is small compared to unity.
Tor infinitesimally thin layers the upper critical frequency f; is infinite,
and we have for f = fo,

Zolyo) & | k- — isgn k)/gds
= (1 —tsenk)Vam|k| f/m, |
which is proportional to ~/f. If the layers are of finite thickness but
k = 0, we have the result obtained in the preceding section,

which is proportional to f up to the critical frequency fo . If neither the
mismatch parameter k nor the layer thickness tf; is zero, then the surface

impedance Zo(yo) cannot be represented by a simple power of f in the
range fe < f S f;. At frequencies above f;, if the layer thickness is
finite, the impedance is approximately that of a solid conductor, namely

Since in general the surface resistance depends upon the two param-
eters 4/6; and k, it is not possible to plot a single curve which shows the
variation of resistance with frequency under all possible conditions of
dielectric mismatch. However if we compare a matched stack of finite
layers with a similar mismatched stack, we see that the asymptotic
behavior of Zo(yo) is the same for both stacks at very low and very
high frequencies. A numerical study of the exact equation for Zo(yo)
shows that in the neighborhood of the critical frequency fe, the resist-
ance of the mismatched stack is higher than the resistance of the matched
stack. (The critical frequency fe as defined in (235) is a function of the
mismatch parameter k, but will be of the same order of magnitude for
a slightly mismatched stack as for a perfectly matched stack.) The
resistance of the mismatched stack exhibits relatively small fluctuations
above and below the resistance of the matched stack in the neighborhood
of the upper critical frequency fs , but this region is not of as much prac-
tical interest as the region near fo, where the stack resistance is defi-
nitely increased by the effect of dielectric mismatch.

An explicit expression for the rate at which the surface impedance of
a mismatched stack begins to depart from its de value as the frequency
is increased has been worked out only for the ideal case of infinitesimally
thin layers. For a plane stack of infinitesimal layers backed by an in-
finite-impedance surface, equation (228) gives, at moderately low fre-
quencies,

2 Bie cc
gly 36? 456+ :
from which the fractional changes in resistance and reactance are

The admissible value of | & |, if the fractional change in resistance is not
to exceed a specified value AR/Rp at a given top frequency fn, is

which is inversely proportional both to fm and to the square of the total
thickness of conducting material in the stack. If we express 7’; in mils,
fm in Mc-sec™’, and assume the conducting layers to be copper, we get

for the three values 4/6: = 0, é/6; = 0.1, and 4/6 = 0.2. For an elec-
trically thick solid conductor we have simply
Re gio, = Ts (247)

hence to get any benefit from the laminated stack we must have
Re g:6:K1 smaller than unity. Actually, if we meet Clogston’s condition by

Fig. 9—Normalized stack resistance Re gié:kK, versus dielectric mismatch
parameter k, for different values of t1/61.

raising the dielectric constant and thus lowering the impedance of the
main dielectric, then since the attenuation constant of the line is pro-
portional to the ratio of stack resistance to dielectric impedance, we
must have Re gi6:4, considerably smaller than unity to obtain a lower
attenuation with the Clogston line than with an ordinary air-filled line
having solid metal walls.

For a plane Clogston 1 line with stacks of equal thickness, the frac-
tional change in the attenuation constant with frequency is equal to —
the fractional change in the resistance of either stack, whether this
change arises from the effects of finite lamina thickness or from di-
electric mismatch or both. The fractional change in the attenuation con-
stant of a coaxial Clogston 1 depends not only on the change in resist-
ance of each stack, but also on the geometric proportions of the cable,
in the manner expressed by equation (200) of Section V.

The effect of dielectric mismatch on the overall attenuation versus
frequency characteristic of a Clogston cable is in general to reduce the
total frequency range (in Mc-sec™) over which the Clogston cable has
a smaller attenuation constant than a conventional air-filled coaxial
cable of the same size. To calculate the lower crossover frequency we
may ordinarily neglect finite lamina thickness effects and use equation
(241) for the stack impedances, while at the upper crossover frequency
the stack impedances are very nearly equal to K1, as given by (217).

It should be remembered that mismatch of the ye product of the main
dielectric will usually be accompanied by a change in the dielectric
impedance ~/ po /eé . Thus under certain conditions the lower crossover
frequency may even be reduced by choosing 6 slightly below the Clog-
ston value, inasmuch as the increase in dielectric impedance may more
than compensate for the increase in stack resistance at low frequencies;
but it appears that this will be paid for in a steeper slope of the attenu-
ation versus frequency curve and a consequent greater reduction of the
- upper crossover frequency.

It would be very useful to make a numerical study of the effects of
dielectric mismatch in Clogston cables having a variety of different pro-
portions; but in the present paper space limitations restrict us to a few
observations concerning orders of magnitude. For the cable which we
considered at the end of the preceding section, it turns out than an
increase or decrease of 1 per cent in the value of ¢) makes a change of
at most a very few per cent in either crossover frequency; with a matched
dielectric, we recall, these crossover frequencies were about 1 Mc-sec™
and about 280 Me-sec™ respectively. However if we had designed a
laminated cable with thicker stacks or thinner laminae or both, so as to
increase the theoretical factor of improvement over a conventional cable
in the working frequency range, we should have found that the tolerable
deviation of ¢ from Clogston’s value, instead of being of the order of
1 per cent, was more nearly of the order of 0.1 per cent or even smaller;
and the greater the improvement striven for, the more stringent the re-
quirement of accurate dielectric match.

Dielectric and magnetic dissipation in the main dielectric and in the
stacks can be taken into account by introducing complex dielectric con-
stants and permeabilities for the lossy materials. Thus we may write

2 = Wy — tue = we (1 — 7 tan f),
where in the most general case the loss tangents may all be different,
though it will be assumed that they are all small compared to unity, so
that the problem may be treated by first-order perturbation methods.

The average rate of energy dissipation per unit volume in a lossy di-
electric by a harmonically varying electric field of maximum amplitude
E is just 4we’ i”, since the imaginary part e” of the complex dielectric
_ constant corresponds to a conductivity g = we”. Similarly the average
rate of energy dissipation per unit volume in a lossy magnetic material
by a harmonically varying magnetic field of maximum amplitude H is
1wu”H’. The part of the attenuation constant which arises from di-
electric and magnetic dissipation is one-half the ratio of power dissipated
per unit length of line to total transmitted power, provided of course
that the attenuation per wavelength is small. Since the loss tangents
of the various materials are assumed small, we can use the fields found
for the lossless case to calculate the transmitted and dissipated power.

If the volume occupied by currents in the stacks is small compared -
to the volume of the main dielectric, so that we can neglect the power
flow in the stacks in the direction of wave propagation compared to the
power flow in the main dielectric, then the part of the attenuation
constant which is due to dielectric and magnetic dissipation is given by
equation (51) of Section II, namely

where X, is the vacuum wavelength and lor , €or are the real parts of the
relative permeability and relative dielectric constant of the main di-
electric. This equation will be derived from energy considerations pres-
ently. It should be noted that the part of the attenuation constant
given by (249) is directly proportional to frequency, provided that the
loss tangents are independent of frequency; but it is the same for both
plane and coaxial geometry and is independent of all the geometrical
factors which describe the size and the relative proportions of the line.
Equation (249) will probably be sufficiently accurate for all Clogston
1 lines having the proportions (stacks thin compared to main dielectric)
which we have considered in Part I. As an example wherein we also take
into account the power flow in the stacks, however, we shall treat a
parallel-plane line with infinitesimally thin laminae backed by high-
impedance walls. Then, according to equations (120) and (121) of
Section IV, the principal field components in the main dielectric are

the propagation factor ¢ ”*"*' being understood throughout. To take .
account of dielectric and magnetic dissipation in the stacks, we write

The average power Pp transmitted through the main dielectric is
obtained by integrating the real part of the z-component of the com-
plex Poynting vector 3E X H* over unit width of the line; thus

Similarly, the average power P; transmitted per unit width of either
stack is

The average power AP, dissipated in the main dielectric per unit length
and width of the line is

while the average power AP, dissipated per unit length and width of
either stack is

which reduces to (249) if we neglect the terms in s/b. The total attenua-
tion is the sum of the metal losses, given by equation (110), and the
dielectric and magnetic losses.

For a coaxial Clogston 1 cable with infinitesimally thin laminae and
high-impedance boundaries, the principal field components are given by
equations (126)—-(128) of Section IV. In the main dielectric we have

I(b? — p’)
2mp(b” — ps3)’
fa ft JOP)
me @’ Qap(b” — p2)*
The average power transmitted through the main dielectric is

while for the average power transmitted through the inner and outer
stacks it will be sufficient to replace the exact expressions by the follow-
ing simple approximations,

For the average power dissipated per unit length of line in the main
dielectric and the inner and outer stacks we have, respectively,

The part of the attenuation constant which is due to dielectric and
magnetic dissipation is therefore

We need scarcely point out that if the loss tangents are not small
compared to unity, it may be impossible to satisfy Clogston’s condition
(102) very closely with a real value of 6, and the resulting mismatch
may reduce:‘the depth of penetration and increase the metal losses in
the stacks. In practice, however, the loss tangents will be of the order
of 0.001 or even 0.0001, and matching the imaginary parts of woe) and
jé will be much less of a practical problem than matching the real parts.

Let pi and pe be the inner and outer radii of a cylindrical shell and
let the thickness t, given by

be less than p; . Then, following Schelkunoff,’ we may replace the Bessel
functions appearing in equation (68) of Section III by their Taylor ex-
pansions, namely

ex (A2)
Iulvps) = Tolees) = Yo "19 (ens,
ad of ! iat = (xt)” (n+1)
Kx(kpo) a Ko(xpe) = a nl Ko (kp1).
It follows that ,
oo i n
Ko(kpi)I 1(kpe) ai Kx(kpo) Lo(xp1) eae a a Bryilkpi),
0o i n
Kaeor)Iu(ees) — Kaleoa)Io(ses) = — 2  Bylups),
ca : (A3)
(xé)”
Kilxpr)Likp2) — Ki(kp2)Li(kp1) = a ae Ansi(kpi),
Ku(xp1) Lo(kp2) Bo Ko(kp2)L1(xp1) = dX uo" A, (xpi),
where
A,(x) = Ip(a)K§” (x) — Ko(x)I8 (2), (M4)

The quantities A,(#) and B,(x) turn out to be finite polynomials in
1/z, the general expressions for the coefficients having been derived in
a rather inaccessible monograph by Pleijel.” When x is large, however,
the leading terms are quite simple. From Pleijel’s analysis, or directly
by substituting the asymptotic series for Jo(x) and Ko(x) into (A4), we
find

If we substitute these approximations into the first of equations
(A38), we obtain

2H. Pleijel, Berdkning af Motstand och Sjdalfinduktion, K. L. Beckmans Bok-
tryckeri, Stockholm, 1906.

The other three equations may be treated similarly. Doing so, and
remembering that

p2o/p. = 1 + t/pi, (A7)
we obtain the results which were quoted in Section . namely
Kp2( Kol 12 + Kyl) & 1+ st ch xt — oa sh Kt,
xpx(Kulo: — Koeln) | 1+ — x sh xt,

Note: Rationalized MKS units are employed throughout. The sub-
scripts 0, 1, 2 applied to symbols representing material constants, such
as €, uw, g, , and 7, have the significance that 0 refers to the main dielec-
tric in a Clogston line, while 1 refers to the conducting layers and 2 re-
fers to the insulating layers in the stacks. Bars denote average values.
Subscripts not included in the present table are explained in the context
where they are used.

a: Distance between outer sheaths of plane Clogston line.
Radius of inner core of coaxial Clogston line.

Characteristic or iterative impedances of laminated stack
(introduced in Section ITI).

A parameter related to dielectric mismatch in a Clogston
1 line (Section VI).

Rectangular coordinate in the direction of magnetic field
ina plane Clogston line.

Rectangular coordinate in the direction normal to the
stacks in a plane Clogston line.

Surface impedance of a plane or cylindrical boundary;
ratio of tangential components of the electric and mag-
netic fields (subscripts explained in context).

I'/(i + t); average propagation constant per unit dis-
tance normal to laminated stack.

Effective skin depth; the depth at which the current den-
sity in an infinite plane stack has fallen to 1/e of its value
at the surface. A small change in a quantity.

tan” '(u”/u’); phase angle of complex permeability.
iwp/(g + twe); intrinsic impedance of medium.

n(1 — y/o)’; characteristic impedance looking in the
y- or p-direction in a homogeneous medium.

(1 + 1)t/6: ; a parameter related to the electrical thick-
ness of a.conducting layer.

(o*? — 7’); transverse propagation constant in the y- or
p-direction in a homogeneous medium.

A parameter related to the propagation constant in a
Clogston 2 (Section XII).

y/a + 4; normalized coordinate transverse to a plane
Clogston 2 line (Section XII).

Angular coordinate in cylindrical system. Phase angle,
tan’ ‘(e’/e’), of complex dielectric constant.

Bessel (Neumann) functions of the second kind.
Modified Bessel functions of the first kind. ~
Modified Bessel functions of the second kind.

is the spectrum. The spectral density (power per unit bandwidth) varies
inversely as the frequency, according to the relation

where the exponent lies between 1 and 1.5 with an average about 1.2.
This type of spectrum will be referred to as a 1/f spectrum. Measure-
ments of the spectra of silicon point contact diodes have been reported
by P. H. Miller’ for the frequency range 20 cycles to 300 kilocycles.
Spectra of point contact transistors measured by the author have been
reported elsewhere’’ * for the range 20 to 15,000 cycles. Typical spectra
for p-n junction type diodes and transistors are shown in Fig. 1. Almost
without exception, our measurements and those reported in the litera-
ture have shown the 1/f spectrum over most of the frequency range
covered. There is some evidence from the related fields of flicker noise
and carbon microphone noise that the 1/f spectrum may extend to fre-
quencies well below 0.1 cycle per second. Some departures from this
type of spectrum have been noted in the neighborhood of 100 ke, as
shown in the curves.

Fig. 1—The spectrum of noise in n-p-n transistors varies inversely with fre-
quency.

Fig. 2—The short-circuit noise current from a point contact or junction diode
generally increases with de bias current.

i 0.2 a 0.6 0.8 1.0 40
COLLECTOR BIAS nN voLTs
Fig. 3—The noise figure of an n-p-n transistor depends in a fairly simple way
on emitter current and collector voltage.

ture on noise behavior. Such experiments have been rather unsatisfac-
tory because the changes in impedance and gain characteristics as a
function of frequency are of the same order as the changes in noise prop-
erties. This makes the interpretation ambiguous. By and large, such
experiments suggest that changes in noise with temperature are rather
small, perhaps of the order of the change in absolute temperature, and
not at all like the exponential changes associated with a diffusion process.
This observation does not necessarily rule out a diffusion-like noise
process; it might indicate merely that we are not looking at the right
part of the spectrum to observe exponential changes with temperature.

It has been observed. that in many semiconductor structures the
noise voltage is approximately proportional to the de bias current.
This relation suggests that the noise is the result of fluctuations of the
conductivity of the material, which modulate the bias current and pro-
duce a fluctuating voltage across the specimen. Such fluctuations in
conductivity could result from variations in concentration of the mi-
nority carrier (holes in n-type material, electrons in p-type). The mag-
nitude of the observed noise and the type of spectrum seem to demand
that the fluctuation be coarse-grained in time to a much greater extent
than could be accounted for by random statistical fluctuations of carrier
density. Experiments of Haynes” on lifetime and transit of injected car- -
riers in rods of germanium have occasionally indicated finite sources of
minority carriers in the material. Our hypothesis is that such sources
of carriers are rather generally distributed over the material (although
mostly too small to be noticed in experiments of the Haynes type), and
that their activity is being modified at a slow rate by some unspecified
local influence in a suitable way to agree with the observed noise spec-
trum. ;

Samples, referred to as “bridges”, have been cut from thin slabs of
single crystal germanium, by a technique devised by. W. L. Bond,”
often of a form shown in Fig. 4. Side arms for both the current and
the noise measuring electrodes have been found necessary to avoid
spurious noise at the electrodes. A large inductance in the bias circuit
greatly reduces the effect of any noise voltage generated at the bias
electrodes. The spurious noise power from this source is seldom more
than a few per cent of that being measured. It should be noted that the
contact area for the noise measuring electrodes should not be on a por-
tion of the specimen carrying bias current, otherwise spurious noise may
be generated at these electrodes. Typical dimensions for the straight
central filmanent of the bridge are 0.05 x 0.05 x 0.7 cm. The side arms
have sandblasted surfaces to suppress holes or electrons injected at the
electrodes. The central portion may be etched, sandblasted, or other-
wise treated at will. The enlarged circular areas are rhodium plated to
provide good contacts to each side arm.

Measurements of the noise spectrum in such bridges with several dif-
ferent etching treatments and with sandblasted surfaces are charac-
terized by the 1/f spectrum over a wide frequency range.* Fairly ex-
tensive measurements have been made in the audio frequency range,
and a few covering the range from 20 cycles to 1 megacycle. A typical
spectrum is shown in Fig. 5.

The current dependence of the noise is shown in Fig. 6 for a number
of samples, mostly n-type, one p-type, and with various resistivities.
The outstanding feature is that noise voltage always increases with de
bias voltage. In many cases there is direct proportionality at the lower
bias values, increasing to a square law at higher biases. There are some

LENGTH
ABOUT 7MM
Fig. 4—Filament with side arms cut out of a single crystal of germanium.

* Departures from the 1/f spectrum at frequencies of the order of 100 kilo-
cycles and above were first discovered by G. B. Herzog and A. Van der Ziel.
See Reference 13.

Fig. 5—Typical spectra of noise in single crystal filaments carrying a de cur-
rent.

exceptions to this trend. Also, there are large variations in the magni-
tude of the noise. An average unit shows a noise voltage about three
times Johnson noise at a bias of-10 volts per centimeter.

The noise behavior at reduced temperatures has been investigated.
Results on ‘three different bridges are shown in Fig. 7. The open circuit
noise voltage is shown as a function of temperature for constant bias
voltage. Although the curves show rather large irregularities, there
seems to be no general trend for noise to decrease with decreasing tem-
perature over the range covered, from —200°C to room temperature.

The surface treatment applied to a bridge may affect the noise very
substantially. A sandblasted surface usually gives the lowest noise.
_ Etching the surface may raise the noise voltage by a factor of ten or
more, though the de resistance changes only a few per cent. The tech-
nique of washing and drying the surface may have an important effect

BIAS FIELD IN VouTs PER CENTIMETER.
Fig. 6—Variation of noise with de bias in single crystal filaments.

on the noise. Some of these processes also affect the lifetime of carriers
in the bridge to a large extent. However, there seems to be no direct
and simple relation between the two effects, since treatments have been
found which change the noise by a large factor with almost no effect on
lifetime, and vice versa.

Fig. 8 shows measurements of noise voltage on several dozen bridges
at a uniform bias of 10 volts per centimeter, all having sandblasted sur-
faces, mostly of n-type but a few of p-type germanium, and with widely
different values of resistivity, produced by varying impurity concentra-
tions. There is considerable scatter in the results, but there is a fairly
obvious tendency for noise voltage to increase in proportion to resis-
tivity. Since Johnson noise also increases in proportion to resistivity in
a structure of fixed dimensions, the conclusion is that with constant bias
voltage the ratio of current induced noise to Johnson noise tends to be

independent of the resistivity of the material. From the data it also
appears that there is no consistent difference between n- and p-type
material. .

Noise does not appear to depend on orientation of the filament with
respect to the crystal axes. Filaments orientated along the 100, 110, and
111 directions and rotated in several ways about these directions showed

An important role for the minority carrier in the noise mechanism
was first clearly indicated in experiments on the effect of a magnetic
field on noise in germanium filaments. It has been found experimentally
that the noise in a single crystal filament may change by a substantial
factor when the filament is subjected to a steady transverse magnetic
field. The following discussion will show that this behavior is in har-
mony with the hypothesis of noisy injection of minority carriers, as set
forth in a preceding section.*

The physical picture on which this treatment is based involves the
random injection of holes into an n-type filament by hole sources which
may be either in the interior or on the surface of the filament.t It is
assumed that the spectrum of the noise arises from the fluctuating na-
ture of the noise source. The effect which any source has will depend
upon the lifetime of the holes which it emits. If these holes remain in
the filament for a long time, they will produce more noise than if they
remain in the filament for a short time. We shall be concerned with
the effect of magnetic fields upon these lengths of time and shall not
deal in this paper with the fluctuations of the noise sources themselves.
If a transverse magnetic field is applied to an n-type germanium fila-
ment, a Hall effect voltage is set up and the holes will be deflected to-
wards one surface of the filament. Since recombination takes place prin-
cipally at the surfaces, this may cause a substantial change in the lifetime
of the holes. In order to determine the effect of the magnetic field on
the noise we proceed along the following lines.

(a) We assume that the observed noise is due to fluctuations in the
conductivity of the filament produced by fluctuations in the hole con-
centration. Since these fluctuations are small, we may take the change
in conducitivity to be proportional to the change in average hole den-

* The following semi-quantitative theory of the dependence of noise on mag-
netic field is taken with some modification from unpublished work of W. Shock-
ley and H. Suhl, on the basis of which the calculations leading to the curves of
Figs. 10 and 11 were carried out. It is hoped that this work may be published in
the near future.

+ To simplify the terminology, the discussion is based on n-type material with
holes as minority carrier. An exactly similar argument could be made for p-type
material with electrons as the minority carrier. There is some experimental evi-

dence of the similarity of behavior of n- and p-type germanium, though most of
the experimental work has been done with n-type.

cussion which follows, and also in the calculation of the curves of Figs.
10 and 11.

* It should be pointed out that a consequence of the hole injection theory of
noise in a filament is that marked frequency dispersion should occur when the
frequency being studied is high enough so that a period is short compared to the
lifetime of holes in the filament. However, we shall neglect this important and
interesting aspect of the problem.

(a) (b)
Fig. 9—Excess hole density across the thickness dimension, (a) with no magnetic
field, (b) with moderate magnetic field.

If we suppose a steady hole current Jo emitted from the left-hand
surface of the filament, then a relatively high concentration 7; of holes
will appear directly in front of the surface. Some of these holes will re-
combine upon the surface, the rate J; being given by

where S is the recombination constant for the surface. The balance of
the holes will diffuse through the filament to recombine upon the right
surface at a rate

and we note that J; + Jz. = J. Because of the high recombination rate,
2 will be very small; hence, J2 will be much smaller than J; . In the ab-
sence of a magnetic field the gradient is uniform, and the concentrations
will be linear, as shown in part (a) of the figure. An identical argument

The constant may be derived by noting that k7'/q is 1/40 volt at room
temperature, and that the effective transverse field, Hy , may be ex-
pressed as follows. (See Reference 14, Section 8.8.)

where 6 = Hall angle .
Hall mobility for electrons (2800 em’/volt-sec)
Uy» = Hall mobility for holes (1500 em?/volt-sec).

The other dimensionless parameter is proportional to the rate of surface
recombination, and is defined as the ratio of the surface recombination
velocity to the diffusion velocity from the center:

The numerical constant is given for holes at room temperature. The
noise changes are expressed in decibles, that is, ten times the common
logarithm of the ratio of noise powers with and without the magnetic
field.

A second case is that in which generation and recombination are on
the surfaces, but the two surfaces have unequal absorption properties.
It might be expected that rather large increases in noise would result
when the magnetic field was poled to pull holes away from the surface
with high absorption properties, and this turns out to be the case when
the calculations are carried out. The results are shown in Fig. 11 for a

SURFACE GENERATION
a~—— VOLUME GENERATION
yp SURFACE RECOMBINATION
CONSTANT

-16 -14 -12 -10 -8 -6 -4 -2 0 2 4 6 8.10 12 14 16 18 20
MAGNETIC FIELD PARAMETER, ®
Fig. 11—Calculated magnetic effect for dissimilar surfaces. Curve 1 of each
pair is for the contribution from the surface having the lower recombination
constant.

-4 Oo 4 8
MAGNETIC FIELD PARAMETER, 6
Fig. 12—Experimental magnetic effect for dissimilar surfaces.

case where the two recombination parameters are 0.1 and 10, and.for
a second case where the parameters are 0.5 and 50. In this figure, sepa-
rate curves have been shown for noise due to holes generated on each
of the two surfaces. The total noise would be gotten by adding the noise
powers represented by the two curves after appropriate weighting for
the contributions of the two surfaces. At present we do not see any way
of determining the weighting factor.

A third case is that in which it is assumed: that the noisy generation
of holes is uniform throughout the body of the filament, but that re-
combination takes place on the surfaces only. These assumptions seem
at first sight to be in contradiction to the statistical mechanical prin-
ciple of detailed balancing, which states that under equilibrium condi-
tions all processes occur with equal frequency in the forward and reverse
directions. Thus it would seem that if holes are generated in the interior,
we must consider recombination in the interior also. Actually this is
not necessary under the non-equilibrium conditions which prevail
during noise measurements. There is no necessity for the noise generated
by a source and a sink for holes to be simply related to the strength of.
this source. Thus we may suppose there are relatively weak sources and
sinks for holes in the interior, but that the hole absorption and genera-
tion of the sources is very noisy compared to the recombination and
generation processes on the surfaces. If this is the state of affairs, most

where EF is the applied field in volts per centimeter and uw is the drift
mobility of holes. Hence, by varying the biasing voltage a large range
of life path values can be obtained.

The correlation is measured by carrying the noise voltages through
separate amplifying channels having identical pass bands extending

from 800 to 1300 cycles per second. A switching arrangement makes it
possible to apply either of the output voltages or their sum or difference
to a rectifier-meter combination. From the readings of the meter the
correlation can be computed according to the relation ‘:

VY, and V2 are rms values of the individual noise voltages, and S and D
are the rms values of their sum and difference. The equivalence to
expression (5) can be seen by noting that

Results of correlation measurements on three bridges are shown in
Figs. 14-16. In each case the calculated curve is shown for reference.
The values of ¢ were calculated from decay measurements on optically
injected holes, as described by J. R. Haynes,” using a value for mobility
of 1700 cm’/volt-sec. In Fig. 14 the agreement with the theoretical model
is very good. The scatter in the points is due to fluctuations in the
noise, which are quite large in the band used for these measurements.
In Fig. 15 the agreement could be made quite good with a lateral shift

Fig. 14—Noise correlation. The solid curve is calculated, the points experi-
mental.

Fig. 15—Noise correlation. The dotted curve includes allowance for losses at
the side arms.

by a factor of two. In Fig. 16 the form of the experimental curve seems
different from that calculated. In particular, the slope is steeper, and
the curve tends to level off at a correlation of about 0.8. It seems pos-
sible to explain the discrepancies between the experimental data and
the calculations on the basis of two considerations which were not in-
cluded in the model. (a) The pair of side arms separating the two seg-
ments of the filament serve to drain off some holes which would other-
wise contribute to the correlation. The dashed curve in Fig. 15 shows
the calculated effect, on the assumption that the absorption in the side
arms is equivalent to an extra section of filament equal in length to half
a segment. The actual distance across the side arms is only 20 per cent
of a segment, but it is not hard to believe that the decay rate in this
region might increase by a factor of two or three due to the reduced elec-
tric field and loss of holes down the side arms. (b) The model assumed a
uniform distribution of noise sources along the filament. There is ex-
perimental evidence that the distribution may be quite spotty. This
can have a substantial effect on the form of the correlation curve. For
example, the dashed curve in Fig. 16 shows the curve calculated for
noise sources lumped at the mid-point of each segment. Other assumed
positions might shift the curve considerably along the horizontal axis.

Fig. 16—Noise correlation. The dotted curve is calculated for lumped noise
sources.

In view of these considerations there seems to be very satisfactory agree-
ment between the experimental results and the model.

Another type of experiment involves noise measurements at frequen-
cies high enough so that the transit time of a hole across a segment is
an appreciable fraction of a cycle. In this case the correlation between
noise voltages from adjacent segments can be improved by putting a
time delay in one channel of the measuring circuit. Measurements were
made by taking the noise voltages from the two segments through sepa-
rate amplifying channels having identical pass bands extending from 17
to 24 kilocyles. The outputs of the two channels were put on the vertical
and horizontal plates of a cathode ray oscilloscope, forming a sort of
Lissajous pattern. The patterns differ from those obtained with sinus-
oidal voltages in that the elliptical figures are filled in solid, due to the
continual variation in amplitude of the noise. A phase shifting device is
included in one channel, and as the phase is shifted to give optimum
correlation, the elliptical pattern narrows down and approaches a, line
inclined at 45°. For a quadrature phase shift, the pattern becomes cir-
cular, and in practice this setting can be determined more precisely
than the in-phase setting, largely because the background noise in the
circuit is less troublesome. With the phase shift for optimum correlation
determined, the delay at the center of the pass band is easily calculated,

Data for a bridge of n-type germanium of resistivity about 20 ohm-
cm are given in Table I. The transit distance, L, after a small correction

for reduced field across the side arm, was taken as 0.305 cm. As noted
in the table, the bridge temperature rose somewhat at the higher bias
values, and the assumed values of mobility have been modified accord-
ing to the inverse three-halves power of the absolute temperature. The
delay required for optimum correlation is shown in the second column
of the table, and the calculated transit time between segments in the
last column. It is seen that the two are in reasonably good agreement,
especially at low fields. When the direction of the field is reversed, an
equal delay is required, but in the opposite channel of the measuring
circuit, as would be expected. Here, again, we have experimental evi-
dence supporting the noisy hole injection hypothesis. The cause of the
discrepancy shown in the table at higher fields is not understood. It is
possible that trapping phenomena increase the transit time over that
calculated from the mobility. There is some evidence for this sort of
behavior in lifetime experiments, but to date there does not seem to be
enough information for any estimate of magnitude of such an effect.

Suppose that a source of holes located at a point 2» in a filament pro-
duces a fluctuating current of holes of rms value J; in a specified fre-
quency band. The hole current is swept down the filament by a field #
and is assumed to decay exponentially according to the relation

where the life path £ may be expressed in terms of drift velocity v, hole
mobility yw, and lifetime 7
Lor = pr.

Assuming that the frequency of measurement is low enough to justify
neglecting the hole transit time, the noise voltage due to holes from a
single source is proportional to the number of holes present in the seg-
ment. This is obtained by integrating (1) over an appropriate range

Under the assumption that the sources are statistically independent,
the total voltage squared is obtained by integrating the square of (2)
over all the sources.

| re an
= K,{1 —F Tze a,
Similarly, the cross product of voltages in two segments, extending say
from 0 to L and L to 2, is

which is the desired relation, from which the solid curves of Figs. 14-16
were calculated.

Relays are produced by a large number of manufacturers in this country.
When we survey their product, we find that there are many kinds and vari-
ettes. They differ widely as to their size, shapes and configurations. Many
of these differences are dictated by the requirements of the task they must
perform and by the environments under which they must work. Other differ-
ences are brought about from considerations of cost and by the design and
fabrication techniques the particular manufacturer employs. However, all
relays have a common. objective. For whatever use they are employed, it 1s
highly desirable that they be reliable. They are expected to function each time
the, are called upon without failure and over the expected life of the equip-
ments in which they are used. This paper deals with the more important
design factors which all relays have in common that greatly influence their
reliability of performance. Contact spring pile-up stability and the impor-
tance of strength of screws, insulating materials with low cold flow and mois-
ture absorption, and manufacturing procedures and controls to achieve this
end are discussed. Coil construction so as to minimize the occurrence of open
windings due to corrosion of the wire and breakage of the lead-out wires 1s
dwelt upon. Contact reliability and how it 1s affected by the material used,
its size and shape, the method of actuation, the presence of contaminating
vapors, and single versus twin contacts are discussed. The degree by which
magnetic materials change their magnetic properties with age and treatments
for alleviating this effect are described. The importance of adequate structural
design so that the relay will be rugged and remain stable so that its perform-
ance 1s substantially unaffected by wear, shock and vibration 1s stressed.
Methods of test to determine how well the relay meets these objectives are
described.

Although a relay is conceptually a simple device, the wide range of
conditions under which relays are required to operate, the many different
characteristics they must have, and the complete dependence placed

cycles of alternate humid and dry environments over the years. Humid
conditions exist during the summer season, followed by a dry exposure
during the winter months when the central offices are heated. Experience
has shown that a relay exposed for six days to 90 per cent relative hu-
midity at 85°F will be comparable to those observed in service in humid
localities. Although the six day exposure is admittedly an accelerated
test, the dimensional changes produced are approximately the same as
those caused by the accumulating effect of fluctuating humidity during
the entire season. Similarly, a period of six days exposure to 120°F pro-
duces the same effect of drying as that which occurs during the heating
season.

By making careful measurements on an adjusted relay of such impor-
tant parameters as operate current, releasing current, contact separation,
contact spring tension, armature back tension, stud gaps, etc., and then
subjecting the relay to repeated cycles of humid and dry conditions,
repeating the measurements after each exposure and noting the changes,
a good appraisal of the relay can be made. A repetition of the test will
reproduce the same pattern of results as the first cycle unless permanent
deformation of the materials in the relay has taken place.

COIL COVER COIL COVER
WAXED_VARNISHED KRAFT PAPER INTERLEAVING
INTERWINDING INTERWINDING

DAYS ON TEST DAYS ON TEST
AT 90% RELATIVE HUMIDITY AND 85°F AT 90% RELATIVE HUMIDITY AND 85°F

CELLULOSE ACETATE INSULATED SPOOL-
WOUND CGIL VINCELLATATE MUSLIN
COVER AND RED ROPE PAPER WASHERS
IN CONTACT WITH WINDING WIRE (NOT
IMPREGNATED).

ITA CELLULOSE ACETATE FILLED COIL
HAVING LEAD WIRES INSULATED WITH
COTTON BRAID,(NOT IMPREGNATED).

DURING EXPOSURE TO 95% RELATIVE HUMIDITY
+100 VOLTS DC WAS APPLIED TO THE WINDING
WITH NEGATIVE POTENTIAL ON CORE.

ae 4—Corrosion comparison of impregnated and non-impregnated relay
coils.

winding washers. Likewise, one group was impregnated while the other
was not. Fifth and sixth groups of coils having cellulose acetate insulation
throughout with and without impregnation were exposed and there were
no failures at the end of the test. This shows that the corrosive effects
of impure materials can be retarded, but not overcome by resorting to
impregnation. In general, impregnation of relay coils is not desirable
because of the risk of contaminating the vital working surfaces of the
relay.

In the normal operation of a relay when the circuit through its winding
is opened, a transient voltage, which may reach hundreds of volts is
generated across the winding terminals by the collapsing flux. If the
insulation between the lead-out wires or between the lead-out wires and
the end turns of the winding is not adequate, electrical breakdown causes
arcing and repeated operation of the relay may cause ultimate disintegra-
tion of the wire and consequent failure of the relay. It is important, there-
fore, to design the coil so that lead-out wires under all conditions are
properly spaced, and to provide adequate insulation between those por-
tions of the winding where high voltages can exist. A test has been de-

Fig. 5—Relay coil employing a motion limiting washer to prevent lead breakage.

Since the opening and closing of contacts are the prime objectives of
a relay, it is extremely important that the contacts themselves are made
reliable. To realize these objectives, several factors should be taken into
consideration. First is the contact material. While much could be said
regarding the behavior of contact metals, space does not permit more
than a brief treatment. The contact should maintain reasonably low re-
sistance and under the environment in which it is used, be able to with-
stand the erosion. Electrical resistivity of most metals is low enough to
be satisfactory from the resistance standpoint, but unfortunately, most
of them develop tarnish or corrosion films when exposed to the atmos-
phere, thus increasing the contact resistance and rendering them unsuit-
able for a contact. These metals are sometimes referred to as ‘‘base”’
metals, and include aluminum, brass, bronze, copper, chromium, nickel
and stainless steel. There is a much smaller group of metals known as
“noble” or precious metals, such as platinum, palladium, gold and irid-
jum. These are relatively free from the tendency to tarnish and will
maintain low contact resistance. Alloys of these metals and certain alloys
in which silver is included are widely used in the telephone plant. Pure
silver is also used and is attractive because of its low cost; however, it
has a tendency to form high resistance tarnish films and therefore has
limitations in its use. It is employed in signaling circuits where the con-
tact makes or breaks current. Its contact resistance remains low because
the films that form on the silver are broken down or destroyed by the
are. It is not employed in circuits carrying voice currents on account of
its tendency to introduce noise.

Enough metal must be provided to give satisfactory life. Each time a
contact makes and breaks an electrical circuit, a small part of the metal
may be lost, so that life may be considered roughly proportional to the
volume of metal available for erosion. The pair of contacts must have
sufficient height to provide enough contact spring clearance to allow for
spring adjustment and to insure that the springs will not touch during
the normal operation of the relay. At least one contact of a pair must
be large enough, that is, present a sufficiently large target area, to insure
full registration of the contacts with normal manufacturing variations
of the position of the contacts on the springs and with the variations in
alignment of the springs during assembly.’

Fig. 6—Two methods of contact spring actuation and their influence on contact
locking.

operation is ‘‘open’”’ contacts due to small insulating particles present
in the atmosphere becoming trapped between the contacts. This causes
high resistance or open circuit and consequent circuit failure. Many at-
tempts have been and are being made to reduce “open” contact troubles.
Examples are filtering the air supply to the central office, enclosing the
relay equipments in closed cabinets, pressurizing the enclosing cabinets,
covering smaller groups of contacts by independent covers, employing
twin contacts rather than single contacts, and enclosing the relay or its
contacts in a hermetically sealed chamber. Even going to the extreme
of completely isolating the relay from its surroundings is not a complete
answer. There is always the possibility of failure by wear particles gen-
erated within the enclosure by the relay actuation.

The most widely employed method to reduce dust failures in the tele-
phone plant is the use of twin contacts in combination with some of the
above mentioned types of enclosures. If the incidence of dust failures fol-
lowed the laws of probability, then elementary considerations would lead
us to predict that if single contacts failed at the rate of once in 1,000 oper-
ations, the simultaneous failure of the two such contacts comprising the
twin would be once in 1,000,000 operations. This is the so-called “square”’
law.. However there are a number of reasons why this is not realized
in practice and why the figure of merit for the twin contact is very much
less than that indicated by the ‘square’ law. In the first place, when
foreign matter becomes lodged on a contact, it seldom falls out on the
first subsequent operation, but will require a number of operations before
it cleans itself; in fact, it may remain inoperative indefinitely. When this
happens to one of the twin contacts, during this period of time, the twin
contact is no better than the single contact. In practice, twin contacts
are generally used with the same total force as the single contact, being
nominally divided equally between the two contacts. This reduction in
force per contact on the twin contacts is of considerable importance in
reducing its effectiveness. Relay designs employing twin contacts that
have been used in the past do not have complete mechanical inde-
pendence of the two members to which the contacts are attached.
Foreign material or protrusions under one contact can adversely influence
the performance of its mate. In a new design of relay which is about to go
into production, the design criterion that twin contacts to be most
effective should be completely and mutually independent has been met.
Laboratory tests and field experience obtained to date show a marked
improvement over the former designs in regard to the incidence of open
contact failures:

OPERATE AMPERE TURNS
Fig. 8—Improvement in relay performance by hydrogen anneal.

ARMATURE MOTION IN INCHES
Fig. 9—Mechanical load of a relay and its friction component.

the displacement and hence the friction is large, aside from the fact
that it is indicative of a rapid rate of wear, the relay would be unstable.

While the wear of the relay parts can be minimized by good design,
it cannot be eliminated entirely, especially for relays required to operate
a very large number of times during their life. For telephone relays the
design objective is for a 40-year life. The effects of wear on performance
to a great extent can ofttimes be counteracted by ingenious design.
Fig. 10 is an illustration of such a case.> The diagram on the left shows
a moving system of a relay in which the contact springs are stud ac-
tuated. The moving springs are tensioned toward the armature and exert
a force tending to open the contacts. When the armature operates, the
stud presses the moving springs into engagement with the stationary
springs. There is no contact force when engagement is first made and
further flexing of the spring is necessary to build up the contact force
to the desired value when the armature reaches its fully operated posi-
tion. As the contacts and studs wear, it is apparent that the contact
force and consequently the load on the armature decreases rapidly. The
stud wear becomes cumulative in its effect on the outside pair of springs
as more springs are added to the pile-up.

The diagram to the right shows a moving system of a relay using
what is called “lift-off” card actuation. The moving springs are ten-

Fig. 10—Two moving systems of relays in relation to the effects of wear on
their performance.

Fig. 13—Comparison of flat and embossed pole surfaces and their magnetic
closed circuit reluctance with misalignment.

done with the ordinary quick-to-release relays, for slow release relays
the armature is allowed to contact the core, finish to finish. When plane
flat pole face surfaces are provided, it is expensive and difficult to insure
in commercial practice that precise and uniform alignment of the pole
face surfaces will obtain. Variations in the alignment of these two sur-
faces will cause variations in the closed magnetic circuit reluctance and
consequently on the release time of the relay.

In Fig. 12 is shown a design where the necessity for holding the align-
ment of core and armature so precisely is not so great.® A spherical surface
of rather large radius is embossed on the front end of the armature, so
that with commercial variations in alignment, the armature always pre-
sents a point on the surface of a sphere for contacting the flat surface of
the core. Similarly, the legs of the armature where they pivot on the
front ends of the hinge bracket are likewise embossed. The results of
the effects of these structural differences on the closed circuit re-
luctance are shown in Fig. 13 for a design with flat surfaces and one
with embossed surfaces. While it is true that with perfect alignment the
relay with flat surfaces will give longer release times, it is apparent that
as variations in alignment occur from time to time and from relay to
relay, it will have larger variations in performance than the relay with
the embossed surfaces. This is a feature which has proven of great value
in the manufacture of slow release relays of reasonable time precision.

iif Oa ponueicet “Cellulose Acetate Filled Coils,’ Bell Labs. Record, 29, p. 514,
ov., 1951.
2. W. C. Slauson, ‘‘Improved U, UA and Y Type Relays,’ Bell Labs. Record, 29,
p. 466, Oct., 1951.

For measuring impedance and admittance parameters, that is 2, L,
C and G, suitable ac bridges, ordinarily simply designated as impedance
bridges, have long held a high place in the Bell System because of their
inherent reliability and precision, and their ability to cover a wide
range of values. The development of many of the original bridges’ ” *' *
for frequencies above the audio range stemmed from the needs of the
earlier carrier systems. With this development came also analysis of
shielding technique,’ standardization of capacitance,’ and a syste-
matic classification of bridge methods® by J. G. Ferguson in 1933, in
which bridges were grouped into two major types designated as ratio-
arm and product-arm, respectively. Following this classification, com-
bined impedance and admittance bridges were developed,” *° utiliz-
ing a single set of bridge standards for both kinds of parameters by
changing the configuration of the bridge network. There have also
been special purpose bridges’ ”'*'“ for use at audio and the lower
carrier frequencies. More recently, coaxial impedance standards” having
values calculable from physical dimensions have been developed.

Bridges for frequencies above one-half megacycle were used in the
Bell System as early as 1919,’° but relatively few bridges were built
until the mid 1930’s when new carrier systems required bridges in the

for their best performance, even though both bridges are useable at
higher frequencies. It will be observed that while there is some over-
lapping of the three ranges, all three methods are necessary to obtain
the impedance coverage shown. It should be emphasized that all the
ranges shown cover both capacitive and inductive reactances. In the
case of the admittance and series-reactance bridges, inductive imped-
ances are measured by using a resonating capacitor, in parallel or
series, respectively, with the apparatus being measured. In the Maxwell
inductance bridge, capacitive impedances are measured by using a
fixed resonating inductor in series with the impedance under test. A
complete accuracy statement for these bridges is necessarily complex,
but in general accuracies of +0.25 per cent for the major component

Fig. 2—One-megacycle Maxwell inductance bridge, shown schematically in
Fig. lc, designed for relay-rack mounting.

Fig. 3—Reactance/frequency chart showing the measurement range of the
bridges shown in Fig. 1.

frequency of twenty megacycles was decided upon as a design objective
with a basic accuracy of +0.5 per cent for the major component. The
immediate need was for a general-purpose bridge, but it was expected
that special-purpose bridges having better accuracy would be required
later.

Fig. 4—Reactance/frequency chart applying to the general-purpose bridge
shown in Fig. 5.

The successful use of a center-tapped transformer for ratio arms in a
465-KKC direct capacitance bridge’ indicated that the resistance ratio
arms RI, r2 of Fig. 1 might be omitted if a suitable transformer could be
developed for higher frequencies. The transformer group of the Labora-
tories succeeded in producing a transformer with a deviation from unity
ratio of less than 0.1 per cent over a frequency range from 0.5 to 20
megacycles. This was made possible by precise location of the windings
in fine milled grooves in the form of reversed helices, cut on a longitudin-
ally-split brass cylinder for the inner winding, and ona surrounding phenol

Fig. 5—Schematic of the 20-megacycle general-purpose bridge showing both
the series (impedance) and parallel (admittance) bridge circuits combined in a
single unit.

fibre cylinder for the bifilar outer winding which serves as the bridge
ratio arms. Electrostatic shielding limits the direct capacitance be-
tween primary and secondary to less than 0.01 uyf. The core material is
compressed powdered molybdenum permalloy. This transformer was
the nucleus around which the general purpose bridge was built, and the
resulting bridge is shown schematically in Fig. 5.

Capacitors cP and cs are worm-driven air capacitors with a range of
about 220 puyf, and were specially designed for this bridge. In the case of
cs, any direct conductance between rotor and stator would result in an
effective series resistance which would vary both with frequency and
capacitor setting, and therefore require laborious correction. This was
avoided by arranging the construction so that the rotor and stator are
mounted on independent insulating supports to the ground panel, thereby
completely eliminating direct conductance from rotor to stator. While
this results in some conductance from test terminal x1 to ground, the
amount is small and its effect is negligible because of the relatively low
‘impedance values measured. In the case of cp, on the other hand, it is
important to minimize series resistance and inductance to avoid con-
ductance and capacitance corrections which would change both with
frequency and capacitor setting. This was accomplished by careful design
of the rotor brush using silver contact surfaces and center-fed connections
to both rotor and stator.

The conductance standard, cp, and resistance standard, rs, were de-
signed to emphasize high-frequency performance. Deposited carbon resis-
tors’ on ceramic rods 3” in diameter and 2” long mounted on small
decade rotors were used, so arranged that only one resistor on a rotor is
in the circuit at any time, and that adjacent resistors are short-circuited
by means of auxiliary shorting brushes to eliminate shunting admittance
which might vary with frequency. For er the resistance values are such
that the two lower decades and the slide-wire rheostat each have a
residual conductance of 333 micromhos, thereby avoiding the use of
resistors exceeding 3,000 ohms in value which would be more likely to —
vary with frequency. The structure is designed to minimize series in-
ductance and to maintain constant capacitance for all settings. For Rs,
on the other hand, it is necessary to maintain constant inductance for
all settings. This was accomplished by adding small wire-loop compen-
sating inductors in series with individual resistors in the 10-ohm and
100-ohm decades when necessary. To minimize the over-all inductance,
the resistor rotors are placed very close together and are driven by gear-
ing from the corresponding dials.

rotor switch on which are mounted five uncalibrated mica capacitors
which enable cs to measure both positive and negative reactance values
up to 10,000 uuf without additional switching. The 20 uyf cr capacitor
covers capacitance measurements up to 60 yuf; the 40 uyf capacitor
covers up to 150 uf; the 80 uuf up to 600 uyf; the 140 uuf up to 10,000
puf; and the 200 yyf capacitor covers all the positive series reactance
measurements. Since the cr capacitor permits the bridge to be balanced
with the test leads short-circuited, the value of the effective resistance
under test is simply equal to the difference between rs readings for the
measurement balance and the short-circuit balance, and the reactance
under test is determined from a computation of the two readings of cs.

A front view of the general purpose bridge is shown in Fig. 6. The four
lower dials are for Gp; above them are the four Rs dials; and above them
is the cr dial. The capacitors cs and cp are located adjacent to the test
terminals, but are operated remotely by the dial knobs at the extreme
right end of the bridge. This was done to remove the operator’s hands as
far as possible from the test terminals. Near the test terminals is a coaxial
connector engraved a. This allows plug-in capacitors (cu in Fig. 5) to
be added in parallel with cp for extending the capacitance range. Com-
pact silvered mica capacitors in steps of 200 uf are used. Fig. 7 shows
the interior of the same bridge with cp and cs in the lower foreground,
ap at the left and Rs in the upper right.

Fig. 6—Front view of the general-purpose bridge shown in Fig. 5. The bridge
is approximately 103 inches high and 19 inches wide.

Fig. 7—Interior view of the general-purpose bridge. The panel edge shown in
the foreground is the left edge of the bridge shown in Fig. 6.

Fig. 8—The five-megacycle Maxwell inductance bridge is approximately 12}
inches high and 19 inches wide. Test terminals are at upper left, and the three
knobs at lower right are zero-balance adjusters.

Fig. 9—Interior of bridge of Fig. 8 showing the shielding for test terminal X1
in foreground; at the left is the calibrated air capacitor; at the right are the con-
ductance decades using glass-sealed deposited-carbon resistors.

Development of capacitors for the L3 coaxial system has required a
new ten-megacycle admittance bridge. Intended especially for determin-
ing temperature coefficient and frequency characteristics of small capa-
citors, the bridge is capable of measuring capacitance values up to 200
uuf with a precision of +0.01 uyf, and a wide range of conductance values.
Unlike the other two bridges described which make grounded measure-
ments only, this bridge is arranged for direct and balanced-to-ground
measurements as well. This is accomplished by using the ratio-arm trans-
former already described in combination with a simple grounding circuit
using a three-position key, as shown in the bridge schematic of Fig. 10.

Fig. 10—Ten-megacycle admittance bridge with three-position key for shift-
ing ground to B for measuring direct admittance, to junction of Cl and C2 for
balanced admittance, and_to D for grounded admittance. Unknowns may be
connected from A to D or Cto D.

Bridges have been developed for the measurement of impedance and
admittance parameters at megacycle frequencies with accuracies hereto-
fore possible only at much lower frequencies. Several of the twenty-
megacycle general-purpose bridges have been built and are furnishing
useful measurements of networks and components. Experience with these
bridges has indicated ranges for which supplementary special-purpose
bridges would be desirable, and two such bridges have been built: a
Maxwell bridge for low-valued inductors, and an admittance bridge for
low-valued capacitors. One feature of all of these bridges not generally
available in commercial measuring instruments for megacycle frequen-
cies is the provision of standards having a range of several decades. These
allow balances to be made with greater precision over a wider range of
phase angles in the apparatus under test, and assure that the absolute
accuracy will not be limited by readability. This added precision is very
useful in comparing similar components or in measuring characteristics
such as temperature coefficient.

1. W. J. Shackelton, ‘‘A Shielded Bridge for Inductive Impedance Measure-
ments,’”’ Bell System Tech. J., 6, pp. 142-171, Jan., 1927.

2. W. J. Shackelton and J. G. Ferguson, ‘‘Electrical Measurement of Communi-
cation Apparatus,’ Bell System Tech. J., 7, pp. 70-89, Jan., 1928.

3. J. G. Ferguson, ‘“Measurement of Inductance by the Shielded Owen Bridge,”’
Bell System Tech. J., 6, pp. 375-386, July, 1927.

5. J. G. Ferguson, ‘‘Shielding in High-Frequency Measurement,”’ Bell System
Tech. J., 8, pp. 560-575, Aug., 1929.

Abstracts of Bell System Technical Papers”
Not Published in This Journal

A Full Automatic Private-Line Teletypewriter Switching System. W. M.
Bacon! and G. A. Locke’. Trans. A.I.E.E., 70, Part 1, pp. 473-480,
1951. (Monograph 1837).

This paper describes a full automatic teletypewriter message switching system
for use in private-line networks involving one or more switching centers and a
multiplicity of local or long-distance lines, each of which may have one or more
stations. This system provides fast teletypewriter communication from any
station to any other station or group of stations in the network. At its point of
origin a message first is perforated in tape accompanied by suitable directing
and end-of-message characters, thereafter it is transmitted automatically, stored
temporarily in perforated tape at a switching office, and then routed at high
speed to its point or points of destination. Important features are the arrange-
ments provided to permit efficient use of long full duplex transmission lines, the
full automatic handling of multiple-address messages with only a single originat-
ing transmission, and the various guards and alarms which.are provided to
protect against loss of messages in case of trouble.

Operational Study of a Highway Mobile Telephone System. L. A.
Dorrr. Trans. A.I.E.E., 70, Part 1, pp. 31-37, 1951. (Monograph
1838).

The Dynamics of the Middle Ear and Its Relation to the Acuity of Hear-
ing. H. Furrcumr’. J. Acoust. Soc. Am., 24, pp. 129-131, March, 1952.

The transformer action of the middle ear as measured by Bekésy is shown to
be the principal cause for the low acuity of hearing for low frequencies. Because
of the very low mechanical impedance across the basilar membrane at low fre-
quencies, large acoustical pressures in front of the ear drum produce appreciable
acoustical pressures across the basilar membrane. For example, at 100 cps this
pressure is thirty times and at 6000 cps it is one-tenth that created across the
basilar membrane.

Diffusion of Donor and Acceptor Elements Into Germanium. C. §.
Fuutuer’. Phys. Rev., 86, pp. 136-137, April 1, 1952.

* Certain of these papers are available as Bell System Monographs and may be
obtained on request to the Publication Department, Bell Telephone Laboratories,
Inc., 463 West Street, New York 14, N. Y. For papers available in this form, the
monograph number is given in parentheses following the date of publication, and
this number should be given in all requests.

A Submarine Telephone Cable with Submerged Repeaters. J. J. GILBERT’.
Trans. A.J.E.E., 170, Part 1, pp. 564-572, 1951. (Monograph 1815).

Physical Structure and Magnetic Anisotropy of Alnico 5. Part I. R. D.
Hemenrercu and E. A. Nessirr’. Jl. Appl. Phys., 23, pp. 352-371,
March, 1952. (Monograph 1976).

It is concluded from electron metallographic results that the high coercive
force and anisotropy of Alnico 5 are caused by a very finely divided precipitate
produced by the permanent magnet heat treatment. This precipitate is a transi-
tion structure rich in cobalt and is face-centered cubic with a9 = 10A and ap-
pears as rods growing along the [100] directions of the matrix crystal when no
magnetic field is applied during heat treatment. The size of the precipitate rods
at optimum properties is approximately 75-100A by 400A long. The spacing
between rows of rods is about 200A. The rods are not distinctly resolved in the
electron images unless they are grown by aging at 800°C. Their orientation and
structure is clearly evident in the electron diffraction patterns at all stages of
growth. The precipitate responds to a magnetic field applied during heat-treat-
ment both by suppression of nuclei making an angle greater than about 70° with
the field and by the forcing of the rods off the [100] direction into that of the
field. The precipitate rods tend to scatter in direction about the field vector when
the field is off the [100] but are aligned accurately when the field is along [100].

It is shown that the band or itinerant electron model of a solid is capable of
accounting for the “exchange stiffness’? which determines the properties of the
transition region, known as the Bloch wall, which separates adjacent ferromag-
netic domains with different directions of magnetization. In this treatment the
constant spin function usually assigned to each running electron wave is replaced
by a variable spin function. At each point of space the spin of a moving electron
is inclined at a small velocity-dependent angle to the mean spin direction of the
other electrons, and this gives rise to an exchange torque which makes the spin
direction of the given electron precess as it moves through the transition region,
the precession rate being just sufficient to keep it in approximate alignment with
the macroscopic magnetization. Physical insight into the mechanisms involved
is provided by a rigorous solution of the wall problem for a ferromagnetic free
electron gas in the Slater-Fock approximation, although it is known that the
free electron gas is not likely to be ferromagnetic in higher approximations.
Rough upper limits to the exchange stiffness constants for actual ferromagnetic
metals can be calculated without using any empirical constants other than the
saturation moment and the lattice constant. The results are only a few times
larger than the observed values.

Elastic and Plastic Properties of Very Small Metal Specimens. C.
HerrinG and J. K. Gaur’. Phys. Rev., 85, pp. 1060-1061, March 15,
1952. (Monograph 1977).

A Scanner for Rapid Measurement of Envelope Delay Distortion. L. E.
Hunt’ and W. J. Aupersuetm’. Proc. I.R.E., 40, pp. 454-459, April,
1952. (Monograph 1967).

possible mechanisms of the current multiplication process in the Type-A transistor
are discussed. One of the mechanisms is based on trapping holes in the collector
barrier of the semiconductor. By means of this trapping model, the effect of
emitter current and temperature on the current multiplication is predicted. It
is shown that these predictions are in reasonable accord with experiment.
Furthermore, assuming this model to hold, the trap density and activation energy
(produced by forming) may be evaluated.

Transistor Forming Effects in n-Type Germanium. L. B. Vaupzs’.
Proc. I.R.E., 40, pp. 445-448, April, 1952. (Monograph 1969).

Some of the effects of electrical forming of the collector of an n-type germanium
transistor are discussed. Evidence is presented for the existence of a region of
p-type germanium underneath the formed electrode, together with some indica-
tion of the size of the formed region. These experiments lend support to the p-n
hook mechanism in that they explain the observed high values of alpha in tran-
sistors. This relation is discussed.

Domain Structure of Perminvar Having a Rectangular Hysteresis Loop.
H. J. Wiuitams’ and M. Gorrrz’. Jl. Appl. Phys., 23, pp. 316-323,
March, 1952. (Monograph 1985).

An investigation has been made of the magnetic domain structure of Permin-
var (43 per cent Ni, 34 per cent Fe, 23 per cent Co) ring specimens having rec-
tangular hysteresis loops after heat-treatment in a magnetic field. Domain
patterns obtained with colloidal magnetite showed curved domain boundaries
extending completely around the rings, forming circles concentric with them.
Changes in magnetization occur when an applied field causes the circular bound-
aries either to expand or contract so that there is a change in the relative values
of clockwise and counter-clockwise flux. A nucleus of reversed magnetization
was formed by making a small notch in a specimen, and this decreased the co-
ercive force and hysteresis loss by a factor of two. It was found that in a 180°
domain boundary it was possible to make the change in spin orientations, which
occurs in going from one side of the boundary to the other, have either a right-
or left-hand screw relation, by the application of a field of appropriate sign per-
pendicular to the surface. The effect of superposing an applied alternating field
was also investigated, and an effective permeability of 4,000,000 was obtained.

Measuring Techniques for Broad-Band. Long-Distance Radio Relay
Systems. W. J. ALBERSHEIM . Proc. I.R.E., 40, pp. 548-551, May, 1952.
(Monograph 1971).

Line-up and maintenance of radio relay systems require sensitive yet rapid
measurements. These are obtained by scanning the systems response as func-
tions of time, frequency, and amplitude. Parameters thus scanned include the

transient response to step functions; frequency characteristics of gain, phase,
impedance and their frequency derivatives; and amplitude characteristics of
output nonlinearity and of intermodulation products.

Aluminum Die Castings—The Effect of Process Variables on Their
Properties. W. Bastncron’ and D. H. Kueprincer!. Proc. A.S.T.M.,
51, pp. 169-197, 1951.

Diffusion in Alloys and the Kirkendall Effect. J. BARDEEN’ and C.
HERRING!. pp. 261-288 of Imperfections in Nearly Perfect Crystals, Wiley
N. Y., 1952, 490 p. Edited by W. Shockley, J. H., Hollomon, R. Maurer
and F. Seitz. Symposium held at Pocono Manor, Oct. 12-14, 1950, by
Committee on Solids, National Research Council.

Lightning Protection for Fixed Radio Stations. D. W. Bopur’. Tele-
Tech, 11, pp. 58-60, 126+, June, 1952.

Common grounds, parallel conducting paths, and discharge gaps provide
three important means for avoiding equipment damage from high current surges.
Protection of connecting facilities must also be considered to preserve service.

Compression Tests on Lead Alloys at Extrusion Temperatures. G. M.
Bouton’ and G. 8. Pures’. Proc. A.S.T.M., v. 51, pp. 761-770, 1951.

Load-deflection measurements made during compression tests on lead and
lead-alloy cylinders at various temperatures show the effects of alloying in-
gredients on the force required to produce deformation. The curves also furnish
clues as to changes taking place in the materials during the course of the test.
The load, P, to produce definite small deformation in pure lead at various tem-
peratures, T, are shown to follow the relationship P = Ae~®7, where A and B
are constants for the material. This is the same relationship found by others in
extrusion studies. The elements added to lead were those most commonly used
in the manufacture of cable sheath, namely, antimony, arsenic, bismuth, silver,
tellurium, and tin. The results show that the stronger alloys now used for cable
sheathing deform less readily at extrusion temperatures than pure lead or the
weaker alloys.

Liffect of Prior Strain at Low Temperatures on the Properties of Some
Close-Packed Metals at Room Temperature. W. C. Exuis’ and E. 8.
Greiner’. J. Metals, 4, pp. 648-651, June, 1952. (Monograph 1966).

The Fatigue Test as Applied to Lead Cable Sheath. G. R. Goun’ and
W. C. Exits’. Proc. A.S.T.M., 51, pp. 721-740, 1951.

This paper discusses the more important factors affecting the design of labora-
tory test methods suitable for obtaining significant fatigue data from reversed
bending tests on cantilever-beam specimens of lead cable sheathing alloys.
Data are presented to show the effect of cycling rate, temperature, shape of
specimen, alloy additions, and aging on fatigue life. The close correlation be-
tween bending fatigue tests on strip specimens and full size sections of cable is
demonstrated. The fatigue data are analyzed in terms of (1) cycle life versus
deflection, (2) cycle life versus strain, and (3) cycle life versus stress. Photo-
micrographs illustrating representative laboratory and field failures are included.

Thermal Conductivity of Germanium. A. Grizco’ and H. C. Monr-
gomErRY. Phys. Rev., 86, p. 570, May 15, 1952. .

Bell System Cable Sheath Problems and Designs. F. W. Horn’ and R. B.
Ramsry’. Trans. A.J.E.E., 10, Part 2, pp. 1811-1816, 1951. (Monograph
1917).

Powdered Standards for Spectrochemical Analysis. E. K. Jaycox’.
Applied Spectroscopy, 6, pp. 17-19, May, 1952. (Monograph 1978).

Lingineering for Low Product Cost and High Product Quality at the
Western Electric Company. A. C. Jonus®. Ind. Quality Control, 8, pp.
53-59, May, 1952.

A successive-approximations method is applied to the selection of network
functions having desired magnitude and phase variation with frequency. The
first approximation, the first set of pole and zero locations, can be selected on
the basis of known solutions to similar problems or through use of a set of curves.
In succeeding approximations the pole and zero locations are adjusted to decrease
the deviation of the earlier approximations from the desired characteristics. The
process adjusts the magnitude and phase characteristics simultaneously. Its
flexibility permits accommodation of practical constraints not possible with
other methods.

The Magnetic Structure of Alnico 5. E. A. Nessrrr’ and R. D. Hewen-
rEIcH. Elec. Eng., 71, pp. 530-534, June, 1952. (Monograph 1981).

In the investigation of Alnico 5, two problems arose. What is the mechanism
which enables the alloy to respond to heat treatment in a magnetic field? What
causes the alloy to have a high coercive force of 600 oersteds? The first problem
has been solved and progress has been made toward solving the second.

Single-Frequency Signaling System for Supervision and Dialing Over
Long-Distance Telephone Trunks. N. A. Newett' and A. Weaver’.
Trans. A.I.E.E., 70, Part 1, pp. 489-494, 1951. (Monograph 1841).

The single-frequency signaling system for long-distance telephone trunks
frees dial calls from the range and other limitations imposed by de signaling
methods. It uses alternating currents in the voice range as the signaling medium
and so can be used with any trunk of any length or type of line facility which
meets voice-transmission requirements. The signaling requirements, design
problems, main features of the circuit and equipment arrangements, and the
operation of this system are outlined in this paper. The system described is the
first practical arrangement of its type satisfactorily to meet all the conditions of
telephone service in the Bell Telephone System.

Experimental Information on Slip Lines. W. T. Reap, Jr’. pp. 129-
1p1 of Imperfections in Nearly Perfect Crystals, Wiley, N. Y., 1952,
490 p. Edited by W. Shockley, J. H. Hollomon, R. Maurer and F. Seitz.
Symposium held at Pocono Manor, Oct. 12-14, 1950, by Committee
on Solids, National Research Council.

On the Geometry of Dislocations. W. T. Reap, Jr.’ and W. SHockuey’.
pp. 77-94 of Imperfections in Nearly Perfect Crystals, Wiley, N. Y.,
1952, 490 p. Edited by W. Shockley, J. H. Hollomon, R. Maurer and
¥. Seitz. Symposium held at Pocono Manor, Oct. 12-14, 1950, by Com-
mittee on Solids, National Research Council.

A Servo System for Heterodyne Oscillators. T. Suoncznwsxr. Trans.
A.J.E.E., 70, Part 1, pp. 1070-1072, 1951. (Monograph 1883).

A constant rate of progression of frequency of a motor-driven heterodyne
oscillator is obtained by comparing its output with a frequency standard. The
result is fed into a servo loop which drives the motor at the proper speed. When
used in connection with a level recorder a linear frequency scale is obtained which
is more accurate than the static calibration of the oscillator.

Metallic Rectifiers in Telephone Power Plants. D. E. Trucxssss’.
Trans. A.I.E.E., 70, Part 2, pp 1464-1467, 1951. (Monograph 1987).

Metallic rectifiers are a comparatively new means of converting power from
alternating current to direct current. Most of the component apparatus used
in the Telephone Systems operates with direct current while the normal power
source is alternating current. Therefore a static device without expendable parts
which is obtainable in small and large current capacity lends itself as a means ox
power conversion in telephone power plants.

A. B. Cuark, B.E.E., University of Michigan, 1911. A. T. & T. Co.,
1911-34; Bell Telephone Laboratories, 1934-. Toll Transmission De-
velopment Engineer, 1929; Toll Transmission Development Director,
1934; Director of Transmission Development, 1935; Director of Systems
Development, 1940; Vice President, 1944. Bell System Chairman of
Joint Subcommittee on Development and Research of the Edison Elec-
tric Institute and Bell System since 1938. Since June, 1951, Mr. Clark
has been in charge of coordinating all Bell System programs at the
Laboratories. During World War II he served both as a consultant to
and a member of various divisions of the Office of Scientific Research
and Development. In 1944 he was appointed Consultant to the Secretary
of War, and in connection with this work made trips to the European
and Mediterranean theaters of operation. Member of I.R.E., Tau Beta
Pi, Sigma Xi, and A.A.A.S. and Fellow of A.I.E.E. and the Acoustical
Society of America.

J.R. Fry, M.E., Cornell University, 1915. Western Electric Company,
1915-25. Bell Telephone Laboratories 1925-. Mr. Fry has been Assistant
Switching Apparatus Engineer in the Switching Apparatus Development
Department since 1946. Except for the years 1941-45, when he worked
on military projects, most of Mr. Fry’s Bell System service has been
devoted to the design and development of electromagnetically operated
switching apparatus such as relays, switches, registers, and selectors.
Member of Eta Kappa Nu.

H. C. Monreomery, A.B., University of Southern California, 1929;
M.A., Columbia University, 1933. Bell Telephone Laboratories, 1929-.
Prior to the war, Mr. Montgomery was engaged in studies of hearing
acuity and the analysis of speech sounds. His recent work in the transis-
tor physics group has been concerned with fluctuation phenomena in
semiconductors.

SAMUEL P. Moraan, Jr., B.S., California Institute of Technology,
1943; M.S., California Institute of Technology, 1944; Ph.D., California
Institute of Technology, 1947. Bell Telephone Laboratories, 1947-. A

ment System, 1944-1945; member, Hoover Medal Board of Award,
1945-1951; and Chairman, Board of Trustees of Volta Memorial Fund,
1949-. He also has been active in the American Standards Association. —
He was long a member of the Board of Directors, and was Chairman of
the Standards Council from 1942-1945 and Vice President 1948-1951.
Since 1949 he has been President of the U. 8. National Committee of the
International Electrotechnical Commission. He is a member of the Joint
Conference Committee on Standards of the Department of Commerce
and ASA, and Chairman of the U. 8. N. C. Executive Council Subcom-
mittee. He is Fellow of the American Institute of Electrical Engineers,
Acoustical Society of America, American Physical Society, American
Association for the Advancement of Science, and of the Institute of
Radio Engineers; and is a member of the American Society for Engineer-
ing Education and of Tau Beta Pi.

J.J. Prturop, E.E. 1908, D.E. (Hon.) 1939, Ohio Northern University ;
A. T. & T. Co., 1908-. From 1910 until 1943 Mr. Pilliod was associated
with the Long Lines Department and the General Engineering Depart-
ment of the American. Telephone and Telegraph Company. From 1914
to 1918 he was Division Plant Engineer in Chicago; 1918-1920, Engineer
of Transmission, New York City; 1920-1941, Engineer in charge of
Long Lines Engineering Department; and 1941-1948, General Manager
of the Long Lines Department. In 1943 he assumed his present position
as Assistant Chief Engineer of the American Telephone and Telegraph
Company. From October 1942 to April 1943 he was Chief of Signal
Section, Production Division, Army Service Force. He is a Fellow of the
A.I.E.E. and is a Trustee of Ohio Northern University and of Vassar
College.

F. F. Surpuey, B.S. in E.E., Purdue University, 1925. A. T. & T. Co.,
1925-34; Bell Telephone Laboratories, 1934—. Since 1948, Mr. Shipley
has been switching engineer in charge of planning large automatic
switching systems, both local and toll. This includes panel, crossbar,
and large step-by-step epee Member of the A.I.E.E., Tau Beta Pi,
and Eta Kappa Nu.

H. T. Wituetm, B.S. in E.E., Cooper Union, 1927; E.E., Cooper
Union, 1936. Western Electric Company, 1922-24; Bell Telephone Lab-
oratories, 1925-. Since joining the Laboratories Mr. Wilhelm’s work has
been with the Transmission Apparatus Development Department, where
he has designed electrical measurement apparatus and developed test
methods. Member of A.I.E.E. and Tau Beta Pi.

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