Line-Power Feed
Line-Protection Switching
Centralized Transmission Surveillance

J. B. FRY, Art and Production Editor
F. J. SCHWETUE, Circulation
W. G. SCHEERER, Coordinating Editor of L5 System Articles

THE BELL SYSTEM TECHNICAL JOURNAL is published ten times a year by
the American Telephone and Telegraph Company, J. D. deButts, Chairman
and Chief Executive Officer, R. D. Lilley, President, J. J. Scanlon, Executive
Vice President and Chief Financial Officer, F. A. Hutson, Jr., Secretary. Checks
for subscriptions should be made payable to American Telephone and Telegraph
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Broadway, New York, N. Y. 10007. Subscriptions $15.00 per year; single copies
$1.75 each. Foreign postage $1.00 per year; 15 cents per copy. Printed in U.S.A.

Coaxial-carrier transmission systems constitute a significant portion
of the Bell System long-haul transmission facilities. These systems
have been developed over several decades to provide the basis of a
high-quality, high-capacity, long-distance communications network.

After extensive exploratory work on wideband amplifiers and coaxial
cable, the feasibility of a coaxial-carrier system was demonstrated in
1936 between New York and Philadelphia. The success of this trial
was followed by development of the first Bell System coaxial-carrier
transmission system, the L1. Placed in service in 1941, the L1 system
was initially capable of carrying 480 four-kHz two-way message chan-
nels per pair of 0.270-inch-diameter coaxial cables, with a repeater
spacing of 5.5 miles. Soon, 0.375-inch-diameter cable became standard
and, with system improvements, the vacuum-tube-operated L1 system
was capable of carrying 600 circuits per coaxial pair with 8-mile
repeater spacing. Its capacity was later increased to 720 circuits.

The major expense of the coaxial system has been in the outside
plant area: cable, cable placement, right-of-way, and buildings,
including aboveground or underground structures for housing re-
peaters. Once this portion of the system is established, development
of electronic equipment to provide maximum utilization of the cable
becomes economically attractive. Each successive generation of
repeaters achieved wider transmission bandwidth and, hence, larger
channel capacity through use of shorter repeater spacing, new tech-
nology, and more advanced system concepts. Use of cables with more

Telephone Typical ‘
Circuits pone PEPIN
First Units Repeater Honor: Other Key
System Services. Pee poi aere Technology Repeater Types Equalization System Features
Per Sheath | Pray
Coax Per (Miles)
Pai Cable*
air
Li 1941 720t 720 4 8 Vacuum tube Manually and automatically | Adjustable static
2,160 8 adjusted regulating repea- and dynamic
ters; equalizing repeaters “bump’’ shapes
L3 1953 1,860 5,580 8 4 Vacuum tube, Buried-thermistor and line- Manual “‘cosine”’ Hardened con-
,300 12 statistically con- pilot-controlled regulating shapes; dynamic figuration
trolled key repeaters; equalizing broad shapes
components repeaters
L4 1967 3,600 32,400 20 2 Discrete transistor, | Fixed basic repeaters; regu- Static ‘‘bump” Noise objective more
printed wiring lating repeaters controlled shapes stringent by 4 dB
board by both a buried thermis-
tor and line pilot;
equalizing repeaters
L5 1974 | 10,800t | 108,000{ 22 1 Discrete transistor, Static ‘‘bump”’ Phase-shaping net-

t Originally 480 circuits per coaxial pair, but widely used at 600-circuit capacity.

t$ Extensions to L5 are being developed to provide 13,200 telephone circuits per coaxial pair, or 132,000 circuits per 22-tube cable.

works to control
third-order modu-
lation addition;
same noise objec-
tives as in L4

coaxial units per sheath increased route capacity and further reduced
per-channel-mile costs.

From the outline of the evolution of coaxial-carrier systems in Table
I, we can see that, in 33 years, the channel capacity of repeatered
coaxial line has increased by a factor of 22.5—from 480 to 10,800
channels, and further increases are anticipated. In the same period,
improvements in cable technology allowed the manufacture of cable
with 5.5 times more coaxial units in the cable sheath—from 4 tubes to
22 tubes—resulting in a 10-fold increase in signal-carrying capacity, not
including the two units reserved for service protection. The total
impact, then, was a 225-fold increase in route capacity. During the
same 33 years, Bell System circuit miles increased 485-fold.

This issue of The Bell System Technical Journal describes in detail
the latest in the line of coaxial systems—the L5 Coaxial-Carrier
Transmission System. The articles include descriptions of an advanced
systems approach and sophisticated repeatered-line and equalization
designs. Also included are the novel concepts in repeatered-line power-
ing, line-protection switching, equipment-performance surveillance,
centralized maintenance, and carrier reference-frequency generation.
Other articles describe the new multiplex and signal-administration
equipment and the important role in the success of L5 played by
innovations in physical design and thin-film techniques and by the
use of ultralinear semiconductor devices. The many computational
aids and measurement facilities that were effectively used in the
development of the system are also discussed.

The initial L5 system—815 miles of cable, 14 stations, 850 manholes,
3400 manhole repeaters, and over 250 bays of transmission equip-
ment—went into service on January 38, 1974, fulfilling a schedule
developed six years earlier. This on-time completion of such a massive
system required the dedicated effort of many individuals in the Bell
System companies. Bell Laboratories people conceived and developed
the system and its components; Western Electric people were re-
sponsible for manufacture of cable and electronic equipment and
for installation of main-station equipment; and AT&T and Long
Lines people were actively involved in system planning, coordination,
route selection, cable placement, installation of line equipment, and
operational testing. It is to this skilled Bell System team that this
issue is dedicated.

Copyright © 1974 American Telephone and Telegraph Company
Tue Bet System TECHNICAL JOURNAL
Vol. 53, No. 10, December 1974
Printed in U.S.A.

The LS Coaxial-Carrier Transmission System is a new solid-state
broadband system designed to transmit 10,800 long-haul message channels
on standard 32-inch (9.5-mm) coaxial cable. Alternatively, high-speed
digital signals may be transmitted in selected mastergroups. This paper
contains an overall system description and describes in detail system design
considerations such as the transmission plan, frequency allocation, and
4000-mile notse performance. Also considered are the significant proper-
ties of the coaxial medium, the ways they affect the hierarchy of equalizers
traditionally found in long-haul coaxial systems, and the manner in
which the several levels of equalization relate to each other and to the
fundamental transmission phenomena within the system.

During the last 30 years, long-haul transmission facilities have been
rapidly expanded to meet the increasing need for both voice and data
transmission, and it is expected that rapid growth will continue in the
1970’s although possibly at a slower rate than the 15 percent per
year which typified the late 1960’s. Thus, during this period there
has continued to be a need for new systems providing increased
message capacity at lower cost to supplement the systems already
available in the long-haul plant. As can be seen in Table I, the L5
Coaxial-Carrier Transmission System is the latest in a family of coaxial
transmission systems developed to date.*

It is evident that the newer coaxial systems have achieved increased
capacity because of better utilization of the transmission medium (i.e.,

*The sheath capacities listed in Table I are the ones most commonly associated
with each of the systems. The initial L3 application, in 1953, was actually on Coax-8
(i.e., a cable which includes eight coaxials) with a total sheath capacity of 5580
circuits on three working pairs, with one pair reserved for protection.

Li L3 L4 L5
First service 1941 19538 1967 1974
Technology Tubes Tubes Transistors | Transistors
and HIC’s
Repeater spacing (miles) 8 4 2 1
Channels per pair 600 1,860 3,600 10,800
Total channels/sheath 1,800 9,300 32,400 108,000
Noise objective (dBrnc0) 44 44 40 40

wider bandwidth) and because of an increase in the number of coaxials
incorporated within the sheath. An important characteristic of such
long-haul systems is the dominant effect which the installed cost of
the transmission medium has on the overall cost. By providing triple
the message capacity on L5 as compared to L4, for example, the cost
of a message-circuit mile on L5 is substantially less than that of a
message-circuit mile on L4, even though the number of repeater
stations is doubled and the electronics required is of wider bandwidth
and of increased sophistication. In addition, the older L-carrier systems
use the same transmission medium, and it is possible to convert many
existing L1, L3, or L4 routes to L5 by adding the required number of
new repeater sites to the existing sites of the earlier system. In this
manner, increased message capacity can be obtained at an incremental
cost that is even more attractive than on a new installation.

As noted in Table I, the L5 coaxial system provides 10,800 toll-grade
quality message channels (on a pair of 0.375-inch disc-insulated coaxial
cables). In a Coax-22 cable configuration, the resultant total channel
capacity is 108,000 channels per cable sheath when a 1:10 protection
strategy is employed. The increase in the bandwidth of the system
required for this capacity is made possible in part by setting the nominal
repeater spacing at 1 mile and in part by several significant techno-
logical advances in such areas as repeater, network, and device
design.!-3

The Ld system includes fixed gain ‘“‘basic’’ repeaters at 1-mile
nominal intervals, adaptive ‘‘regulating”’ repeaters at 7-mile maximum
intervals, adjustable gain ‘‘equalizing” repeaters at 38-mile maximum
intervals, and power-feed main station repeaters at 75-mile maximum
intervals.* Line protection switching is provided on a 1-for-10 basis at
maximum intervals of 150 miles.®

The adjustable equalization of the L5 repeatered line is partially
manual and partially automatic. Manually adjusted equalizers
compensate for the transmission deviations, which are essentially
time-invariant. These equalizers are adjusted to minimize the mean-
squared error of the broadband channel upon initial installation, and
any subsequent readjustments are normally done out of service. The
use of out-of-service, mean-squared-error adjustment procedures
permits a substantial improvement in the transmission quality for a
particular array of adjustable equalizers.

Automatic equalizers compensate for the time-variant transmission
deviations. These equalizers are located in each regulating, equalizing,
and main station repeater and fall into two categories. The automatic
equalizers for the cable-temperature effect are located in each of these
repeaters and utilize both open-loop regulating circuits, which detect
the resistance of a locally buried thermistor, and closed-loop regulating
circuits, which detect the level of a full-time line pilot. The automatic
equalizers for the repeater-temperature effect and the regulating
repeater tracking error are located only in the receiving main stations
and utilize closed-loop regulating circuits equipped with ‘‘memory,”’
which detect the levels of four full-time line pilots.

Provisions are included at appropriate main stations for adding,
dropping, or branching any jumbogroup (3600 channels) at line
frequency. ‘Dropping’ involves connection to the jumbogroup
multiplex (Jmx), a new frequency-division multiplex (rpm) terminal
designed for L5, which translates up to three basic jumbogroups
(0.564 to 17.548 MHz) to appropriate line frequency assignments
between 3.124 and 60.556 MHz.’ “‘Branching”’ involves connection,
at line frequency, to or from another L5 facility. ‘‘Adding”’ involves
the addition to an L5 line of circuits originating from the mmx. The
mastergroups of a basic jumbogroup are administered much like the
mastergroups of an L4 signal. The basic jumbogroup can consist of
any available combination of analog and digital mastergroup signals.
When frequency assignments permit, direct connection with other
facilities is made possible by connecting all mastergroups to the Jmx
via a basic jumbogroup trunk bay (BsetT). This trunk bay permits
mastergroups from several different sources, such as mastergroup
multiplex (amx-2), L-carrier mastergroup digital (LMp) terminals,
and radio systems, to be combined into a basic jumbogroup.® The
carrier generation within the ymx requires reference frequencies of an
accuracy heretofore unavailable in the Bell System and consequently

a jumbogroup frequency supply (sFs) is included at all smx locations
for this purpose.®

At selected main stations, an L5 transmission surveillance center
(tsc) will be utilized to centralize and partially automate the surveil-
lance and fault location of several hundred miles of repeatered line
and associated main station and multiplex equipment.’ In the initial
L5 installation between Lillyville, Pa., and Hillsboro, Mo., a single
Tsc serves these functions for 664 miles of ‘“‘backbone”’ route Coax-22,
and 151 miles of ‘‘sideleg’’ Coax-12.

All long-haul coaxial systems share the common objective of toll-
grade signal-to-noise quality in a 4000-mile circuit. In the L5 system,
this involves equalization of over 100,000 dB of cable loss at the upper-
most message frequencies and almost 132,000 dB at the uppermost
“controlled” frequencies near 70 MHz. Equalization of this loss is
accomplished with a combination of wideband circuits, all of which
must be low in noise, highly linear, and extremely stable. It is in the
achievement of the 4000-mile transmission objective that the familiar
coaxial system hierarchy of increasingly complex repeaters evolves.
Among the general features of the system, the frequency allocation
will be considered first. Although many different factors must usually
be considered during the specification of a frequency allocation, only
a few can be reviewed here.

The controlling consideration in establishing the frequency allocation
is the overall bandwidth involved—both in terms of hertz of bandwidth
and in terms of the ratio of uppermost to lowermost frequencies to be
used—which appears to be consistent with the state of the art at the
time, or within the evolutionary lifetime of the system development.
An important and relevant experience of previous coaxial-systems
development has shown the difficulty and expense of simultaneously
controlling the transmission characteristic at the highest and lowest
frequencies of a spectrum that spans a frequency range of as much as
five to six octaves. Realizable transformers, inductors, and capacitors
are in most cases far from ideal components, and parasitic effects be-
come especially troublesome when the frequency range is more than a
few octaves. As the system bandwidth approaches five octaves or so,
parasitic effects frequently become dominant. Thus, for a given band-

width, there may be significant advantages to raising, within reasonable
limits, the lowest frequency requiring careful control.

Compatibility with the existing rpm hierarchy is another relevant
factor and, in the case at hand, the basic building block was taken to be
the six-mastergroup assembly formed in accordance with the L4
transmission plan. The six mastergroups are arranged between 0.564
and 17.548 M¥Uz and are designated the ‘“‘basic jumbogroup.”

A basic system objective was the transmission of at least three
times the L4 message capacity of a single jumbogroup. The assignment
of the jumbogroup line frequencies and the spacing between the jumbo-
groups depends mainly on the bandwidth considerations already
discussed and the types of signals and signal administration procedures
planned for the system. In L35, the capability to select any jumbogroup
from the line spectrum at line frequency (‘branching and dropping’)
was a basic system objective. Questions relating to the design of
appropriate bandpass and band-stop filters substantially affect the
guard spacing required between jumbogroups. In addition, at the time
that it was necessary to make the frequency allocation decisions, the
ability to displace an entire analog jumbogroup with a high-speed
digital signal was also a basic objective.

The frequency allocation of the L5 system is shown in Fig. 1, with
six mastergroups, each containing 600 message channels, making up
the basic jumbogroup. In some cases, a mastergroup in the L5 line
spectrum will transmit a 13.29-megabits-per-second digital signal by
means of the L-carrier mastergroup digital (LMp) terminal," a digital
repeater spaced at maximum intervals of 300 miles. These digital
channels will be used to carry a variety of digital signals including
wideband data, pps, and digitized Picturephone® signals.

The regions around 1.59 and 68.65 MHz are used to transmit fault-
locating signals which in turn are used to remotely locate repeater
faults from a centralized location equipped with an L5 transmission
surveillance center.

The region around 68.78 MHz is used to transmit signals required
by the line-protection-switching system (Lpss-3) for interstation com-
munication, and control and verification of Lpss operation.

Pilots are transmitted below, between, and above the jumbogroups
to control the L5 automatic equalizers. The automatic equalizers for the
cable-temperature effect, located in the regulating repeaters, are
controlled by the incoming level of the 42.880-MHz pilot (which is
often referred to as the ‘‘temperature pilot’’). The automatic equalizer

IN MHz 70.144(C) FL: FAULT-LOCATING SIGNAL
42.496(C) E: EQUALIZING PILOT
SW: LPSS TELEMETRY
42.496(C)
45.056(C)
91.648(C)
40.448(C)
LLeo Oo ° SsSsom
ao + 8 aSBR Ss
eminent Se ; © oo OO
-- N Qa g oon oo
I (goeood|Geoooo (feoooo I I
S 88 8 5 :
: & 8 8 8 8

Fig. 1—L5 frequency allocation. The basic jumbogroup shown is equivalent to the
L4 line spectrum. The carrier frequencies indicated are those used to translate the
basic jumbogroup to the L5 line frequency assignment.

for the repeater-temperature effect and the regulating repeater tracking
error, located in receiving main stations and designated the E3 equal-
izer, is controlled by the incoming levels of all four pilots.

After the establishment of at least an approximate frequency
allocation, the central concept in the design of the coaxial system
becomes the equalization of the 4000-mile cable loss. Furthermore,
it must be done in such a way that the residual transmission deviations
are less than 0.01 percent of the top channel cable loss, while accum-
ulated noise from all sources, including associated multiplex, must sat-
isfy the long-haul noise objective of 40 dBrncO. The general features
of the system layout selected to meet these goals are described next.

The most general features of a repeatered-line layout in long-haul-
coaxial systems depend mainly on power-feeding techniques, equaliza-
tion strategy, reliability requirements, and convertibility of older sys-
tems. The equipment providing power for the remote repeaters, the fine-
grained equalization of transmission, and the automatic line-protection

switching will generally be of such size and sophistication as to require
a station environment as opposed to the manholes that are suitable
for the simpler repeaters. The cost of these main stations is many times
that of a manhole, and it is essential that the number of such stations
be minimized for system costs to be competitive with alternative ways
of providing the circuits.

The power-feed configuration maximizing the length of the power
span for a given voltage is the double-ended connection shown in
Fig. 2.

The L5 line power supplies provide a nominal 0.91 A de via the
center conductors of the coaxial cables for powering up to 75 miles
of remote line repeaters. This is a convenient submultiple of the L4
power span length and requires a maximum line voltage of only
+1150 V, with the result that physical designs of high reliability
should be readily achievable, and corona ‘‘popping”’ in the installed
system should be minimal.

The basic layout is affected by the equalization strategy mainly by
the effect the latter has on repeater signal loading and the extent of
signal level misalignment that must be accepted by the line repeaters.
The L5 strategy in most cases uses pre- and postequalization in equal
parts and, as a result, repeater load requirements can be reduced for a
given transmission deviation from nominal or, for a fixed repeater
load capacity, the permissible distance between equalizers can be
increased, since the magnitude of the transmission deviation is usually

= GRD PROT
(OPERATES WHEN
IMBALANCE IN TWO LINES
EXCEEDS PRESET THRESHOLD)

Fig. 2—Simplified diagram of line power feed. Plus or minus voltage is developed
by dc-de converters and combined with message in power separation filters prior to
application to coaxial line.

proportional to distance. Finally, when the misalignment is contained
to within about +5 dB, it can be shown that this approach achieves
optimum signal-to-noise ratio in both second- and third-order inter-
modulation-limited systems. Figure 3 shows the signal-to-thermal
noise penalties associated with single-ended equalization, at either
the receiver or the transmitter only, and with equal parts of pre- and
postequalization.

Finally, a relatively simple adjustable equalizer is included in the
midspan manhole repeater so that considerations of equalization and
load capacity do not entirely determine the main station interval.
The midspan equalizer provides coarse adjustment of signal levels as
they traverse the power feed span, and not the fine-grain compensation
of transmission deviations, which is accomplished only at the main
stations.

Pre- and postequalization is applied throughout the L5 repeatered
line except for the automatic equalization of the repeater-temperature
effect, the controlling considerations in this case being simplification
of system operation and reduction of system complexity. The repeater-
temperature effect is equalized only in the receiving portion of main
station repeaters, which are up to 75 miles apart.

Reliability affects system layout, since the reliability of the system
electronics determines the length of the protection-switching span and

HO” 38 6. «4 22 0 2 4 6 8 10
MISALIGNMENT 1N DECIBELS
Fig. 3—Signal-to-thermal noise penalties resulting from signal misalignment. For
misalignment less than about +5 dB, pre- and postequalization in equal parts
(curve 2) also results in negligible penalties in second- and third-order intermodula-
tion noise (not shown).

SWITCHING REGULATING MIDSPAN SWITCHING
POWER-FEED REPEATER (EQUALIZING) POWER-FEED POWER-FEED
MAIN STATION \ REPEATER MAIN STATION MAIN STATION

BASIC \
REPEATER __¥
see RECEIVING | tons RECEIVING
aan REPEATER | fora NeR REPEATER

the number of lines required as standby for any given availability
objective to be achieved. The predicted reliability of the L5 line
repeaters permits up to two power-feed spans to be traversed before
automatic line-protection switching is required, and then on a 1-for-10
basis. In the context of the repeatered line, the basic building block
of the system is the switching span and, in the absence of signal
administration needs (dropping, branching, etc.), such spans as are
shown in Fig. 4 are connected in tandem until the distance between
terminal stations is bridged.

In addition to the provision of repeatered-line power, fine-grain
equalization, and (sometimes) line-protection switching, other im-
portant functions are carried out at the different types of L5 main
stations, and, depending on the particular features required of main
stations, they fall into one of three categories. In order of decreasing
complexity they are:

(¢) Terminal station/terminal main station (Ts/TMs).
(21) Switching power feed main station (SPFMS).
(i272) Power feed main station (PFMs).

The functions performed at these stations which are directly related
to the repeatered line are alike. The differences among them are
in the signal and pilot administration (line connecting) functions,
the line-protection switching, the transmission-surveillance capabilities,
and the multiplexing and related signal-processing functions.* Some
main station and repeater functions are summarized in Fig. 4, and
some possible terminal interconnections for jumbogroup administration
are shown in Fig. 5.

A basic objective in the design of the L5 system was to achieve a
system requiring a minimum of day-to-day routine involvement of
craftspeople. The L5 main stations, as well as the repeatered line, will
generally be operated without personnel present. This has been made
possible in part by developing equalization procedures that should
require activity only rarely, and in part through the implementation
of centralized alarm, surveillance, and fault-locating techniques. The
layout of the initial L5 commercial installation between Lillyville, Pa.,
and Hillsboro, Mo., is shown in Fig. 6 and, for purposes of the follow-
ing discussion of system organization and administration, is treated as
typical.

Installation of the L5 repeatered line is accomplished by telephone
company personnel and is controlled and supervised at ‘‘maintenance
centers.”’ Six such maintenance centers are shown in Fig. 6; the four
at Danville, Xenia, Morgantown, and Noble have responsibility for
all but the 30 repeater sections near Lillyville and the 14 repeater
sections near Hillsboro. Those four stations are consequently each
responsible, on the average, for the installation and maintenance of
nearly 200 miles of repeatered line which, when fully equipped, will
include approximately 4000 line repeaters.

Upon being received from the factory, every L5 repeater is carefully
tested at one maintenance center before installation along the re-
peatered line. The tests made involve primarily transmission measure-
ments that are executed, on site, by a computer-operated transmission
measuring set and associated test stand. A hard copy of the test
results is produced during the process and, in the case of an acceptable
repeater, is retained in the maintenance center as part of the permanent
history file on that repeater. Regulating repeaters and fault-locating
oscillators are among system components requiring on-site adjustment

RECEIVING TRANSMITTING
' REPEATER i = REPEATER |
7
|
COAXIAL | / paranoid TRANSMITTING : | COAXIAL
Line | |P BRANCH Plot LINE
S EV/EZ/E3 | CIRCUIT circuit [23 heal E1/E2 S
I |e S| “(e2) an
| RECEIVING TRANSMITTING
EQUALIZER i) = EQUALIZER
TO OTHER FROM OTHER
L5 SYSTEM L5 SYSTEM

Fig. 5—Jumbogroup administration. In the hypothetical case shown, sal is through-connected at line frequency and JsG2 is
branched on to another L5 system also at line frequency. Ja3 is dropped via JMx and the Byer to basic jumbogroup for interconnec-
tion with mmx-2, L4, a radio system, or a combination thereof.

BEAUCOUP KENTUCKY
HILLSBORO MAIN ROUTE — 664 MILES
SIDELEGS | — 151 MILES
TOTAL — 815 MILES
MISSOURI (a)

(2) HILLSBORO, MO.
[>] BEAUCOUP, ILL.
() NOBLE, ILL.
[>]  SHELBURN, IND.
(2) MORGANTOWN, IND
[>]  CONNERSVILLE,
?
(2) S XENIA, OHIO
w
[>] HILLIARD, OHIO
(>) DANVILLE, OHIO
[3]  WAYNESBURG,
(2) LILLYVILLE, PA.
Gy CINCINNATI, OHIO
GY? covumaus, ox10
or DAYTON, OHIO
<q WILLIAMSTOWN, KY

(b)
Fig. 6—(a) Geographical layout of initial L5 commercial installation between

Lillyville, Pa., and Hillsboro, Mo. (b) Administrative layout of Lillyville-Hillsboro
route.

in accordance with the particular repeater station to which they are
assigned, and those adjustments are normally made utilizing the same
test setups.

Installation of the L5 main station equipment is usually done by
Western Electric installation personnel and, prior to turnover of the
equipment to the telephone company, a variety of transmission, noise,
and operational tests is usually necessary. In most cases, these tests
utilize the same kind of test equipment required by the telephone
company to install, maintain, and operate the system.

After the testing and installation of the repeaters and fault-locating
oscillators have been completed between a pair of power-feeding points
(such as Lillyville and Waynesburg in Fig. 6), and after the main
station equipment has been installed and tested, power is applied by
operating the line-power-feed dc-dc converters at both ends of the line.
The converters at each end are adjusted to share equally the load of
the intervening repeatered line (except in the case of very short power
spans, which can be powered by converters at one end only). With
power applied to the line, a detailed step-by-step process, known as
the L5 line acceptance procedure and appropriately documented, is
executed, which permits the systematic identification and elimination
of any system faults that may have escaped detection during the
earlier testing of the individual repeaters and cable sections. In
addition, such engineering or operational errors as incorrect line-
build-out specification, incorrect fault-locating oscillator adjustment,
or improper pre-regulator settings can, with a combination of engineer-
ing and on-site analysis, readily be detected. The line acceptance
procedure flows from a detection of the gross faults that may have
existed, through a regulator-by-regulator pilot level measurement
sequence, to the measurement, evaluation, and equalization of the
channel between main stations, and ultimately to the noise loading
of the tandem power-feed spans making up a protection-switching
span (such as between Lillyville and Danville in Fig. 6). Upon com-
pletion of this procedure, the repeatered line is ready for the next
level of system testing, which would consist of such procedures as
exercise and validation of the protection-switching system, including
tandem operation (such as from Lillyville-to-Danville-to-Xenia in
Fig. 6) and multiplex-to-multiplex measurements and evaluation
(such as between Lillyville and Xenia).

A wide variety of status and alarm indications has been made
available in the L5 system design to provide for complete and ready

remote verification of system health as well as diagnosis of system
troubles when they occur. Status indications are provided for such
operating parameters as the dynamic equalizer range setting and the
condition of the protection-switching system, while alarm indications
are developed based on the power level of any line or jmx pilot, the
condition of any fuse, and a variety of other parameters. These alarm
and status indications are transferred to a central alarm station which,
in Fig. 6, is located at Williamstown, Ky., off the L5 route. This station
is part of the E2 Status Reporting and Control System and, in the
following discussion, references to E2 will be to this system, not to the
E2 equalizer that is an integral part of the L5 system. Transmission
between the Ld stations and the E2 central station takes place over a
data network which is a four-wire private line, meeting requirements
similar to those of a standard voice-band data service. (In the case of
the initial L5 installation, the data facility is located internal to the co-
axial cable, utilizing two of the loaded 19-gauge pairs, as far as
Cincinnati, and utilizes an external private line circuit between
Cincinnati and Williamstown.) Each L5 main station is equipped with
at least one E2 “remote unit,’”’ which provides the interface between
the L5 system alarms and the E2 data facility. The larger Ld stations,
such as Xenia and Morgantown on the initial route, require two E2
remote units, because of the quantity of equipment to be located there.
The total load on the Williamstown E2 central is 16 remote units, the
upper limit for a ‘manual’ E2 central on a single data network.

In its normal operating mode, the E2 central sequentially interro-
gates (alarm polls) each remote unit for its alarm poll response. When
all remote units have been polled, the process is successively repeated.
A manual operation at the central or a demand by the L5 transmission
surveillance system (Tss) for use of the data facility will interrupt this
otherwise automatic and repetitive alarm polling. The manual opera-
tions at the central include calling up a particular station display for a
detailed alarm report as well as controlling such L5 system operations
as effecting a line switch or operating the line-feed converters.

surveillance center (Tsc), located at Xenia. Equipped with a mini-
computer central processor, and loaded with appropriate data-base,
operating-system, and applications programs, that Tsc controls the
fault-location and surveillance processes for all the repeaters and main
stations in the initial system. (The total mileage represented by
Fig. 6 is approximately 815 miles. When fully equipped, the equipment
“domain” of the tsc at Xenia will include over 17,000 line repeaters
and more than 250 transmit-receive bays, as well as associated multi-
plex.) Virtually all the periodic maintenance measurements planned
for the system will be carried out at the Tsc and, by supplying a full
duplex data circuit between Xenia and Williamstown, operation of
the tsc can be transferred to Williamstown during Xenia’s unmanned
intervals.

The transmission plan of an analog coaxial system is determined by
adjusting the bandwidth, repeater spacing, and signal transmission
levels so that prescribed channel objectives are realized over the life
of the system. To transmit over a particular bandwidth on a specified
cable, the repeater spacing is determined by the available gain, noise
figure, load-carrying capacity and linearity of the repeater, and the
margins required to allow for cable and repeater variations resulting
from aging, temperature changes, and manufacturing limits.

Table IT illustrates how the basic repeater noise, load capacity, and
linearity design objectives became progressively more stringent as
coaxial-system design proceeded from L3 through L5. In comparing

System Data L3 L4 L5
Top frequency (MHz) 8.3 17.5 60.6
Repeater performance, referred
to top channel:
Noise figure (dB) 1] 6.5 5.5
Load capacity (dBm) 16 21 24
Nonlinearity
Second order (dB) —61 —70 —70
Third order (dB) — 96 — 100 —110*
Insertion gain (dB) 44 33 31
Transmission level (dB) —lil —14,2 —13.6

the repeater design parameters of the table, it is important to note
that the required noise, load capacity, and linearity improvements
from L3 through L5 are achieved at progressively higher top channel
frequencies. Table II shows, for example, that operating at one-half the
L4 repeater spacing and more than three times the L4 top channel fre-
quency, the L5 basic repeater requires a 1-dB better noise figure, 3-dB
more load capacity, and 10-dB better third-order nonlinearity than
the L4 basic repeater, with all parameters referred to their respective
top channel frequencies. Top channel insertion gain is slightly lower in
the L5 repeater despite the higher top transmitted frequency because
of the 2:1 reduction in repeater spacing. °

The L5 system signal-to-noise design objective is based on meeting a
4000-mile per channel noise objective of 40 dBrncO0. The 40-dBrnc0
total is allocated as 39.4 dBrncO to the high-frequency line and 31.5
dBrnc0 to the terminals (Fig. 7 shows how the 31.5 dBrnc0 terminal
allowance relates to a possible multiplex deployment). In the signal-
to-noise design of the system, high-frequency line transmission levels
are shaped with frequency to realize noise performance in every
channel as nearly equal as possible. In L5, second-order modulation

JMX JMX JMX JMX JMX
JUMBOGROUP —— >»
(3600 CHANNELS)
MMX-2 MMX-2 MMX-2 MMX-2 MMX-2
MASTERGROUP- —-> \A
(600 CHANNELS) t
LMX-3(SG) LMX-3(SG) LMX-3(S6) § EXPOSURES LMX-3(SG)
SUPERGROUP. ——-»> \A
(60 CHANNELS) t
LMX-3(GRP) LMX-3(GRP) 3 EXPOSURES LMX-3(G RP)
GROUP- ———— > Ss
(12 CHANNELS) |
AG 3 EXPOSURES AG
ORIGINATE TERMINATE

Fig. 7—Possible multiplex terminal deployment in a long-haul interconnection.
This example includes 11 intermediate multiplex operations between the originating
and terminating stations, which is well above the number typically encountered.

noise is the controlling modulation contributor because of the use of
the phase-shaping networks discussed below, which serve to reduce the
third-order intermodulation distortion to a relatively insignificant
level. In L5, application of signal level shaping techniques results in
the transmission levels shown in Fig. 8a and in the predicted zero
level noise shown in Fig. 8b. The curve labeled ‘‘Total’’ in Fig. 8b
is the predicted noise under nominal conditions. The uppermost curve,
labeled ‘‘Total (with allowances),’’ is the predicted noise performance
including penalties for misalignment, misequalization, and the like.
As can be seen, the average noise over all channels is predicted to
be 39.7 dBrncO under nominal conditions. The use of frequency
frogging of the jumbogroups and mastergroups will tend to cause
the typical 4000-mile L5 channel to approach that average value in
noise performance. In addition, measurements on the L6 field trial
and first commercial installations indicate close agreement with
4000-mile predicted noise proportioned to the appropriate test system
length. A controlling factor in the determination of the allowable
second- and third-order modulation coefficients listed in Table II is
the use of repeatered line phase-shaping networks in each Ld regulating
repeater. Such networks are used to produce a significantly less linear
repeatered line phase-frequency characteristic, resulting in reduced
correlation of third-order (A + B — C)-type product addition from
repeater to repeater. In the absence of phase-shaping, such product
addition would occur on a nearly in-phase or voltage-addition basis,
with the repeater-to-repeater terms approaching 20 log N, where N
is the number of tandem line repeaters.*

In the presence of the phase-shaping networks, such as those used in
L5, substantially lower correlation of (A + B — C) product addition
results. For the LS 6-mile regulating-section phase-frequency charac-
teristic shown in the upper curve of Fig. 9, the repeater-to-repeater
addition term within a frogging link is approximately 15 log N, cor-
responding to a per-frogging-link (A + B — C)-type product reduction
of about 15 dB relative to the “‘no-phase-shaping” values. Thus, in Ld,
the use of phase shaping reduces third-order intermodulation noise to
relatively insignificant values, and the system is second-order modula-
tion and thermal-noise limited.

* Product addition according to a particular law occurs over “‘frogging”’ link inter-
vals of up to 800 miles. Because of the randomizing effect on product and source
frequencies produced by frogging, random or power addition is assumed to occur
from frogging link to frogging link for all modulation product types.

Fig. 8—(a) Repeater output transmission levels. The discontinuities in the TL shown result from using a stepped approximation to
the optimum (smooth) preemphasis curve, derived by transmitting Ja2 1.7 dB hotter than jel and ja3 1.8 dB hotter than 3q@2 at
the ymx output. (b) Predicted zero level noise for 4000-mile L5 system. The prediction is based on extensive individual repeater
and transmit-receive bay measurements, and has been confirmed by noise loading measurements made on lengths of the initial
system up to 750 miles long.

Fig. 9—Relative phase (linear component subtracted) of L5 regulating section—
with and without phase-shaping networks in regulating repeater. The parameter }
is a measure of the “‘stiffness’’ of the ¢-f characteristic of the networks, and f, is
the frequency at which the phase shift of each of the four sections is 180 degrees.

The typical L5 cable (new installation) will be a Coax-22 and will
contain 22 disc-insulated coaxials and 42 polyethylene-insulated

conductor pairs. (A cutaway view of a Coax-20 is shown in Fig. 10.)
The paired conductors in these cables are used for support systems
such as order wire, fault location, and air-pressure telemetry.

The loss characteristic of the cable over the band of interest in L5
can be approximated by

The first term of the right-hand side is a result of the resistance loss
of the copper conductors and, at L5 frequencies, owing to skin effect,
this loss is proportional to the square root of frequency. The second
term reflects the dissipation in the polyethylene disc spacers that
maintain the relative positions of the inner and outer conductors, and
it increases linearly with frequency. (This term is often called the
“power factor” term.) The third term reflects the temperature depend-
ence of the copper resistivity, and is also proportional to the square
root of frequency.

Although the cable loss representation in eq. (1) is perfectly adequate
over the transmission band of Ld and its predecessors, distortions in the
cable loss will be encountered as system bandwidths are extended
further, which in some cases may limit those bandwidths. For example,
structural return loss (SRL) spikes are encountered, which result from
the cable-stranding operation during manufacture. The strander
periodically deforms the coaxials as they precess about the cable
core, and, for a fixed stranding lay, the sru spikes are coherent and
result in relatively narrow-band transmission deviations from the
behavior predicted by eq. (1). The current lay length for the outer
coaxials in Coax-20 (cables having 20 coaxial tubes within the sheath)
and Coax-22 is 36 inches +3 inch and results in sri spikes at 157 + 2.5
MHz. Other distortions are caused by seam interaction or ‘back
twist,’’ which is also a product of the stranding operation. These occur
in the outer coaxials of the Coax-20 and Coax-22 around 170 MHz.
The inner units of Coax-20 and -22 (8 in each case) and all the units of
Coax-12, since they have shorter lay lengths, exhibit these effects at
higher frequencies than those noted for the 36-inch lay length. For the
purpose of designing the hierarchy of repeaters and equalizers that
make up the repeatered line, the loss described by eq. (1) becomes the
basic objective for repeater gain per mile of system, as discussed next.

Fig. 10—Coax-20 of serrated seam construction. This cable is the backbone of
nearly all existing L4 routes, and includes 52 paper-insulated wire pairs in addition
to the 20 coaxials. New L5 construction will normally utilize Coax-22, specifically
developed for L5.

Figure 11 shows that the loss of 1 mile of 2-inch coaxial cable reaches
about 33 dB at 70 MHz. It accumulates to over 130,000 dB in 4000
miles, and is compensated by a variety of equalizers located throughout
the system. When the transmission characteristic or effect being
equalized is both predictable and time-invariant, fixed equalizers are
appropriate, but when the effect to be equalized is either unpredictable
(in a given installation) or time-invariant, some form of adjustable
equalization is necessary.

4.2.1.1 Nominal cable loss. The principal loss in the system that is
both time-invariant and predictable is the nominal cable loss, which is
compensated by the fixed-gain basic line repeaters. If the 4000-mile
system transmission objective were to be met under nominal conditions
by adequately matching the gain of the basic line repeaters to the loss
of the adjacent cable section, a match would be required of approxi-
mately 0.0008 dB at the uppermost L5 frequencies, which is neither
realistic with respect to design complexity nor consistent with measur-
ing and control capabilities of present-day manufacturing processes.
The principal transmission components of the basic repeater are a

4.2.1.2 Deviations from nominal repeater spacing. When the repeaters can-
not be installed at the nominal 1-mile intervals, the resultant spacing
deviations are compensated for by the abovementioned LEO net-
works which simulate the loss of a length of the coaxial cable, and
are available in 0.1-mile increments from 0.1 to 0.5 mile. Because
of obstacles, access problems, or other geographical limitations, it
is usually impossible to achieve exactly 1-mile spacing between ad-
jacent repeater sites. In such cases, the line repeaters are assigned
one of the available LBo’s to increase the loss of shorter cable sections
to that of 1 mile of cable. For unusual obstacles such as river crossings,
for example, a limited number of long repeater spacings may be
permitted by requiring the repeater sections on both sides of the long
section to be shorter than nominal. The administrative rules for all
these cases are described in detail in engineering documentation
available to the telephone company engineer at the time of route
layout and route planning.

4.2.1.3 Average design error. In previous coaxial systems, such as L3
and L4, the equalization of the average difference between line repeater
gain and cable loss (i.e., the average “design error’) has been attempted
with families of fixed equalizers called “design deviation equalizers”’
or simply ‘‘deviation equalizers.’’ The functions of such equalizers are,
first, to contain the transmitted signal levels at close to nominal
values and, second, to enhance, in conjunction with all the other
equalizers in the system, the ultimate transmission response of the
system. While the first of these can often be guaranteed with relatively
high confidence, the enhancement of the equalized transmission
response is more difficult to achieve as a product of such equalization
strategy.1 This section discusses some reasons for this and some
considerations involved in seeking an optimum overall set of equalizers
(fixed plus adjustable).

As mentioned before, the initial design objective for a deviation
equalizer is the complement of the average design error of the line
repeaters. As used here, this error is the difference, on the average,
between the gain of the line repeaters and the loss of the associated
nominal length of coaxial cable. Ideally, the basis for computing the
average difference would be the universe of all manufactured line

repeaters and cable. In reality, the basis must be the largest sample
of each that is available before final manufacturing commitments
must be made on the equalizers.

Even if the optimum information were available when desired,
network realization limitations leave a residual error requiring further
equalization, and when the deviation equalizers consist of a family of
several different designs (as in L4, for example) there are several
different realization errors. Naturally, the specific errors for different
physical models of the “same” network also tend to be different
(because of what is usually called the manufacturing deviation).
Finally, even if the realizations were ideal, the average design error
characteristic of the repeaters used as a basis for the deviation equalizer
design will not always be characteristic of the set of repeaters in any
particular line section.

The collective impact of these factors is a tendency to introduce
“ripples” into the frequency response of the channel. Even though
characterized by smaller amplitude deviations than the unrippled
deviation, the rippled frequency response can often be more difficult to
equalize. Of equal importance, the variations from line to line in the
rippled characteristic resulting from the above factors will often be
more difficult to track with a selected set of adjustable equalizers than
would the line-to-line differences in the unrippled deviation (i.e.,
that existing before the application of the deviation equalizers).

To the extent that these factors contribute significantly to the
ultimate residual error of an equalized channel, the transmission
response of that channel would apparently benefit from minimizing
the number of different types in the family of networks, and/or
minimizing the number of locations at which the deviation equalizers
are inserted. The straightforward application of these guidelines leads
to a single deviation equalizer design inserted as infrequently (along
the repeatered line) as possible, while the most extreme interpretation
would lead to no deviation equalizers as such, a condition that probably
could be achieved only with the addition of relatively costly adjustable
equalizer stations, since the signal levels must still be contained
within the repeater load capacity as the signal traverses the repeatered
line. *

It is possible that the latter might be a desirable solution for a single
equalizing section, which in L5 will be up to two power-feed spans.

*The cost of a fully equipped L5 manhole equalizing station in a medium-cost
zone is about twice that of a fully equipped regulating repeater station, which would
normally be used otherwise.

However, when there is no attempt whatever (such as with deviation
equalizers) to reduce the average misalignment over all equalizing
sections to a value nearer zero, then all the adjustable equalizers in a
long circuit would tend to be set to the same side of their nominal
setting. As a result, there would be no tendency whatever to cancel
(or randomize) the errors and transmission distortions originating in
the equalizers themselves and in their inherent equalizing limitations,
and these would accumulate systematically in a long circuit, quite
possibly becoming the major source of residual overall transmission
deviation. Short of this extreme, a solution based on a single network
design and application in a minimum number of locations is attractive
from several viewpoints, and such a solution has been incorporated
into the L5 design.

The L5 deviation equalizer design provides one-third the total
“design error’ equalization required by a 66-mile repeatered line
(approximate average power span length) in single networks located
in the transmitting and receiving main stations and at the midspan
equalizing station. The network has been designated the ‘22X”
deviation equalizer, since the design objective is approximately the
design error of 22 miles of L5 repeatered line. The particular method
used to realize the deviation equalizer design is described in detail in
Ref. 1. As described there, the L5 deviation equalizer, while in principle
correcting the average design error, is realized in the final design stages
using computer optimization routines that interact with the available
adjustable equalizers. Figure 12 shows the signal misalignment as a
function of distance, per 3 dB of 66-mile correction, employing the
22X approach and assuming uniform accumulation of the misalign-
ment. As far as signal-to-noise penalty is concerned, it can be seen that
this arrangement is equivalent to pre- and postequalization, in equal

Fig. 12—Equalization of the average design error. The relative signal level is
shown for a case involving 3 dB of misalignment in 66 miles; three identical deviation
equalizers are placed at the transmitter and the receiver, and midway between.

X in miles
TRMTG RPTR MIDSPAN RPTR RCVG RPTR
55 to 75 Yes Yes Yes
33 to 55 Yes No* Yes
11 to 33 No * Yes
Less than 11 No * No

parts, of 44-mile sections having +1-dB signal excursion in tandem
with 22-mile sections having +0.5-dB signal excursion, each section
incurring signal-to-noise penalties very near zero (see Fig. 3).

Naturally, administrative rules are required to assist the telephone
company engineer in the determination of his 22X equalizer needs.
With ‘“‘quantum steps” of 22 miles of fixed equalization, it is possible to
specify the fixed equalization to within +22/2 miles of the ideal value,
and the application rules specified for L5 are shown in Table III. Note
that execution of the tabulated rules should approach compensation, on
the average, of the design error in a long (multispan) system that oper-
ates at a uniform mean temperature. In so doing, the rules should result
in slight over- or undercompensation of the individual equalizing
sections which will be of varying lengths. This effect, along with the
significant variations in mean cable temperature that normally will
be characteristic of long systems, should result in a tendency to
randomize the adjustable equalizer settings, thereby mitigating the
systematic accumulation of the errors and distortions originating in
the adjustable equalizers themselves, and in their inherent equalizing
limitations.

4.2.2 Compensation of transmission effects requiring adjustable equalizers

As discussed above, those transmission effects that are ezther time-
variant or unpredictable (or both) require some form of adjustable
equalization. Adjustable equalizers can be automatically or manually
adjusted and the process can be carried out either in- or out-of-service,
as particular circumstances and considerations warrant. In the case
of L5, each of these possible types of equalizer and equalization
processes is utilized among the various members of the equalization
hierarchy.

4.2.2.1 Cable-temperature effect. The largest time-variant transmission
deviation in a coaxial cable system, normally orders of magnitude
greater than any other, is the cable-temperature effect, shown in
Fig. 13 for 1 mile of 3-inch cable and for a 20°F change in cable tem-
perature. Extrapolated to 4000 miles, the temperature dependence
depicted here corresponds to about +3000 dB of loss variation at the
uppermost L5 frequencies, uniformly distributed over that distance.
In LS, this is corrected at intervals not greater than 7 miles in “‘regu-
lating repeaters”? which, in addition to compensating for the nominal
cable loss, provide automatic equalization of the cable-temperature
effect. The regulating repeater compensates for the changing cable
loss within a “‘regulating section”’ which, in typical installations, has
an average length of approximately 6 miles.

In addition to the components found in a basic repeater, the major
transmission components of the regulating repeater are a pair of
adjustable equalizers having the adjustable loss frequency capability
shown in Fig. 13 for the cable-temperature effect. The first equalizer
operates in a closed-loop feedback mode and postequalizes for a portion
of the preceding regulating section. The second equalizer operates in
an open-loop mode by sensing the resistance of a thermistor buried in
the ground near the cable, thereby pre-equalizing for a portion of the
cable-temperature effect of the ensuing regulating section. These two
equalizers restrain the maximum level deviations resulting from the

Fig. 13—Effect of changing earth temperature on 1-mile cable loss. The indicated
temperature range is only rarely exceeded within the continental U.S. at the cable
burial depth of 4 feet.

cable-temperature effect to about 2.5 dB within a regulating section
through the use of the pre- and postequalization method. When the
system is properly aligned, the pre-equalizer in one regulating repeater
and the postequalizer in the subsequent regulating repeater each
compensate for one-half the cable loss variation of the intervening
cable section. The remaining components and a more detailed descrip-
tion of the regulating repeater operation are included in Ref. 1.

4.2.2.2 Repeater-temperature effect. The other major time-variant trans-
mission deviation is the repeater-temperature effect. This results
from both the regulating repeater tracking error associated with
equalizing the cable-temperature effect (a maximum of 0.10 dB per
regulating section) and the direct effect of changing temperature on
the transmission responses of the various line repeaters (less than 0.02
dB per repeater section). Suitable allowances had to be made in the
repeater signal loading requirement to permit this effect to be equalized
at the receiving end only of each power-feed span (up to 75 miles long).
This is accomplished in the ‘‘dynamic equalizer,” a cause-associated,
adaptive equalizer with memory,* the important characteristics and
operational features of which are described in Ref. 1.

4.2.2.3 Statistical manufacturing deviation. The largest time-invariant
transmission deviation requiring adjustable equalization is the sta-
tistical manufacturing deviation that results from the (usually small)
differences in electronic component values within their specified
tolerance ranges. Thus, the exact frequency shape of this deviation is
not predictable for any given collection of repeaters in a particular
installation, although it can be predicted to fall within certain over-
all limits. In L5, this deviation is compensated by equalizers El
and E2 which are manually adjustable because the deviation is time-
invariant once identified. These equalizers consist of 28 adjustable
Bode-type equalizers (“bump” equalizers)—18 in the E2 and 10 in the
El. Both E2 and El equalizers are located in the transmitting and
receiving main stations, while only the El equalizer, mechanically
adapted for manhole use, is utilized at the midspan equalizing re-
peater. The deployment chosen for the equalizing stations is the result
of seeking a reasonable balance between the amount of misalignment
that would have to be tolerated by the line repeaters, with resultant
signal-to-noise penalties and effects on repeater power requirements,

*The memory in the dynamic equalizer permits the transmission response of the
equalizer to remain unchanged if any equalizing pilots are lost.

and the proliferation of the relatively complex (and relatively costly)
equalizing repeaters that would otherwise be required to keep the mis-
alignment and associated signal-to-noise penalties small.

To estimate the static misalignments to be encountered, it is neces-
sary to know some transmission properties of the repeaters which, at
the outset of a system design, either are in the form of repeater require-
ments or, at times, are based on previous experiences with similar
transmission components. In the case of the L5 design, a requirement
was imposed on the line repeaters that (z) the average gain of the
repeaters match the nominal cable loss within +0.1 dB and (772) the
spread among the gains of all manufactured repeaters would be less
than +0.1 dB with respect to that average. This, in combination with
the deviation equalizer strategy described in the preceding section,
made it possible to specify and design a single midspan (between
power-feed stations) adjustable equalizer having both adequate range
and adequate margin (against the possibility, for example, that the
repeater transmission requirements turned out to be unrealistic, or
too costly to achieve, and might thus have been relaxed somewhat).
As mentioned earlier, this also resulted in the spacing of the relatively
expensive power-feed stations being limited by line power considera-
tions, rather than by equalization and repeater load capacity
considerations.

The selection of the type of equalizer terms, or ‘‘shapes,’’ to be used
and the number of terms in the equalizers, as well as the deployment
of the shapes in the frequency domain, was based in part on computer
simulations of the L5 channel as it was estimated to be, and in part
on engineering judgments that evolved from experiences with earlier
systems such as L3 and L4. For example, it was judged possible to meet
the line repeater-cable loss matching requirement with about four or
five peaks in the matching error as a function of frequency. (Iive was
the actual number.) It was further reasoned that (7) the action of the
deviation equalizers, intended to complement the line repeater error
function, would tend to double that number of peaks in the resultant
“fixed equalized” characteristic; (77) the midspan equalizer, while as
simple as possible, should have a number of terms equivalent to the
number of deviation peaks expected in the characteristic for which it
was to compensate (8 to 10); (iz) the action of the midspan equalizer
would tend to increase the number of deviation peaks in the resultant
“ceoarse-equalized”’ characteristic by about the number of terms in
the equalizer (presumably, 8 to 10). This leads to a main station
equalizer being required to deal with a transmission characteristic

having potentially 16 to 20 peaks, whereupon the equalizer would
correspondingly be required to have a similar number of terms. This
type of reasoning provided starting conditions for various computer-
aided optimization routines that ultimately resulted in the 10-term
E1 equalizer and the 18-term E2 equalizer. Similar computer simula-
tions were utilized to evaluate the effectiveness of such different types
of equalizer shapes as the Bode-equalizer-derived ‘“‘bump”’ shapes (as
used in L4) and cosine equalizers (as used in L3). In each simulation
undertaken, a given number of terms could be deployed more effectively
with “bumps” than with alternatives, consequently leading to the
selection of the “bump”’ equalizers for L5. While the initial analyses
and selection of equalizer characteristics had to be based on expected
L5 channel performance, the final deployment of ‘‘bump”’ shapes,
as manufactured for the initial commercial application, was based on
actual L5 repeatered line realizations in the L5 field trial. (The field
trial took place between Cedarbrook and Netcong, N. J., from mid-1970
to mid-1972, with each of four coaxial lines equipped and evaluated
over a distance of approximately 100 miles. The trial resulted in a
variety of important system improvements. )

The manually adjustable equalizers are intended to be set upon
initial installation and only rarely thereafter—for example, when a
substantial increase in the amount of equipment operating in the
manholes results in significant changes in the annual mean operating
temperature of the repeaters. Readjustments of the El and E2 equal-
izers under such conditions will restore the mean-annual setting of the
dynamic equalizers (E3) to midrange, will improve the overall trans-
mission response (since the 28-term E1/E2 combination is more effec-
tive at equalizing the temperature offset effects than the four-term
dynamic equalizer E3), and will improve signal-to-noise performance
by reducing the maximum misalignment. Such readjustment would be
accomplished out-of-service and would involve determination of the
channel misalignment at.a set of frequencies (three per adjustable
term) selected to permit optimum channel equalization in a mean-
squared-error sense. The misalignment of the channel is measured
using programmable transmission measuring equipment under the
control of what amounts to a special-purpose computer designated the
Equalizer Adjustment Unit. The detailed implementation of equalizer
adjustment is described in Ref. 1.

4.2.2.4 Other factors affecting system transmission. For completeness, two
other factors that result in time-variant transmission effects are

mentioned here. One of these is the aging of the elements making up
the system. Aging effects usually are much smaller and much slower
than any effects already discussed. In a solid-state system designed
with careful attention to component selection and component stresses,
there is no aging mechanism comparable to that which was charac-
teristic of the electron tube systems L1 and L3. In L5, aging effects are
expected to be small enough to be insignificant with respect to trans-
mission quality or with respect to such operational aspects as equalizer
adjustment intervals.

The second of these effects is that associated with maintenance
which, along with the aging effect, is normally one of the smallest
time-dependent sources of change in the transmission response of a
system. Over long periods of time there will inevitably be occasional
repeater failures with subsequent replacement of these repeaters by
spares not having the same transmission response as the failed repeater.
(Present estimates of repeater reliability suggest a mean-time-to-
failure in a maximum-length power span of 75 miles because of repeater
failure at between one and two years. Individual failures are, of course,
protected by the tess.) When they become large enough to be detected,
both aging and maintenance effects would normally be compensated
by one of the occasional readjustments of state equalizers El and
F2.

When the characteristics of the L5 repeatered line are examined,
it is evident that it shares many features with previous systems, and
thus in some ways is a hybrid offspring of the L3 and L4 predecessors.
For example, while the adjustable equalizers El and E2 consist of
bump shapes like L4, the adjustments are made out-of-service and
in such a way as to minimize the residual mean-squared error of the
equalized channel, like L3. The cable-temperature effect is compen-
sated by using both open- and closed-loop regulators which pre- and
postequalize the changes in cable loss in equal parts, like L4. But
the 22X deviation equalizer strategy of L5 is much more akin to
the L3 approach. (In L4 there are five different designs, one of which
is installed in every regulating repeater.) While all three systems
utilize line-build-out netowrks to provide some spacing flexibility,
they have fault-locating systems that are quite different. And only
the L5 system supplements the basic fault-location system with
a centralized, computer-operated transmission surveillance system

capable of measuring and monitoring a wide variety of system oper-
ating parameters.

At the time of this writing, the development of the L5 system in its
initial configuration had been completed with all important noise and
transmission objectives having been met. Continuing efforts in the
software offerings of the transmission surveillance system are expected
to increase the already considerable capabilities for automatic and
semiautomatic system maintainance, fault location, and trouble di-
agnosis. A modest level of continuing effort involving the optimiza-
tion of deviation equalizer designs and modification of equalizer
deployment should further improve the system transmission response
and reduce system noise.

Effort is also under way to replace the existing mastergroup multi-
plex equipment (mmx-2c) with individual mastergroup translators,
resulting in reduced noise contribution from that level of multiplex
at lower cost. The possibility of alternative protection strategies for
the jumbogroup level of multiplex are also being examined. In addition,
the basic frequency allocation described in this paper is being reviewed
with an eye toward possible spectrum reorganizations that would
permit an increase in L5 capacity to 22 mastergroups using a new
family of mastergroups and multimastergroup multiplex equipment.

Finally, questions continue to be raised with respect to future
coaxial systems—whether or what or when they may be. Although
advanced coaxial systems having substantially greater capacity than
L5 seem clearly to be within the technological capabilities of the
relatively near future, the need and economic desirability of such
systems has not yet been established. Whether or not L5 becomes the
last of the major long-haul coaxial developments will be determined
in the years ahead primarily by considerations of circuit growth
patterns and economic conditions and by the relative viability and
availability of alternative instrumentalities such as possible new micro-
wave radio, millimeter waveguide, and optical fiber systems.

1. E. H. Angell, Y. S. Cho, K. P. Kretsch, and M. M. Luniewicz, ‘““L5 System:
Repeatered Line,” B.S.T.J., this issue, pp. 1935-1985.

2. F. A. D’Altroy, R. M. Jacobs, J. M. Nacci, and E. J. Panner, ‘“L5 System:
Ultralinear Transistors,” B.S.T.J., this issue, pp. 2195-2202.

3. J. L. Garrison, A. Olsen, Jr., and T. H. Simmonds, Jr., ““L5 System: Transmission
Networks and Magnetic Components,” B.S.T.J., this issue, pp. 2203-2248.

5. J. H. Green and R. W. Sanders, ‘“L5 System: Line-Protection Switching,”
B.S.T.J., this issue, pp. 2011-2034.

6. R. K. Bates and D. J. Zorn, “L5 System: Signal Administration and Inter-
connection,” B.S.T.J., this issue, pp. 2129-2145.

8. J. F. Barry, 8. Narayanan, and J. F. Oberst, ‘“L5 System: Jumbogroup Frequency
Supply,” B.S.T.J., this issue, pp. 2109-2127.

9. J. L. Thomas, R. E. Anderson, and P. J. Baun, ‘“L5 System: Centralized Trans-
mission Surveillance,” B.S.T.J., this issue, pp. 2035-2064.

10. J. F. Gunn, J. S. Ronne, and D. C. Weller, ‘‘The Pzcturephone® System: Master-
group Digital Transmission on Modern Coaxial Systems,” B.S.T.J., 50, No. 2
(February 1971), pp. 501-520.

11. Bell Laboratories Staff, Transmission Systems for Communication, Fourth Edition,
revised, pp. 331-333.

The Ld repeatered line is presented from the viewpoint of a distributed
equalization process. Reliable transmission of 10,800 circuits over 4000
miles of coaxial cable with minimum noise is the sole objective of this
process. The strategy 1s to provide equalization in cause-associated incre-
ments that place specrfic bounds on signal-level excursions, thereby in-
suring an ultralinear, low-noise predictable transmission medium. The
individual causes of signal misalignment, both static and dynamic, are
examined and the realization of the strategy, which forms a hierarchy of
equalizers, 1s described.

The basic function of the Ld repeatered line is to provide a lossless
transmission facility between any two L5 terminal stations, which
may be located as far as 4000 miles (6400 kilometers) apart, with
minimum noise penalty. The objective is to establish and maintain
the insertion loss of the 4000-mile line over the message band to be
within +4 dB and the noise to be less than 39.4 dBrnc0 in any of the
10,800 channels transmitted.

The transmission medium for the L5 line is the standard Bell System
0.375-inch disc-insulated coaxial cable. This cable has a loss* that can
be conveniently expressed as follows:

* A 2.1-percent factor has been included in this loss to account for miscellaneous
factors such as stranding.

There are two frequency bands of interest. The first is the message
band from 3.1 to 60.6 MHz over which the 4000-mile objectives must
be met. To meet these long-haul objectives, however, and to accommo-
date other system functions, a wider band from 1.6 to 66 MHz is
controlled. In eq. (1), the loss associated with the first bracketed term
is a function of frequency only. This is called the static loss. The second
bracketed term is not only a function of frequency, but also of cable
temperature and, hence, of time. This component of cable loss is called
the dynamic cable loss.

The strategy to equalize both static and dynamic losses is to provide
a cause-associated equalization hierarchy—that is, the various equip-
ment making up the L5 repeatered line each compensate for a particular
cause of line misalignment. This relationship is shown in Fig. 1.

This paper discusses the misalignment causes and the associated
equipment design and realization.

Static equalization of the cable is provided in three levels. The first
is provided by the basic repeater. With the repeaters spaced at 1-mile

intervals, the gain required to match the static component of the cable
attenuation given in (1) varies from 4.97 dB at 1.6 MHz to 32.04 dB
at 66 MHz.

The second level is provided by the deviation equalizer. It is neither
economically nor technically feasible to design the basic repeater
with a gain response that exactly matches the cable loss function.
The realization of the repeater design results in a slight mismatch
between the average basic repeater gain and the average cable loss.
This mismatch is referred to as the average design error. An objective
was established to hold this error to less than +0.1 dB over the
frequency band in a 1-mile section or to less than +7.5 dB in a 75-mile
power-feed section. Compensation for this misalignment is provided
by the deviation equalizer located at the transmitting and receiving
ends of a power-feed section and in midspan.

Even if it were possible to exactly match the nominal attenuation
of the cable by the gain of the basic repeater, there would still be
appreciable misalignment in line sections involving many repeaters.
This is due to the statistical manufacturing deviations in repeaters
and cables, and is impossible to predict for a given line section. The
resulting misalignment from both causes is referred to as static line
misalignment or static deviation. Equalization of this static deviation
forms the third level of static equalization in the L5 system, and is
accomplished with adjustable equalizers referred to as El and E2
equalizers.

Specific causes for the statistical manufacturing deviations are
known. Repeater gain deviations are the result of component and
assembly tolerances, while cable loss deviations are the result of
variations in the copper conductivity, cable geometry, and dielectric
disc conductivity. Copper conductivity and cable geometry influence
the A parameter of the cable loss equation and typical variations (one
sigma) resulting from this effect are 6 dB at 66 MHz over 75 miles.
Dielectric disc conductivity, on the other hand, affects the B parameter
of the cable loss equation and amounts to a 1.5-dB change for a one-
sigma variation at 66 MHz over 75 miles. However, changes in disc
material have occurred since the initial cable production some 25
years ago. Measurements made on a number of these earlier cables
indicate a variation in cable loss of about 23 dB at 66 MHz per 75
miles. Since, in addition to new installations, L5 is also intended as a
retrofit system for some of these earlier cable applications, this varia-
tion becomes important.

Equation (1) shows that the cable attenuation varies as the tempera-
ture of the cable changes. In the L5 system, the cable is buried four
feet beneath the ground surface, where the maximum temperature
deviation expected in the United States is +20°F. This results in a
52-dB change in cable attenuation at 66 MHz for 75 miles. Such
change in attenuation is compensated for by repeaters that are called
regulating repeaters. These repeaters automatically compensate for
changes in the cable attenuation by sensing the temperature of the
earth at cable depth and by detecting the level of a pilot tone which is
inserted into the message signal at L5 main stations.

This is not the only component of dynamic misalignment that must
be equalized, however. Since it is not possible to match exactly the
dynamic change of the cable over the entire frequency spectrum, there
will be a slight misalignment or slight difference between the change
in gain of the regulating repeater and the change in loss of the cable,
referred to as the tracking error. In addition, slight differences occur
between the change in gain of the basic and regulating repeaters
themselves as the temperature of their environment changes. These
two effects, taken together, are referred to as the residual dynamic
transmission deviation. While over any short-line section this deviation
is very small, the accumulated effects over a 75-mile section are
typically +3 dB at 66 MHz. This effect is automatically equalized
by an E3 equalizer. The time-varying transmission deviations of an
Ld line are detected by four pilot tones spaced across the line frequency
spectrum. Four networks in the E3 equalizer automatically respond
to the pilot levels to correct the residual dynamic transmission
deviation.

The function of the basic repeater is to provide a fixed gain to
compensate for the attenuation of 1 mile of coaxial cable. Its charac-
teristics have the dominant effect on the overall system performance.
Thus, the requirements it must meet are stringent to assure that the
overall 4000-mile system objectives are met. These requirements cover
numerous areas such as specification of gain match to cable loss,
intermodulation distortion, power-handling capacity, noise figure,

return loss, and temperature coefficient. The simultaneous achievement
of all these specifications is a major accomplishment in the design of
the L5 system.

Two of the most important design requirements are (2) that the
average gain of the basic repeater must match the loss (at 55°F) of one
mile of coaxial cable over the message band to within 0.10 dB and
(27) that the two-sigma limit on the distribution of repeater gains must
not differ from the average characteristic by more than 0.10 dB. These
requirements necessitate an amplifier design with an overall square-
root-of-frequency gain shape of 6.91 dB at 3.1 MHz to 30.69 dB at
60.6 MHz.

The second- and third-order interchannel modulation distortion
(hereafter referred to as intermodulation noise) of the repeater must
be extremely small. The second-order intermodulation coefficient
objective varies from —105 dB at the low end of the spectrum to —70
dB at the high end. The third-order intermodulation coefficient
objective varies from —128 dB at the low end to —110 dB at the high
end. These objectives are derived from calculations involving the
repeater noise figure, output spectral density of the signal, and the
assumed law of addition, from repeater to repeater, for intermodulation
noise.

To avoid a rise in system noise because of peak busy-hour message
load, the basic repeater must be capable of sustaining a load of +24
dBm before overloading. For the L5 system, the conservative definition
of overload is used as the point where the modulation coefficient
degrades by 0.5 dB. An ultralinear high-power amplifier is obviously
required as the result of the overload and intermodulation objectives.

Noise figure of the repeater obviously affects system signal-to-noise
performance and therefore should be kept to a minimum. The noise
figure objective is 8.5 dB at the low frequencies and slowly decreases
to 5.5 dB at the high frequencies. Contrary to overload and inter-
modulation, this requirement calls for a low thermal noise amplifier.
Additional objectives that influenced the amplifier design were surge
protection, return loss, and temperature coefficient.

To summarize, the basic repeater objectives dictate a shaped-gain,
low-noise, high-power ultralinear repeater. One method of achieving
the shaped-gain requirement would be to design a flat-gain amplifier
and a separate network for the shaped loss. The shaped loss would be
equal to the maximum gain required minus the minimum gain required

(for signal transmission). This amount of loss (25.13 dB), if placed at
the input, would lead to an excessively high noise figure at low fre-
quencies; if placed at the output of the amplifier, this loss would
excessively penalize intermodulation distortion and overload. On the
other hand, these penalties may be avoided by designing the amplifier
with a square-root-of-frequency shape (Vf) network in the feedback
path, thus creating a shaped-gain amplifier. As a result, additional
negative feedback is available; it improves the intermodulation dis-
tortion and gain deviations resulting from u-path variations and
improves return loss. However, the shaped-gain feedback amplifier
must be properly compensated to avoid high-frequency stability
problems.

The realization of the basic repeater includes two feedback amplifiers
and a number of passive networks shown in block diagram form in
Fig. 2. Both amplifiers have shaped loss networks in their feedback
paths to accomplish the Vf gain required. Details of the amplifiers
and other networks are given in the sections that follow.

The twin jacks, shown at the repeater input and output, are located
behind the mounting surface for the repeater in the manhole. Their
purpose is to provide connection from the coaxial line to the repeater
with an additional port exposed for outside connection. The outside
connection serves two purposes. First, it allows injection of fault-
locating tones through bridging pads to the repeater input and output.
Second, it allows for ‘power patching” of repeaters. This is done by
removing the bridging pads and replacing them with a coaxial cable
patch. The repeater may then be removed and replaced without
interrupting the coaxial line de power. This simplifies and expedites
repeater replacement.

The bridging pad has an impedance of about 2000 ohms facing the
twin jack. Since it connects with a coaxial plug, there is also a parasitic
capacitance to ground. To maintain the 75-ohm impedance required
at the repeater input and output, an inductor was designed into the
twin jack. This is shown schematically in Fig. 3. By choosing Z such
that VL/C = 75 ohms, a lumped section of 75-ohm coaxial cable was
approximated, and the impedances were maintained. The shunting
effect of the 2000-ohm resistor is compensated for by the impedance
of the low-frequency networks that are discussed in Section 2.33.

COAXIAL CABLE USED IN POWER PATCHING
Ie ee ee ee ee ee we ee we ee we ww os we ae ae ~~

The effect of the twin jack and bridging pad are included in the
performance specifications of the repeater.

The earth-ground filters, shown at the repeater input and output,
consist of sections of high-voltage capacitors and ferrite beads which
form a filter. They provide transmission of signal and power through
the outer wall of the repeater (which must be kept at earth ground
for safety reasons) to an inner “floating ground.” Signals developed
across the high-voltage output capacitors are then sufficiently attenu-
ated by the filter so as not to couple into the input high-voltage capaci-
tor, and vice versa. Failure to provide this filtering action would result

in signals coupling from output to input, which produce unequalizable
ripples across the frequency spectrum of the repeater. The earth-
ground filters are the only high-voltage components in the repeater ;
all other components operate with respect to the internal ‘‘floating
ground.”

The two low-frequency networks (see Fig. 2) differ slightly in internal
component choice, but perform essentially the same functions. The
original purpose of the low-frequency networks was to provide some
loss at the very low frequencies of the L5 spectrum to compensate for
excess gain of the amplifiers. However, during the evolution of the
repeater design, other functions were added to the low-frequency
networks. A simplified schematic of a low-frequency network is shown
in Fig. 4. The network, comprised of Z1, Z2, and ZO, provides the
low-frequency loss shaping previously mentioned. Resistor R1 provides
both an impedance match to 75 ohms (in conjunction with the 2000
ohms in the bridging pad) and a current limiting function for surge
protection. It has a small flat loss effect on the repeater response,
which is compensated for by the main amplifiers.

G1 is a gas tube surge protector having a de voltage breakdown of
about 90 volts and, under normal circuit conditions, introduces only
1.5 pF of shunt capacitance. Inductor L1 provides for the separation
of the dc line current from the L5 message signal. It maintains a high
reactance over the frequency band to keep signal losses to a minimum.

The line-build-out network (LBO) is a passive network that provides
loss equivalent to a given length of coaxial cable. It is used to reduce
the repeater gain from the equivalent of 1.0 mile of cable down to
0.5 mile in steps of 0.1 mile. This allows flexibility for those cases in
which the sections of cable are shorter than 1.0 mile (usually for easy
access to the repeater location or because of a physical obstruction in
the location of the manhole). Since the line-build-out network is an
integral part of the basic repeater, a separate repeater identification
(code) must be associated with each of the six line-build-out networks.
This is in contrast to earlier L-carrier systems in which the line-build-
out network was provided to the field as a separate network. The
advantage of this approach is that it allows for precise control of the
gain interaction between the line-build-out network and the input
impedance of the power amplifier.

To achieve the required gain, noise figure, and distortion require-
ments of the basic repeater, new ultralinear wideband transistors were
required. The characteristics of these transistors and other transistor
design considerations are covered in a companion article.?

As previously mentioned, each amplifier uses negative feedback.
Good stability of the amplifier requires minimization of the length of
the w8 path, which was achieved by employing thin-film technology
with appliqued components, yielding a tri-level circuit realization:
(see Fig. 5). The three layers thus formed are comprised of (7) thin-film

resistors and metal system interconnections, (77) discrete leaded
components (mostly inductors, transformers, and transistors) and
(217) leadless ceramic chip components (capacitors and thick-film
resistors). This tri-level circuit minimizes the w8 path length and
increases the available feedback. In addition, since thin-film resistors
and ceramic chip capacitors have no external leads, parasitic effects
are minimized, thus improving the circuit performance at the high
frequencies required in the L5 system.

The feedback networks of both amplifiers required precision com-
ponents to achieve the required gain characteristic. An initial tolerance
of +0.1 percent is maintained with the thin-film-shaping resistors,
and selection limits of +0.03 dB flatness across the L5 spectrum are
required on the transformers.

Other details of the physical realization of the repeater are discussed
in more detail in a companion article.’

2.3.5.1 Preamplifier. The preamplifier is a negative feedback, hybrid-
input, hybrid-output amplifier depicted in Fig. 6. This circuit configura-
tion was chosen, after careful computer simulation of various alter-
natives, for its low-noise figure while still maintaining good inter-
modulation performance and stability. The input and output hybrid
transformers have impedance ratios of 75:65 + 28 ohms. This ratio
was chosen to minimize the amplifier noise figure, while still maintain-
ing physical realizability and reproducibility of the hybrid. Decreasing
the “‘through loss’”’ of the hybrid to decrease the noise figure results
in increasing the loss through the uf ports of the hybrids, which then
results in an increased minimum gain of the amplifier. It is this mini-
mum gain, resulting from the hybrid effect, that is compensated for
by the low-frequency networks.

The input stages were chosen as common emitter, biased at 30 mA
and 5 volts, to achieve a low noise figure and to provide sufficient
gain to minimize the effect of the noise figure of the output stages on
the overall amplifier noise figure. The output stages are biased at 110
mA and 12 volts for optimum intermodulation performance. Both input
and output stages consist of two transistors connected in parallel to
reduce the intermodulation noise as follows. The input signal current
divides equally, and each transistor then carries one-half the total
current. The controlling third-order nonlinearity is primarily current-
dependent and, as a result, the output distortion of each transistor is
reduced to one-eighth [($)*]. When combined from both transistors,
the output distortion is one-fourth that of a single stage, or a 12-dB
improvement. Similarly, for second-order distortion, a 6-dB improve-
ment is achieved. To achieve these improvements, the transistors are
paired by matching their current gain.

Two adjustable elements are in the preamplifier—a capacitor and a
potentiometer. Each has a very limited range and is used for factory
adjustment of the gain response, particularly at high frequencies.
Computer sensitivity runs were used to choose the variable elements for
matching the normal amplifier gain shape variation. The capacitor
affects the balance between loop and local feedback and hence in-
fluences the closed loop gain through the yf effect, which varies the
repeater gain only at the very high frequencies. The potentiometer
is in the bridged-T 6-network. Changes in its resistance directly vary
the 6 loss and, hence, the amplifier gain. This adjustment changes a
broad, high-frequency shape and is made after the total repeater has
been assembled, to set the gain at 42.880 MHz (the temperature pilot)
to within +0.01 dB of the nominal value. This precise gain adjustment

minimizes gain error introduced by the basic repeater at the tempera-
ture pilot frequency and therefore reduces the range requirements of
the regulating repeater.

2.3.5.2 Power amplifier. The most difficult objective to meet in the
basic repeater was the intermodulation distortion. After an extensive
program of computer modeling of transistor and circuit nonlinearities®
and laboratory evaluation, the circuit topology shown in Fig. 7 was
chosen for the power amplifier. It has shunt feedback at the input
and hybrid feedback at the output. The » path is a common emitter-
transformer-common-base arrangement. The interstage transformer
provides current gain to minimize the effect of distortion in the
common emitter stage and also provides a more nearly optimum set of
interface impedances between the stages. An autotransformer was
chosen as the output hybrid to obtain maximum bandwidth and min-
imum phase shift.

As in the case of the preamplifier, all transistors are paralleled for
improvement in distortion. The bias point of 15 volts. and 110 mA
was chosen to minimize the third-order distortion.

Computer simulation revealed that the closed-loop gain at high
frequency was sensitive to the capacitance at the common base output
node. This node capacitance varies as a result of variation in transistor
capacitance, surge diode capacitance, choke winding capacitance, and
circuit parasitic capacitance. Because these capacitances cannot be
controlled to the desired tolerance, an adjustable capacitor was added.

To meet the distortion requirements, the loop feedback was maxi-
mized. To maintain stability margins at the maximized feedback, an
input hybrid was not used. The input network serves to form part of the
overall amplifier gain shaping, but at the expense of presenting a
nonconstant impedance to the preamplifier (or LBo, if used). This
impedance mismatch creates a gain interaction when line-build-out
networks are placed between the preamplifier and the power amplifier.
To offset this interaction, power amplifiers intended for use in repeaters
with line-build-out networks are adjusted to a different nominal gain.

Buried cable systems, although shielded, are still subjected to a
number of transients induced from within and without the system.
These include (2) high-voltage line turnup,® (22) power patching and
subsequent insertion of an uncharged repeater into the high-voltage
line (227) lightning, and (zv) 60-Hz induction. To protect the repeater

One performance feature that must be determined very early in the
development of a coaxial system is the match of the gain of the line
repeaters to the loss of the cable. This information 1s necessary for the
design of the deviation equalizer, which is described in the next
section. Although prototype information was available on the degree
of match, the results of actual production runs with final component
types are required to establish a sufficiently accurate frequency
response of the repeaters. These data were obtained with the coopera-
tion of the manufacturer, Western Electric, from measurements of the
early production repeaters by Bell Laboratories personnel. The details
of the accuracy and environment of these measurements are described
in a companion paper.’ The results of these measurements, when
compared to the installed line measurements, established a high degree
of confidence in the reproducibility of the design and in the validity
of the parametric equation describing the cable loss. The results also
provided repeater temperature coefficients which were required for
the design of the network shapes in the E3 dynamic equalizer.

The following illustrations demonstrate some performance features
of the basic repeater. Figure 8 illustrates the match to cable loss and Fig.

Fig. 8—Average deviation of basic repeater gain from the average loss of 1 mile of
coax cable.

9 represents typical input and output return loss. Overload and temp-
erature coefficients are shown in Figs. 10 and 11, respectively. The
noise figure of a basic repeater is plotted in Fig. 12. Second- and
third-order modulation are depicted in Fig. 13.

It has been pointed out that exact equalization of the static attenua-
tion component of the cable loss is not feasible. The design trans-
mission objective of the basic repeater is to match the nominal cable
attenuation characteristic to within +0.1 dB. The difference between

the gain of the average repeater and the nominal loss of the cable is
called the average design error and is obviously predictable. Equali-
zation of the average design error is accomplished by providing fixed
equalizers in the transmitting and receiving repeater in the main
stations and in the equalizing repeater located at approximately the
midspan location between main stations. In previous L-carrier systems,
the loss characteristic of the deviation equalizer was determined as

Determine the average design error as a function of frequency .
for each type of line repeater.

Determine the cumulative average design error for the average
expected line section length.

Modify the cumulative design error to compensate for any
design error at the temperature pilot frequency by adding (or
subtracting) the function KV f/fr, where K is the cumulative
design error at the temperature pilot frequency, fr. This is
necessary because any gain offset at the pilot frequency will
be operated on by the regulating repeaters as if it were caused
by a line temperature change.

Apportion the resultant characteristic, which is the desired
total insertion loss characteristic (inverted in sign, of course)
of all line deviation equalizers, equally to all equalizers.
Realize a network to match the desired characteristic according
to some error criterion (usually a minimum mean-squared
criterion).

In any actual line section, a residual gain or loss characteristic will
still remain, since the line will not contain all average repeaters and
cable, and will be of a different than average length. It is difficult to
determine beforehand whether this residual characteristic is equalizable

within the capabilities of the variable equalizers provided in the
system.

In the L5 system, a different design approach was taken. Rather
than the deviation equalizer being designed to match the average
design error, it was designed by a computer program so that the
difference between the deviation equalizer and the average design
error is optimally equalizable by the L5 adjustable equalizers.

The results can be seen in Fig. 14. Curve (a) shows the average design
error of the average L5 power-feed section consisting of 60 basic and
10 regulating repeaters. Curve (b) shows the difference between the
average design error and the deviation equalizers. Curve (c) shows the
equalized characteristic predicted by the computer program using
mathematical descriptions of the equalizers.

This approach has another advantage in terms of the system
development time. Repeater development and variable equalizer
development both require long intervals. Using the former approach,
optimum variable equalizers cannot be developed concurrently with
the repeaters. In the new approach, the deviation equalizer is used to
optimally “marry” the repeater design error and the equalizers. The
development time of the fixed deviation equalizer is very short com-
pared to the repeaters and variable equalizers.

As indicated in the previous section, gain deviations of the line
repeaters resulting from manufacturing differences from repeater to
repeater and differences in cable loss from section to section, taken
together, result in a static misalignment of a line section. Since there
are a large number of contributors to this static misalignment, mis-
alignment of any line section as a function of frequency is extremely
difficult, if not impossible, to predict a priort. An equalizer designed
to compensate for this misalignment must therefore be able to accom-
modate a wide range of shapes. This implies an adjustable equalizer
with considerable flexibility.

On the other hand, the static misalignment of any line section will
be a very slowly varying function of time. These considerations led
to the design of manually adjusted equalizers that are adjusted upon
initial installation and, at very infrequent intervals thereafter, gov-
erned by such factors as route growth and line equipment replacement.

In-service versus out-of-service adjustment is another important
consideration affecting adjustable equalizer design. While in-service
equalization is preferable from an operational standpoint, the con-
comitant constraints imposed on the location of equalizer adjustment
tones in the frequency spectrum make the achievement of an optimum
equalizer setting difficult. Since the equalizers will be adjusted in-
frequently, however, out-of-service adjustment imposed no system
penalty and, in fact, is considered necessary to achieve the objective
of +0.4 dB in a switching section.

In wideband coaxial systems, two types of equalizers are commonly
used for amplitude equalization—transversal equalizers’ and Bode

equalizers.’ After extensive studies conducted early in the L5 develop-
ment, Bode equalizers were selected primarily because the algorithm
for adjusting the equalizer setting to match the shape of the line is
simpler and converges more rapidly to the optimum settings than that
for transversal equalizers. This is an important consideration. Achiev-
ing the 4000-mile equalization objective depends to a large degree on
the ability of the telephone company craftsperson to achieve the
optimum equalizer settings within the individual line sections. There-
fore, it is a distinct advantage to have a simple algorithm. Furthermore,
a simple algorithm is much more amenable to mechanization. Mecha-
nization, in turn, insures the achievement of optimum equalization.
This is discussed in detail later.

In the L5 system, two manually adjustable equalizers, the El and
E2 equalizers, are provided to compensate for the residual static
misalignment of the Ld line sections.

The El equalizer contains ten broad shapes spaced across the L5
frequency band from 1.6 to 66 MHz plus one very narrow shape
centered at the L5 temperature pilot location. They are shown in
Fig. 15. In each L5 section are three El equalizers, one in the trans-
mitting main station, one in the receiving main station, and a third
at the equalizer manhole. From a functional standpoint, the three E1
equalizers are identical. The manhole E1 equalizer differs from the
others in that it is packaged to fit in a manhole apparatus case and
also in that it derives its power from the de current on the center
conductor of the coaxial cable. The primary function of the El equal-
izer is to provide relatively coarse equalization of the L5 line and to
contain the message signal within relatively narrow levels as it tra-
verses the line so that excessive noise penalties will not accrue.

The nominal insertion gain of the El equalizer—that is, the gain
of the equalizer when all bumps are in their reference state—is 0 dB.
The gain or loss around zero of each bump can be set independently
of any other bump by means of a manual control on the front panel
of the equalizer. Note that, with the exception of the pilot bump,
each bump shape has a zero crossover close to the center frequency
of the bump immediately preceding it. Taken collectively, we refer
to this property of the El bumps as being one-way orthogonal with
respect to their center frequency. The significance of this property
is that, if the required gain setting of bump 7 is determined by a
measurement of the line with only bumps 7 — j,7 = 1, 2, ---,; 7-1

previously set, then the required gain will not be altered by the
subsequent setting of bumps 7+ j, 7 = 1, 2, ---, N —7 (where N
is the total number of bumps in the equalizer). It is this property that
allows rapid convergence of the equalizing algorithm.

The pilot bump has a very specific purpose. The transmission level
layout of the L5 line is such that the absolute power level of the tem-
perature pilot at the input to the El equalizer is always the correct
level. If the E1 equalizer were to change the pilot level (by insertion
of gain or loss by bump number 7, shown in Fig. 15, for instance),
then the regulating repeaters in the line following the El equalizer
would act to correct the change with Vf shape. In general, this would
prevent the equalization from converging and the line could not be
equalized. The purpose of the pilot bump, then, is to maintain the
pilot leaving the E1 equalizer at its correct transmission level regardless
of any other gain setting within the equalizer. The pilot equalizer is
narrow enough so that it does not affect the equalization of the message
band surrounding the pilot frequency.

The E1 equalizer is realized by a series of four amplifiers alternately
interconnected with four passive networks, as depicted in Fig. 16. Two
networks are series double-bump Bode networks and the third is a
triple-bump Bode network. The theory of Bode network design and
the realization of the networks are discussed in a companion paper.®
The amplifiers in the E1 equalizer serve several purposes. First, the
gain of the amplifier makes up for the flat loss of the networks so that
the overall insertion loss of the equalizer is 0 dB. Second, they provide
isolation between the Bode networks so that the return loss interaction
inherent in these networks is minimized. Finally, each amplifier
provides one bump shape by the inclusion of a Bode network in its
feedback path. A typical amplifier schematic is shown in Fig. 17.

The most difficult design problem in the El (and E2) equalizers
was the achievement of the overall frequency response. The gain or
loss of each amplifier or network comprising the equalizers is adjusted
during its manufacture to be within +0.05 dB of its respective nominal
value. The requirement on the overall equalizer is that, in its reference
state, the gain of the equalizer should be within +0.1 dB of zero when
all the Bode networks are in their flat condition. This has been achieved
by providing a trimming network adjusted during the assembly of
the El equalizer to trim the characteristic of the El to within the
desired requirements. Because the network and amplifier gains are
tightly controlled, the trim equalizer is a simple network containing
only a flat and a slope term.

AMPLI- EQUALIZER 2. AMPLI- EQUALIZER 3 AMPLI- EQUALIZER 1) = AMPLI-
FIER 4 FIER 3 FIER 2 FIER 1

The E2 equalizer is similar in design to the El equalizer. It contains
18 narrow shapes spaced across the 1.6- to 66-MHz spectrum. E2
equalizers are located in main stations at both the transmit and receive
ends of a power-feed section. The location of the E2 bumps are indi-
cated in Fig. 18. The primary function of the E2 equalizer is to provide
the fine-grained equalization of the L5 signal so that, coupled with the
E1 equalizer, the overall equalized response of a switching section is
within +0.4 dB of zero from 1.6 to 66 MHz.

The E2 equalizer contains seven amplifiers alternately intercon-
nected with seven networks. Six networks are series double-bump
Bode networks. The seventh network is a trim equalizer similar to
the one in the El equalizer. Six amplifiers contain Bode networks in
their feedback paths, while the seventh is a flat-gain amplifier. Salient
features of the El and E2 equalizers are shown in Table I.

Frequency Response (MHz) 1.6 to 66 1.6 to 66
Nominal Gain (dB) 0.0+0.1 0.0 + 0.1
Input Return Loss (dB) >30 >30
Output Return Loss (dB) >30 >30
Number of Shapes 10 Broadband 18

+1.0 Pilot Adjustment
Number of Amplifiers 4 7
Number of Networks 4 7
Full Load Output Power (Single +6.5 +4.5

The 4000-mile equalization objective for the L5 system is a very
difficult objective to meet. Therefore, great attention has been given
to the selection of the Bode network shapes used in the El and E2
equalizers and to the determination of the best equalization algorithm
so that optimal equalization is achieved from section to section.

From a transmission viewpoint, we can consider the El and E2
equalizers as a single equalizer with 28 Bode networks whose frequency
response, HQL(f), is given by

where f indicates the frequency and g,B;(f) represent the gain and
frequency response of the kth Bode network, respectively.

If M(f) represents the misalignment to be equalized, the residual
error, H(f), will be, after equalization,

The purpose of the equalization algorithm is to minimize a certain
measure of E(f) in eq. (3) over the frequency range of interest. The
ideal condition would be to make E(f) equal to 0 over the entire
frequency range by proper selection of B,(f) and gx. In reality, how-
ever, this is not possible. The L5 equalization strategy is to minimize
the mean-squared-error (MsE) function of E(f), which is defined as

The optimum set of gains, g;, which results in the minimum MsE
can be found if the gradient, G;, of the msE with respect to each gain,
gz, becomes zero.?°! That is,

The MsE algorithm given in Ref. 10 solves eq. (7) with G = 0 by
the steepest descent method. This may be readily implemented in an
automatic equalizer control circuit, but not in the manually adjusted
equalizers. The steepest descent algorithm requires simultaneous
adjustment of all the gain settings. An equalizer adjustment algorithm
based on the Gauss-Seidel iterative method discussed in Ref. 11 is
suitable and hence adapted for the manual El and E2 equalizer
adjustment. This algorithm calls for the following equalizer gain
adjustment procedure:

(z) Adjust the first gain, g1, until its corresponding gradient Gi = 0.
(72) Repeat for the second gain, gz, through the last one, g2s, until
corresponding gradients become zero, thus completing one
iteration.
(177) Repeat the above steps.

So far, it has been assumed that the gradient G, can be continuously
calculated during the adjustment of the kth Bode network gain, gy.

One approach is to compute the gradient by eq. (5). The gradient is
obtained by cross-correlating the equalizer shapes B:(f) with error
E(f). This procedure requires continuous error information over the
frequency range of interest and results in a rather complex hardware
realization.

It is shown in Ref. 10, however, that the gradient of the msE with
respect to a particular network gain can be very closely approximated
by the following simple relationship :

where E(f) is the channel residual error defined in eq. (3), fz2 is the
center frequency of the Bode network B;(f), and fii and f,3 are lower
and upper side frequencies of B,(f) such that

Bi(fis) = Bu(fis) = 2Be( fre).
If Bi(fi2) is normalized and its value is 1, eq. (9) becomes simply
Gi, = 2E (fia) + E (fiz) + Eas). (10)

In other words, by sampling the channel error at three frequency
points and by properly weighting these errors, we may approximate
the necessary gradient information. This approach is called the
simplified msE algorithm.

In Ref. 11, it is shown that a further approximation has resulted
in a very simple way of computing the gradient:

That is, the gradient is approximated by sampling the channel error
at a single frequency (usually the center frequency of the Bode net-
work). An equalizer adjustment method with the gradient obtained
as in eq. (11) has been referred to as the zero forcing (zr) algorithm,
and has been used, for example, in the L4 line equalization. When
the set of shapes B;, are near-orthogonal, the zF algorithm results in a
rapid convergence to the optimum gain settings.

Extensive computer simulation and field testing have been conducted
to compare the effectiveness of the three adjustment algorithms in
which the gradients have been calculated according to eqs. (5), (10),
and (11). The results are shown in Fig. 19, which indicates that there
is considerable improvement in the simplified Ms& algorithm over the
zF algorithm. However, little improvement is achieved when the
gradients are calculated according to the more complex relationship

in (5). In the L5 equalization plan, the simplified sr algorithm has
been implemented with the aid of a unit called the equalizer adjust-
ment unit (BAU).

In the equalization process, the BAv is used in conjunction with the
90-type transmission measuring set composed of the 90G oscillator,
90H detector, and 90F digital control unit,!2 which is located in every
L5 main station.

For equalization, two separate EAU’s and transmission sets are
required, one at the transmitting station and one at the receiving
station (see Fig. 20). During equalization, each EAv is set by its
operator for a particular Bode network, say B;,, and the Eau outputs
the appropriate control information to the 90 set for that network.
At the transmitting station, the command portion of the EAU instructs
the 90G to generate specified levels and proper frequencies corre-
sponding to the network to be adjusted. In the zF mode, the EAU
causes the 90G to generate only frequency fi2, which corresponds to

MANUAL CONTROL MANUAL ‘CONTROL MANUAL CONTROL AUTOMATIC
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t 1 t
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90F __J to 90F
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EAU EAU.

the center frequency of B,. If the MsE mode is selected at the Eau,
the 90G will generate sequentially frequencies fi, fizz, and fi3 as
defined in eq. (10). These frequencies are then transmitted on the
Ld line.

At the receiving location, the EAv initially establishes synchroniza-
tion between the transmitting 90G and the receiving 90H. The 90H
then receives these signals, measures the received level, and transmits
the resulting level information to the arithmetic portion of the Eau.
There it is processed and the resultant gradient information is displayed
to the operator in numeric form. The displayed gradient is calculated
by either eq. (9) or (11), depending on whether the mss or zF mode,
respectively, has been selected by a front panel switch located on
the BAU.

In the equalization procedure, since the gradient information is
displayed only at the receiving end of the line, the adjustment is under
the control of the operator at the receiving station. The receiving
operator relays the correct gain setting for each equalizer bump to
the operator at the transmitting station and to the craftsperson at the
manhole equalizing repeater location. To achieve the proper pre- and
postequalization during the adjustment of the E1 equalizer, one-half
the error is compensated for by the equalizer in the manhole and one-
quarter of the error is compensated for in both the transmitting and
recelving main-station equalizers. In the case of the E2 equalizer,
one-half the correction is inserted at both the transmitting and re-
celving main stations. The ultimate objective is to adjust all the equal-
izer shapes so that the resultant gradient, as displayed on the Bau, is
zero for all shapes.

In the equalization procedure, the zF algorithm is first used because
of its rapid convergence to the near-optimum gain settings. The MsE
algorithm is then used to ‘“‘fine tune” the equalization. The iterative
procedure is as follows:

(22) Select El equalizer, shape B,. Adjust the equalizers (one-half
the correction in the manhole equalizer, one-quarter in both
transmitting and receiving main stations) until the gradient
of the error is less than 0.05 dB.

(tv) Select E2 equalizer, shape Bi. Adjust the equalizers (one-half
correction in both transmitting and receiving main stations)
until the gradient of the error is less than 0.05 dB.

(v) Repeat (iv) for E2 shapes By through Bos.
(vt) Select the msE mode of gradient calculation.
(vit) Iterate steps (77) through (v) until the gradient is within +0.05
dB of zero for all shapes.

In practice, convergence of the iteration is quite rapid so that usually
two or three iterations are sufficient.

In both the L4 and L5 systems, the coaxial cable is buried at a
depth of four feet. The change in cable loss resulting from a +20°F
temperature change, for power-feed spans approaching 75 miles in
length, is +52 dB at 66 MHz. This is the magnitude of the equalizing
function of the regulating repeaters at the high end of the L5 frequency
spectrum.

The strategy in providing compensation for this seasonal dynamic
change in loss is focused at minimizing the signal-to-noise impairment
resulting from misalignment caused by the temperature effect. In
the L5 system, the distance between regulating repeaters is a maximum
of seven miles. With this bound established, the misalignment resulting
from cable temperature changes is reduced from +52 dB to +4.8 dB.

Fig. 21—Regulating section—66-MHz misalignment at the regulating repeater
output and input owing to cable temperature variation.

When both pre- and post-correction are provided, the misalignment
is further reduced to +2.4 dB as shown in Fig. 21.

A second effect that the regulating repeater must compensate for
is not directly related to temperature, but more to distance. Since the
basic repeater is a fixed gain amplifier designed to equalize the loss
of one mile of cable at 55°F, adjustments are made in the nominal
spacing of the repeater manholes when cable is placed in areas having
other than a mean annual temperature of 55°. In warmer areas, the
manholes are placed at intervals shorter than 1 mile; in colder areas,
placement intervals are greater than 1 mile. A 5-degree error in mean
annual temperature estimation for system design results in 2158 feet
of cumulative error in manhole placement over 75 miles. This amounts
to 18.1 dB of misalignment at 66 MHz or, as in the case of the L5
system, 1.2 dB in a 7-mile regulating section. The loss relationship as
defined in (1) is predominantly Vf.

There is a third source of misalignment. One inevitable situation
that plagues those who engineer cable systems is the inability to place
a manhole at the locations dictated by the nominal spacing. Right-of-
way procurement, convenient access, terrain considerations, and
densely populated and urban areas all contribute to a deviation from
nominal spacing. The accumulation of these deviations within a
regulating section results in additional misalignment that is accommo-
dated by the line-build-out networks in 0.1-mile increments, and the
remainder is equalized by the closed-loop regulating action of the
postregulator described in the next section. Therefore, equalization of
cumulative spacing deviations of +0.05 mile, equivalent to +1.6 dB
at 66 MHz, is another objective of the regulating repeater.

Other sources of misalignment, such as the deviation in basic
repeater gain from the nominal gain, the uncertainty in placed cable
lengths, and others, are estimated to be +0.4 dB. Table II lists the
misalignments and establishes the total range required of both the
post- and preregulator circuits at 66 MHz for 7 miles.

Table Il— Misalignments allocated to the regulating repeater
Cause Magnitude
Change in cable loss caused by temperature variation (+20°F) +4.8 dB
Cumulative length error (+0.038 mi) +1.2 dB
Cumulative spacing deviation (£0.05 mi) +1.6 dB
Other +0.4 dB
Total +8.0 dB

The preregulator automatically accommodates for +2.4 dB of the
cable loss variation resulting from temperature changes, and the post-
regulator must accommodate the remaining -+-5.6 dB. This effect is
illustrated in Fig. 22.

This suggests a range requirement of 11.2 dB for the postregulator.
However, because of circuit realization limitations, only 8.7 dB was
attainable in either a post- or preregulator network while still meeting
a broadband objective of less than +0.015 dB per dB of tracking
error. It was found that line acceptance procedures overcome this
limitation if the uncertainties and deviations accumulate in such a
way that the range of the regulating repeater is exceeded. These
procedures locate the excess variations which are then compensated
for with either reassignment of line-build-out networks or readjustment
of preregulators.

(7) To equalize the dynamic temperature-dependent term of the
cable equation in a pre- and postequalization strategy.

(it) To accommodate automatically the uncertainties and cumu-
lative spacing deviation in a postequalization strategy.

This total function is aimed at minimizing the signal-to-noise
impairment by maintaining the distributed misalignment at small
magnitudes.

Fig. 22—Regulating section—66-MHz misalignment at the regulating repeater
output and input resulting from cable temperature variation and other sources of
misalignment.

The regulating repeater consists of those circuits that comprise a
basic repeater plus additional circuits to perform the pre- and post-
regulating functions. These additional circuits are located between
the preamplifier and power amplifier as shown in Fig. 23.

It has been established that the misalignments allocated to the
regulating repeater are predominantly losses or gains that vary with
frequency in a Vf manner because of temperature and length. There-
fore, the equalization networks must also vary as Vf. The misalignment
allocated to the preregulator is one-half the variation resulting from
temperature. Control of the preregulator is established by sensing earth
temperature in the vicinity of the manhole. To establish proper track-
ing of the cable temperature, the sensor must be placed in an earth
environment similar to that in which the cable is buried. The depth
of the sensor is particularly important. Errors in depth significantly
affect the amplitude and delay of earth temperature variations with
respect to the cable.

Control of the postregulator is established by sensing the level of a
single frequency, designated the temperature pilot, which is applied
at the transmitting end of the cable. This method of control has the
advantage of automatically correcting for all misalignments of this
pilot. Therefore, the remaining half of the temperature effect on cable,
cumulative spacing deviations, and uncertainties in cable length and
design temperature are equalized in the postregulator. However, pilot
control may be troublesome if the losses incurred are not Vf, since
the correct equalization shape will not be provided. In the L5 system,
this difficulty is avoided by the proper selection of the pilot frequency
and the attention given to the basic repeaters and to the fixed and
adjustable equalizers at this frequency. Although maximum sensi-
tivity to cable temperatures occur at the high end of the L5 spectrum,
the pilot was chosen to be at 42.880 MHz. The reasons for this choice
are

(i) The temperature coefficient of the basic repeater (not vf)
could be made to approach 0 at this frequency, but is a maxi-
mum at higher frequencies.

(771) Return loss is better controlled and therefore interactive
effects are minimized.

(ziz) This frequency is in the guard band between the second and
third jumbogroups.

a ee eee ee ZERO LOSS AT NOMINAL
T TEMPERATURE AND MANHOLE SPACING ——————————~——— 7
TEST PORT
| PHASE-SHAPING
DEVIATION BUFFERED BUFFERED
EQUALIZER YF NETWORK Yf NETWORK

FROM
INPUT OUTPUT
DC GAIN WITH @EROM
THERMISTOR
COMPENSATING ihe TEMPERATURE
ZERO AND SENSOR
DRIVER NONLINEAR
MAPPING CIRCUITS
DRIVER
rn AND DRIVE
BAND AMPLIFIER—
PILOT DETECTOR

Fig. 23—Additional circuits that perform the pre- and postregulating functions.

Each pre- or postregulator contains identical Bode equalizers (vf
networks?) with loss adjustment controlled by varying a single resistive
element. This element is an indirectly heated thermistor with the
heater winding electrically isolated from the thermally controlled
resistor.

Gain is provided in each regulator to compensate for the loss in the
Bode equalizer and the associated networks by single-stage thin-film
hybrid integrated-circuit amplifiers. Two amplifiers and the Bode
equalizer are packaged to form the Vf network. The amplifiers also
provide the proper terminating impedances for both ports of the
equalizer to achieve an accuracy that is better than 1 percent in
matching the loss variation which is Vf. The type of transistor used in
the basic repeater circuits is also used in the single-stage amplifier to
achieve the noise figure and linearity requirements.

One critical aspect in the design of the preregulator control circuitry
is to accurately map the nonlinear resistance-versus-temperature
function of the buried ground temperature sensor into a linear loss
(in dB) function. In addition, provision had to be made to control the
sensitivity of the function since regulating sections can vary signifi-
cantly in length, and to adjust for the mean annual temperatures since
the same sensor is placed in all climatic areas. The response of the
temperature sensor is shown in Fig. 24. The range expected is +20°F

The nonlinear mapping was accomplished through the use of
operational amplifiers. With the use of negative feedback and scaling,
the control circuit provides a temperature tracking function that is
accurate to better than 2°F. The mean annual adjustment is made
accessible so that readjustments may be made safely in the manhole
when the repeater is installed and powered, should line acceptance
procedures indicate that the postregulator range is being exceeded.

In the postregulator, the temperature pilot is selected from the
message band with a crystal filter, amplified, and converted to a direct
current in a peak detector. Comparison of this direct current signal to a
reference results in an error signal that is amplified and applied to the
heater of the thermistor in the Bode equalizing network. The heater
sets the thermistor resistance in the Bode network and the pilot level
is maintained constant at the output for changes in level at the input.

The goal in this control circuitry is to achieve a reasonable response
time while maintaining a well-damped, nonenhanced system of many
postregulators in tandem.

A system of tandem regulators may be modeled as shown in Fig. 25.
In this figure,

G(s) = the control loop transfer function of a single regulator, where
sis a Laplace transformed variable.

The normal expected response of this system is to have H,; vary
seasonally while tracking each cable section loss change that results
from the cable temperature change. This variation is extremely slow
(one cycle per year) and, since G(s) is large for such a slow variation,
the envelope gain of the pilot is nearly zero while the message envelope
is properly tracking (A = Vjfm/fp) the pilot envelope to maintain the
levels constant at the output of each regulator. Therefore, each
regulator compensates for its preceding section of cable.

For very fast variations in the pilot envelope, the envelope gain to
the pilot is unity, and the message envelope gain is nearly zero.
Message channels, therefore, are not affected by the faster per-
turbations.

When G(s) has a single pole, no gain enhancement (£,./E»; > 1)
will occur, and minimum overshoot to transients will result. When
G(s) has more than a single pole, gain enhancement will occur where
G(s) approaches unity, and the system will not be critically damped
to transients.

In the postregulator, the loop response contains several poles. These
poles are a result of the indirectly heated thermistor. A plot of the
frequency response of the loop gain is shown in Fig. 26.

To restrict the gain enhancement to a tolerable magnitude, the
response is compensated for by using an operational amplifier to
provide a pole cancelling zero in the vicinity of 2 Hz. This results in
an enhancement response as shown in Figs. 27 and 28 where computed
and measured data are compared and in a transient response as shown
in Fig. 29.

One final aspect of the circuitry pertains to a transfer arrangement
in the event of pilot failure. The loss of the temperature pilot initiates
a switch to the spare line. This pilot loss could also cause the envelope-
controlled regulators to go to an extreme gain condition. To prevent
this high gain from occurring, heater control of the postregulator is
transferred from envelope control to preregulator control. Under these
circumstances, those misalignments allocated to the postregulator
are no longer equalized. However, in an L5 switching section (150

Fig. 27—Computed pilot envelope response for 80 regulators in tandem with and
without thermistor pole cancellation. Measured data points are compared.

Fig. 28—Computed channel envelope response for 80 regulators in tandem with and
without thermistor pole cancellation.

Fig. 29—Transient response of 80 regulators in tandem. Upper trace is the input
excitation of 0.8 dB.

miles), the misalignments owing to cumulative spacing deviation and
uncertainties tend to be both positive and negative, and hence there
is some cancellation. The net result is that gain is maintained near
normal. Also, regulator response time is fast so recovery from typical
no-pilot misalignments of 6 dB is complete in less than 5 seconds for a
switching section containing 25 regulators.

The networks in the transmission path of the regulating repeater
contribute to the noise and linear performance of the repeater. These
parameters are listed and compared to the same parameters of the
basic repeater in Table III.

The final level of equalization in the L5 system is performed by an
automatic pilot controlled equalizer which equalizes for a number of
dynamic effects. The primary effect is that associated with the small
but significant change in the gain of the L5 line repeaters because of
seasonal temperature changes. Repeaters located in manholes are
subject to ambient temperature changes roughly equivalent to the
+20°F temperature change of the coaxial cable itself. Line repeaters
having an average temperature coefficient of less than 0.002 dB per
degree Fahrenheit at 66 MHz introduce less than 3-dB change in line
gain over a 75-mile power-feed section. The variation of this deviation
with frequency is predominantly non-Vf and is therefore not compen-
sated for by the regulating repeaters.

In addition, as discussed in the previous section, a tracking error is
associated with the inability of the regulating repeaters to accurately
match the change in loss of the cable as a function of frequency and
temperature. Over a 75-mile power-feed section, deviations of up to
1.6 dB at 66 MHz can be expected.

While seasonal temperature variations have the main effect on
line gain change, there are other second-order, longer term effects
such as component drift, aging, and increases in the manhole ambient
temperature as the number of equipped coaxial lines in a route is
increased. All these taken together represent the total dynamic
deviation of the Ld line and are equalized by the E3 dynamic equalizer.

The realization of the E3 equalizer has been a function of a number
of system considerations. In previous sections, it was pointed out that
the static deviation, being a result of statistical manufacturing devia-
tions of many components, was impossible to predict a priori. Thus
the El and E2 equalizers are very flexible and have many degrees of
freedom to match a wide variety of shapes.

On the other hand, we expect the dynamic deviation to be much
better behaved. In previous L-carrier systems, the gain of the line
repeater as a function of the repeater temperature was a broad function
of frequency. Predictions based upon gain sensitivity analysis indicated
a similar behavior for the L5 repeatered line. We would expect then
that an equalizer designed to compensate for this effect would also be
a broad function of frequency with correspondingly fewer degrees of
freedom. Furthermore, this function could be measured a priori by
measuring the gain deviation of a sample of repeaters for various
ambient temperatures.

While the dynamic misalignment of the L5 line is a slowly varying
function of frequency, the overall equalization objective of the L5
line dictates that automatic equalization be employed rather than
manual equalization, as in the case of El and E2 equalizers. While
there are many ways we could conceive of providing automatic
equalization, the use of pilot tones in the composite message signal
and independent of the message signal itself allows a relatively simple
and reliable method of providing such equalization. Therefore, pilot
frequencies have been included above and below the message band as
well as between jumbogroups to provide for dynamic equalization.
The pilot between jumbogroup 2 and 3 is the same pilot used for
temperature regulation.

All these considerations have led to the E3 dynamic equalizer with
four degrees of freedom, each degree of freedom being automatically
controlled by a line pilot.

Unlike all other equalization in the L5 system, the E3 dynamic
equalization is provided only on a postequalization basis. There are two
reasons for this: In the first place, the expected misalignment over a 75-
mile power feed span is about 3 dB at 66 MHz. The noise penalty as-
sociated with postequalization only, under these conditions, is on the
order of 1 dB and has been included in the system margins. In addition,
any scheme to provide both pre- and postequalization on a dynamic
basis for all four pilots would be considerably more expensive and is
not economically justifiable. Thus, the E3 dynamic equalizer is located
in main stations at the receive end of each Ld power-feed section.

Figure 30 is a block diagram of the E3 dynamic equalizer. Basically,
it consists of four high-frequency networks through which the L5 line
signal passes. Each network has a variable gain shape such that the
combined frequency characteristic of all four networks closely matches
the expected residual dynamic misalignment of the L5 line. Three of
these shapes are Bode bumps, while the fourth shape is a flat function
of frequency (Fig. 31).

In Fig. 30, the first network is a series Bode network imbedded
between two flat-gain amplifiers. The amplifiers provide an overall
gain of 6.55 dB and provide good input and output return loss. Im-

bedded in the Bode network is a thermistor whose resistance is con-
trolled by the 42.880-MHz temperature pilot at the E3 equalizer
output through the control circuit feedback path. This resistance in
turn controls the gain of the Bode network such that the level of the
42.880-MHz pilot at the output of the equalizer is held at a fixed
reference level. .

The second network is a broadband variolosser that provides a flat
gain or loss around a nominal loss of 12 dB. The amount of gain or
loss is controlled by a thermistor in the variolosser whose resistance is
controlled by the level of the 20.992-MHz pilot at the output of the
equalizer.

The third network is a series Bode network similar to the first
network. In this network, the gain of the Bode network is determined
by the level of the 66.048-MHz pilot at the output of the equalizer.

The fourth network is a broadband feedback amplifier of nominal
14.0-dB gain with a Bode network in its feedback path. This network
also contains a thermistor whose resistance is controlled by the level
of the 2.976-MHz pilot at the equalizer output.

The E28 equalizer control circuits contain a pilot pick-off circuit
where the four E8 line pilots are separated from the message signal
and converted to four de voltage levels, each proportional to the
absolute power level of the respective pilot. The remainder of the
control circuits are digital circuits. The heart of each digital circuit
is a binary up-down counter coupled with a digital-to-analog converter
that acts as a digital integrator in the feedback loop. The system
works in the following fashion: If the pilot level at the output of the
equalizer deviates more than about 0.05 dB from nominal, it is detected
by a change in the de pilot voltage. Under this condition, the up-down
counter is allowed to count. This count is converted by the digital-to-
analog converter to a de current, which is then applied to the thermistor
in the corresponding network. This causes the network gain to change
in a direction to restore the output pilot level to its correct value.
When the pilot returns to within +0.05 dB of nominal, the counter is
stopped and the gain of the network is held at that level.

There are several advantages in using digital circuits in the feedback
loop when space and power consumption are not critical.

(2) With a digital integrator within the control loop, an equiv-
alently higher loop gain can be obtained than with an analog

integrator. The steady-state accuracy of the control loop is
determined primarily by the size of the quantizing steps, the
tracking between the digital-to-analog converter, and the
network control characteristic (input voltage versus network
gain in dB).

(it) If an abrupt change in pilot level or loss of pilot occurs because
of an abnormal line condition, the digital memory holds the
equalizing network control fixed.

(tit) The midrange gain of each equalizing network is precisely
defined by the state of the up-down binary counter so that the
network may be set at midrange with a local key or by remote
command.

(wv) The control loop dynamic response can be controlled by the
rate at which pulses are clocked into the up-down counter.

(v) The binary counter state can be monitored to indicate equalizer
gain and provide for control and alarm functions such as
locking the gain at any given state and giving a warning of
end of equalizing range.

If M(f, t) represents the time-varying misalignment to be equalized,
the residual error after equalization will be similar to eq. (3) and can
be expressed by

where f and ¢ indicate the frequency and time, g, and B; are the gain
and input-output relationship of the kth network, respectively, and
there are four adjustable networks in the equalizer.

The E83 equalizer is adjusted continuously in-service by sampling
channel misalignments at four pilot frequencies. In this way, four
pilot signals generate the four equalized channel errors. The network
gains are adjusted until the four errors become zero.

The block diagram shown in Fig. 30 can be represented by the
functional block diagram shown in Fig. 32. Figure 32 indicates that
the E8 equalizer is a multivariable system whose input and output
relationship can be expressed by the following equation.

Referring to Fig. 32, the gain g is obtained in the feedback loop and
expressed by

where G isa 4 X 4 control matrix, and W isa 4 X 4 weighting matrix.
G is an operator and determines the dynamic behavior of the four
feedback control loops and is expressed approximately by

where K and T are the constants andI isa 4 X 4 unity matrix. From
eqs. (13) and (14), the input-output relationship of the equalizer
becomes

All the main and power-feed stations along the L5 coaxial line are
equipped with E8 equalizers. Hence, any disturbance leaving one
equalizer will be propagated to the following equalizers whenever
those equalizers are controlled by the same pilot signals. If pilot
signals are blocked and reinserted every Nth station, then up to N
equalizers are effectively connected in tandem, and when the individual
feedback loop is designed, it is necessary to consider the effects on the
subsequent N — 1 equalizers.

The dynamic behavior of the E3 equalizer is mainly determined by
B and G in egs. (13) and (14), respectively. As shown in eq. (13), the
matrix B is determined by the network shapes in E3 and pilot fre-
quencies. When B and G are finally designed, a further improvement
may be obtained by a suitable choice of the weighting matrix W shown
in eq. (14).

In reality, the control loop transfer function, G, includes nonlinear
elements (e.g., thermistor), and the frequency domain approach to the
analysis and synthesis of the control loop to satisfy the transient
behavior becomes less accurate. A digital computer simulation in the
time domain was developed to predict the behavior of the N-tandem-
connected equalizers when the transfer function of the single control
loop, G, is known. The computer results were used to modify the
control loop transfer function to satisfy the system requirements of
the N-tandem-connected equalizers.

Figure 33 shows a transient response of four E3 equalizers connected
in tandem when input pilots are step-disturbed by 2 dB. Figure 34 is

similar to Fig. 33, but includes the transient effects of 40 regulating
repeaters within four power-feed sections (about 250 miles). These
results were measured in the first L5 installation.

The concept of a hierarchy of repeaters in which the basic repeater
is the fundamental building block was originated in the L4 system.
Experience with the L4 system has proven this concept to be sound,
and the L5 repeaters are similarly based. Design emphasis was focused
on the realization of an ultralinear low-noise repeater with a frequency
response that is consistently reproducible in manufacture. This

Fig. 36—Measured line noise (noise loading) extrapolated to 4000 miles (measured
length: 750 miles).

1. G. H. Duvall and L. M. Rackson, ‘L4 System: Coaxial Cable and Apparatus,”’
B.S.T.J., 48, No. 4 (April 1969), pp. 1065-1093.

2. F. A. D’Altroy, R. M. Jacobs, J. M. Nacci, and E. J. Panner, “L5 System:
Ultralinear Transistors,” B.S.T.J., this issue, pp. 2195-2202.

4. J. L. Garrison, L. P. Labbe, and C. C. Rock, “L4 System: Basic and Regulating
Repeaters,” B.S.T.J., 48, No. 4 (April 1969), pp. 841-889.

. R. M.-M. Chen, C. F. Hempstead, Y. L. Kuo, M. L. Liou, R. P. Snicer, and
E. D. Walsh, “5 System: Role of Computing and Precision Measurements,”
BS.T.J., this issue, pp. 2249-2267.

Copyright © 1974 American Telephone and Telegraph Company
Tue Brevi System TECHNICAL JOURNAL
Vol. 58, No. 10, December 1974
Printed in U.S.A.

The repeaters in a pair of LS coaxial-carrier transmission lines are
powered in a series loop with two dc-to-dc converters at each end of a power
span. The loop ts normally grounded at one end with the converter voltages
balanced to minimize the voltage to ground at the floating end. As with the
L4 system, automatic grounding of the floating point is provided to limit
the line voltage to ground and to enable the two lines to be powered in-
dependently during turnup and under trouble conditions. A new dc-to-dc
converter powered by a 140-V battery has been developed. It employs a
single power stage to deliver 910 mA at output levels up to 1150 V. A
dc-to-dc converter was also developed for operation on 24-V battery power
for use at stations where installation of a new 140-V battery plant is not
warranted. The dc-to-de converters employ hybrid, thin-film, integrated
circutts to achieve the needed level of precision without dependence on field
adjustments. They also employ many new automatic control features that
simplify operation of the four converters in a series loop and permit
turnup by remote control via the E2 status reporting and control system.
Another new feature is a spare converter with reversible polarity and a
decentralized high-voltage patching arrangement.

The repeaters and equalizers in the Ld coaxial-carrier transmission
system are spaced at approximately one-mile intervals along the cable
system. Dc-to-de converters operating on station battery power and
spaced up to 75 miles apart are employed to furnish a constant current
of 910 mA at voltages up to 1150 V dc over the coaxial line to the
series-connected repeaters and equalizers. This article describes the
line-power-feed system and equipment with emphasis on the features
that simplify (or automate) system operation and that minimize the

effect of foreign voltages which may be injected into the system
between stations.

The L5 repeatered lines, like those of the L4 system, are powered
in pairs using a series loop including two dc-to-de converters at each
end of a power-feed section (see Fig. 1).! For cables shorter than 37.5
miles, the pair of converters at one end is omitted. The system is
permanently grounded at one station and the output voltages of the

REPEATERS AND FOR LESS
EQUALIZERS THAN
37.5 MILES
7
1 ge
0 TO 1150
Vdc
dc INPUT
(STATION
BATTERY) de INPUT

SEPAR— > _ : > SEPAR~
TRANSMISSION SLTER FEA TRANSMISSION
SIGNAL SIGNAL
OUTPUT + 910 mA de | INPUT

four series-connected converters are balanced under normal conditions
such that the voltage is near zero between the floating and local-ground
points at the distant station. Therefore, the highest voltage to ground
is 1150 V and no path exists for the flow of direct current through
earth-metal interfaces between stations. This avoids electrolytic
‘corrosion at such interfaces under normal steady-state conditions.

Under certain abnormal conditions, grounding of the floating point
is desired. If, for example, a foreign longitudinal voltage appears
between grounded points in the system, grounding of the floating-
ground point may be needed to limit the peak voltage appearing
between the center conductors and outer conductors of the coaxial
cables. This arises from the tendency for such voltages to be super-
imposed on the output voltage of the converters connected to the
floating ground point. A second abnormal condition which warrants
grounding of the floating point is the appearance of a cable fault or a
converter failure that would reduce the current in the power-feed loop
below an adequate level. By detecting such a fault and automatically
grounding the floating point, normal current (and transmission) can be
maintained in one of the coaxial lines.

The principal electrical performance requirements that influenced
the design of the line-power-feed system are summarized in Table I.
The allowance for de earth potentials as high as 15 V per mile is
larger than that used in designing previous systems of this type.
Such earth potentials are caused by geomagnetic storms that ac-
company solar flares. The effects are largest on east-west routes at
high latitudes in areas with high soil resistivity. Large de earth po-

Table |— System requirements
Nominal line-current 910 mA
Loop voltage drop 120 to 4600 V
Line-current variation (normal +40 mA for combined variations of
limit) +100 V in loop drop, +2 V per

mile de earth potential, 10-50°C
ambient temperature, 120-152 V
converter dc input voltage.

Line-current variation (during +110 mA for earth potentials up to
unusual geomagnetic disturbances) +12 V/mile.

System-withstand capability (no DC earth potential up to +15
protective shutdowns or equip- V/mile. 60-Hz induction up to
ment damage) 1500 V rms for 0.1 s.

tentials occur infrequently and persist only for a few minutes in a
given locality. The design objectives of avoiding major transmission
impairment up to 12 V per mile, and avoiding protective power shut-
downs up to 15 V per mile are considered to be quite conservative,
inasmuch as earth potentials approaching these levels are expected to
occur only rarely and in limited areas of the United States.

High levels of longitudinal 60-Hz induction can be caused by large
unbalanced currents under fault conditions on major commercial
power lines which parallel the cable route. The “system-withstand”
capability of 1500 V is considered adequate for most applications.
Where routes have unusual exposures to high levels of 60-Hz induction
(i.e., more than 1500 V), additional measures, such as supplementary
cable shielding, must be considered.

A major factor affecting converter design was the desire for sim-
plicity of system operation. In particular, it was desired to be able to
turn up converters at unattended stations by remote control. This
implied a need to design the converters and protective ground circuits
to perform several functions automatically. These functions include:

(2) Soft start—To increase line current gradually over a period
of about 15 seconds.
(71) Converter voltage limiting—To avoid an excessive output
voltage from the first converter turned on.
(2) Automatic grounding—To ground the floating point during
turnup.
(iv) Automatic ungrounding—To unground the floating point
upon completion of turnup.
(v) Converter current limiting—To avoid producing excessive
current into a short circuit.
(vt) High-impedance shutdown—To prevent turnup into an open
circuit (avoids unnecessary personnel hazards).

To provide for faster power restoral after a protective shutdown
caused by an unforseeable momentary transient, the converter would
be designed to automatically make a single attempt to restart. To
maintain the converter and to provide for fast restoral of power in
the event of converter failure, a manually patchable spare converter
and test-load facility would be available at each power-feed station.

Simple in-service pushbutton tests would be employed to verify
that protective converter shutdown and grounding circuits are func-

tioning. The need for adjustments to the system in the field would be
minimized.

In summary, it was thought that the goal of high system reliability
would be best served by accepting moderate increases in the complexity
of the power-feed equipment to simplify its use in the field and permit
remote control.

The basic element in the converter is the power conversion section
which changes the de input voltage available from the station batteries
to a higher voltage that is controllable and is isolated from the input.
Power converters for the L5 system were designed to operate on 24-V
and 140-V battery plants. The power conversion technique for 24-V
input is similar to that developed for the L4 system,! and employs up
to five pulse-width-modulated power stages. Each power stage employs
power transistors switching at 20 kHz and delivers approximately
200 W at 230 V and 910 mA.

A new type of transistor-converter circuit?-* was developed for use
with the 140-V battery plants that will be used increasingly in the
Bell System.> The new converter circuit employs transistors rated at
200 V and 12 A and delivers approximately 1 kW from a single unit.
The new circuit, illustrated in Fig. 2, separates the pulse-width-control
function from the power-inversion function by employing a switching
regulator (Q1, D1) operating at 40 kHz on the input side of a bridge-
inverter circuit (Q2, Q3, Q4, Q5) that operates at 20 kHz. A principal
feature of this circuit is the presence of the inductor which averages
the voltage pulses at the input to the inverter rather than placing the
inductor in the conventional position at the output of the rectifier
bridge. This provides a number of advantages:

(z) Eliminates voltage and current surges that ordinarily
appear in the transistors and diodes as a result of rectifier-diode
recovery and/or transformer saturation.

(tv) Permits effective utilization of techniques to reduce switching
loss.

By using nonlinear networks to improve the switching locus of the
transistor, the switching losses were kept quite low even at a switching

frequency of 40 kHz using transistors with 0.75-us rise and fall times.
These techniques are illustrated for the switching regulator section
of the converter in Fig. 3. In Fig. 3a, the switching transistor operates
into a highly inductive load (L1) with the inductor voltage clamped
by fly-back diode D1. In the absence of the nonlinear loci-control
networks, A and B, the adverse switching loci shown in Figs. 3b and 3c
are obtained at turn-on and turn-off, respectively, of transistor Ql.
The peak and average switching losses in transistor Q1 are 1500 W
and 45 W, respectively. Inclusion of networks A and B yields the im-
provements in switching loci shown in Figs. 3b and 3c. This reduces
both the peak and average transistor switching losses by a factor of 10.
The total switching losses are reduced by a factor of 3, taking into
account the losses in the added resistor, Rl. The achieved conversion
efficiency of 90 percent (96 percent for the switching regulator alone)
is considered quite high at an operating frequency of 40 kHz.

The desired output voltage-current characteristic for a converter
intended to feed a line of maximum length is illustrated in Fig. 4. There
are three modes of regulation corresponding to the three linear segments
of the voltage-current characteristic.

Normal operation of the converter is in the middle section called
the impedance mode. The slope resistance of this section, 3370 ohms,

/ NETWORKS SWITCHING
LOCUS WITHOUT
NETWORKS
_ IMPROVED ee
/ SWITCHING ing
; LOCUS

Fig. 3—(a) Switching regulator with locus-control networks. (b) Q1 switching
locus at turn-on. (c) Q1 switching locus at turn-off.

is chosen to limit the variations in cable current to +110 mA in the
presence of de earth potentials up to 12 V per mile. The ‘‘level”’ of the
impedance segment is adjustable in the field to accommodate cables
of various lengths and to provide means for balancing the voltages
produced by the four converters in the series loop. None of the other
output characteristics, alarm, or protective shutdown levels require
adjustment in the field. Instead, the necessary precision (approxi-
mately +2 percent) is built into the equipment using precise integrated
voltage regulators, thin-film tantalum resistors, and integrated opera-
tional amplifiers.

At current levels above 1000 mA, the regulating mode automatically
changes to the current-limiting segment, where the slope resistance
increases to 24,500 ohms. This limits the output current to less than
1050 mA for any short-circuit fault on the line. Normal operation in a

OUTPUT CURRENT NORMAL
CURRENT REGULATING MODE - OPERATING POINT
t (24,500 OHMS) “4 (MAXIMUM
0 N 1000mA 7 i eang

mode with such a flat (i.e., high-impedance) slope would be undesirable
because of the difficulty of maintaining good voltage balancing be-
tween the four converters. At output-voltage levels above 1520 V,
the regulating mode automatically changes to the voltage-limiting
segment, where the slope resistance decreases to 43 ohms. This limits
the converter output voltage to 1550 V while power is being turned up.
Without this feature, the first converter turned on in the series loop
would deliver higher output voltage in attempting to supply power
to the complete loop. When the converter is turned on, the output
voltage of the converter is increased slowly in the open-loop control
mode over a period of 15 seconds to minimize the voltage and current
transients applied to the system and to enable the feedback regulator
to take control smoothly and restrict the output voltage and current
levels not to exceed the characteristic shown in Fig. 4.

The circuit implementation of the triple-mode regulator is indicated
in Fig. 5. The output voltage and current of the converters are sensed
using saturable reactors to achieve dc isolation. A negative de voltage
proportional to converter output voltage is presented at point a and a
negative dc voltage proportional to converter output current is pre-
sented at point b. These two voltages are compared to a reference
voltage in three different ways at the inputs to each of three “‘current-
summing” operational amplifiers. The outputs of the three operational
amplifiers are combined via diodes D1, D2, and D3 to control the

output of a 40-kHz pulse-width modulator which, in turn, controls the
output voltage and current of the dc-to-de converter. The three diodes
assure that only one operational amplifier (the one with the highest
level) controls the converter. In the current-limiting mode, operational
amplifier No. 2 is controlling. Note that the output-current level is
compared to a reference level at current summing point c. In the
voltage-limiting mode, operational amplifier No. 1 is controlling with
the converter output voltage compared to reference at current-summing
point d. In the impedance mode, operational amplifier No. 3 is in
control. In this case, a linear combination of the output voltage and
current signals is compared to reference at summing point e. The slope
of the converter output characteristic in the impedance mode is
determined by the ratio of the combining resistors R1 and R2.

VOLTAGE CURRENT
SENSING SENSING
SATURABLE SATURABLE
REACTOR REACTOR
=—_— \ 7
TO CABLE
R
E —
140 V Cc i
co)
T
TO GROUND
VOLTAGE ip & io

The feedback regulator circuit and the ramp generator (used for
soft starting) were realized as a hybrid integrated circuit using beam-
lead operational amplifiers, thin-film tanalum resistors, and ceramic-
chip capacitors. The resistors in the current-summing networks are
matched to 0.25 percent using an anodizing process.

General alarm and protective features were included in the converters
as follows:

Low-current alarm (—5%) High-current shutdown (+32%)
High-current alarm (+7%) High-voltage shutdown (1750 V)

A circuit is also employed to detect line-circuit fluctuations resulting
from an arcing fault in the repeatered line and to activate converter
shutdown. As a supplementary measure to protect personnel from
high-voltage hazards, a circuit is provided which prevents turn-up
of the converter if the coaxial line has an open circuit. These functions
are realized with three additional hybrid integrated circuits. The
precision of these circuits (+2 percent end of life) is such that no field
adjustments are needed. Means are provided for simple (pushbutton)
20, no-go in-service tests of these protection and alarm functions.

The function of the ground-protector circuit shown in Fig. 1 is to
automatically ground or unground the system’s floating-ground point
under appropriate conditions. When the converters are turned down,
the floating point is grounded. The ground is automatically opened
after two conditions are satisfied: (7) the sum of the magnitudes of the
two line currents exceeds 1500 mA and (77) the difference between
the magnitude of the two line currents is less than 34 mA. This assures
that all four converters have been energized and that the opening of
the ground connection will not produce a voltage at the floating
ground greater than 129 V.

The floating point is automatically grounded after a short delay if
the voltage exceeds 250 V to ground or instantaneously if the voltage
exceeds 800 V +15 percent to ground. Possible causes of voltages
sufficient to actuate the automatic grounding circuit are a converter
failure, a line fault, the appearance of an abnormal de earth potential
between stations, a nearby lightning strike, or abnormal 60-Hz
induction. As in the case of other automatic features, a pushbutton,

in-service test is provided to verify the proper functioning of these
features.

Vil. SYSTEM RESPONSE TO DC EARTH POTENTIALS BETWEEN
POWER-FEED STATIONS

As noted earlier in this paper, abnormal solar-flare activity can
cause the appearance of earth potentials up to 12 V per mile. Such an
earth potential appears as a voltage between station grounds as shown
in Fig. 6. The effect of earth potential will be discussed for the case of a
maximum-length system where the effect is maximum. Until the earth
potential reaches 3.33 V per mile (250 V for 75 miles), the earth
potential appears at the floating ground point. The line currents are
not affected since the system is grounded at only one point. As the
earth potential increases further, the threshold for the automatic
grounding circuit is exceeded and the system becomes grounded at
both stations. The earth potential now appears as an aiding voltage
in one loop (cable plus two converters) and as a bucking voltage in
the other loop (see Fig. 6). Thus, in a pair of cables, the cable current
increases in one cable and decreases in the other.

Upon taking into account the nonlinear output characteristic of
the converters and the major nonlinearities in the repeaters, the
overall effects of de earth potentials are shown in Fig. 7. The hysteresis
effect is the result of the margins purposely designed into the automatic

grounding and ungrounding circuits to avoid unstable operation. The
current variations in the cables are less than 110 mA for earth po-
tentials less than 12 V per mile (900 V in a 75-mile power-feed section).

The appearance of a burst of 1500-V, longitudinal, 60-Hz induction
near a power-feed station will cause a peak line voltage to ground of
2300 V and a peak line current of 2.2 A. If the 1500-V, 60-Hz induction
is closer to the center of a power-feed section, the peak voltage will
be less, but the peak current can increase to 3.5 A.

The action of the diodes in the converter output rectifiers is such
that the effect on the converter is similar to the application of an
alternating series of open and short circuits. Fast current-limiting
circuits are included in the converter to reduce the peak currents in
the switching transistors. A delay is included in the high-current
shutdown circuits to avoid an unwanted power shutdown during such
bursts of 60-Hz induction, which are expected to be less than 100 ms
in duration.

IX. HIGH-VOLTAGE PATCHING ARRANGEMENT FOR SPARE CONVERTER
AND TEST LOAD

A spare converter is provided for use in quickly restoring power to a
line in the event of failure of a regular line-feed converter. A manual

high-voltage patching arrangement is provided to permit connecting
the spare converter to substitute for the original converter for any
coaxial line and, at the same time, connecting the original converter
to the test load contained within the spare converter. To achieve this,
the output of the spare converter and the test load are multipled to
jacks in a protected area at the top of each converter, as shown in
simplified form in Fig. 8. There are two types of plugs shown in Fig. 8,
Pl and P2. The plugs are shown in their normal positions in Fig. 8.
Note that there is a Pl type plug for each converter normally associ-
ated with a particular coaxial line, but only one P2 type plug which is
normally associated with the spare converter. In the normal state,
each individual converter feeds a particular line and the spare con-
verter 1s connected to its internal test load. Under this condition, the
spare converter may be exercised into the test load to assure its
operability should it be needed to substitute for an individual con-
verter. The spare converter also provides a means for checking opera-
tion of the spare converter plug-in units at each station.

When an individual converter fails, the Pl plug from the failed
converter is manually interchanged with the P2 plug from the spare
converter. As can be seen from Fig. 8, the failed converter will connect
to the test load and the spare converter to the coaxial line. After the
failed converter has been repaired and tested, power is turned off on
the line and the patch plugs are restored to their normal positions.

The patching is somewhat more complex inasmuch as there are both
positive and negative converters for the individual lines and the
converter output polarity of the spare converter is reversible. The

REGULAR LINE FEED CONVERTERS SPARE CONVERTER
FEEDING INDIVIDUAL CABLES AND TEST LOAD
(USUALLY ONE PER STATION)

P1 plugs from positive converters are slightly different from P1 plugs
from negative converters and this difference is used to control the
output polarity of the spare converter when it is used to power a
coaxial line. There are safety interlock features built into the patch
plugs to assure that converters cannot be energized unless the plugs
are in valid positions and the covers over the patch panels are closed.
Major design objectives for the patching arrangement were to keep it
simple and safe for use by the telephone craftsperson in the field and to
facilitate the addition of converters without requiring a power shut-
down on all cables at a station.

The physical design of the line-power-feed equipment for the L5
system was based on the following factors:

(t) Two power sources (converters) and their associated ground
panel are always provided as a group to power a pair of re-
peatered lines (see Fig. 1), and thus can be manufactured and
subsequently installed in the field as a single entity.

(it) The two paired power sources (converters) have the same
output voltage magnitudes but different polarities.

(i2t) The required line-feed voltages can be predetermined and the
power-feed equipment can be ordered from the factory with
the proper output-voltage capability.

These factors resulted in a physical design which integrates two power
sources (converters) and a ground panel into a single structure (see
Fig. 9). While the structure has well-partitioned volumes for each
function—converters and ground panel—they are not distinct, separ-
able, physical entities.

The physical design of the line feed has been influenced, in detail, by
other factors. For instance, the L5 line-feed equipment had to be
compatible with L4 line-feed equipment in some respects to facilitate
station rearrangements when L4 lines were converted to L5. The
height and width had to be the same as in L4 or some rational sub-
multiple thereof. The depth of the L5 equipment was designed to be
approximately one-half the depth of the L4 power-feed equipment.

To facilitate service restoral and maintenance, replaceable plug-in
modules and a spare converter were included in the design. Personnel
protection was required because of the high-voltage potentials present.

Other design factors were the usual requirements for (2) means for
heat removal, (27) convenient cabling design, (777) human engineering
features, and (zv) esthetic considerations.

The line-power-feed equipment cabinet shown in Fig. 9 contains
two converters and a ground panel. High-voltage patching compart-
ments and a common area for battery input and alarm connections are
also provided. Figure 9 shows the 24-V-battery input design, which is
27 inches wide, 133 inches deep, and 7 feet high. The 140-V-battery
input version, designed for stations with the new 140-V de distribution
system shown in Fig. 10, is narrower. The same partitioning pattern
has been followed in the 140-V design. Three 140-V cabinets placed
side by side are equivalent to two 24-V-input designs and are thus
compatible with floor plans.

Normal operating controls are mounted on the front panel. No
other facilities for adjustments are provided within the equipment.
Trouble indicators are a lamp at the top of the cabinet, and a lamp for
each converter.

Figures 11 and 12 show the regular line-feed cabinets with their
doors open to illustrate the plug-in modules. The lower doors are inter-
locked to deactivate the equipment (one converter at a time) since
the high voltages are generated in this area. Access doors to other
high-voltage areas, including the patching area, are also interlocked
and protective covers are furnished.

The cabling plan is designed to minimize cable installation within
the cabinet. Field-connected cable terminates at the top of the cabinet.
The high-voltage outputs and the connections to the spare converter
are protected by flexible conduit and enclosed metal ducts.

Replaceable plug-in modules, which contain the critical electronic
circuits, are used to facilitate rapid field repair and manufacture. The
modules for a 24-V input converter are shown in Fig. 13. The active
electronic circuits for each converter, except for the power stages and
oscillator, are contained on five printed-wire boards which provide the
plant automatic features described earlier. Each board consists of a
double-sided epoxy-glass board assembly with a standard plastic face

plate. Connection to the unit is through printed-circuit contact fingers
which plug into connectors mounted on the housing for the five
boards. Each converter has an identical complement of five boards.
Although the boards were initially designed for application in the
24-V-input version, they were applied without change in the 140-V-
input version to simplify manufacture and repair. Other printed-wire-

board assemblies are component parts of the power stages, oscillators,
and ground panels.

The feedback regulation, alarm, and protective features incorporated
in the converters utilize four specially developed hybrid integrated

circuits shown in Fig. 14. The hybrid circuits include thin-film tantalum
resistors with precise ratios (0.25 percent initial) to provide long-term
stability and eliminate the need for adjustments over the anticipated
life of the equipment. The resistors, positioned in close proximity on a
ceramic substrate, are aged prior to final trim anodization to achieve

the required accuracy and stability. Additional advantages in circuit
performance are realized by including associated components on the
same substrate, which assures minimum-length paths for critical
interconnections. These additional components include beam-leaded
operational-amplifier chips, ceramic-chip capacitors, and beam-leaded
diodes. Stitch crossovers are also used for interconnections on the
substrate. Solder reflow is used to attach the capacitors and thermo-
compression bonding is used for the operational amplifiers and diodes

Fig. 15—Unencapsulated development model of feedback regulator with hybrid
integrated circuitry.

as well as the substrate lead frame. An early development model of
the feedback regulator hybrid, unencapsulated, is shown in Fig. 15.
The hybrid circuit is mounted to the printed-wire board by reflow
soldering the lead frame to presoldered terminal areas on the boards.
The use of hybrid integrated circuitry eliminates a significant number
of soldered connections, which would have been required in a discrete
realization and thus should enhance reliability.

Each converter (line and spare) has an associated high-voltage
patch compartment containing the patch plug and mating receptacle.
Access to the patch plug is gained by opening an interlocked hinged

panel, as shown in Fig. 16. A small window in the line-feed converter
panel is provided for visual observation of the patch status, i.e., to
determine whether the normal plug or the spare plug is in place in the
line-feed converter. A status lamp is provided at each line-feed con-

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verter and indicates whether the spare converter has been placed in
service as a substitute for a regular line feed or is available for
assignment.

The spare converter services a number of line-feed converters, the
quantity being a function of the acceptable system repair time. It
should be noted here that the transmission-system-protection switching
facility maintains service in the event of a converter failure. Arrange-
ments are available to provide a multiplicity of spare converter systems
for regular line feed within a single office if system-reliability require-
ments warrant.

Spare converters are available for —24-V and 140-V input voltages.
The cabinet sizes are identical to those of the regular line-feed
converters.

The precision obtained through the use of hybrid-integrated-circuit
technology permitted the inclusion of many automatic control features
in the power-supply system for the L5 repeaters. It is expected that
the resulting simplification of system operation and the minimal
number of adjustments required in the field will contribute significantly
to overall system reliability.

A. P. Walsak was responsible for physical design, and circuit de-
sign responsibilities were shared by R. Ostapiak, P. P. Untamo, R. E.
Schroeder, and J. R. Meszar.

2. E. T. Calkin and B. H. Hamilton, ‘‘Circuit Techniques for Improving the Switch-
ing Loci of Transistor Switches in Switching Regulators,’ IEEE Conf. Record
of 1972 Seventh Annual Meeting of IEEE Industry Applications Society,
pp. 477-484.

. Ibid., “A Conceptionally New Approach for Regulated de to de Converters
Employing Transistor Switches and Pulse Width Modulation,” pp. 485-494.

Copyright © 1974 American Telephone and Telegraph Company
THE Bey System TECHNICAL JOURNAL
Vol. 53, No. 10, December 1974
Printed in U.S.A.

A new system has been designed to provide protection switching capa-
bility for the LS repeatered line. This line-protection switching system,
called tpss-3, provides one standby line for protection of as many as 10
service carrying lines. Switching equipment ts located in terminal, ter-
minal main, and switching power-feed stations. Economic and reliability
considerations allow switching spans of up to 150 miles. Either tempera-
ture pilot deviations or excessive total signal energy initiates automatic
switching. Manual control of all switching functions 1s provided locally
and via telemetry.

A dedicated pcm signaling system maintains interstation switching
system coordination. Identical information ts transmitted in complemen-
tary form over two adjacent L5 channels. Parity and complementary
checking as well as automatic retransmission of failed codes enhances
error immunity.

The need for new switching capabilities and the desire for a modern
switching facility for the L5 Coaxial-Carrier Transmission System
have led to the development of a new line-protection switching system
called tpss-3. Several objectives were paramount at the beginning of
the new development. One was to design an interstation signaling
system that is relatively immune to both line noise and hits (short
interruptions of transmission), since a signaling error can cause a
service failure because of improper switch activation. Another objec-
tive was to provide a design that requires a minimum of effort for the
addition of new lines.

A newly installed coaxial-carrier system usually has only one or
two of the ultimate regular lines equipped and connected. Additional
lines are equipped and connected to the system as traffic growth
requires. Thus, the process of adding lines to a working system is
important. The upss-3 bay is factory-wired and tested for its full
capacity of 10 regular lines and one standby line. Equipping switching
capability for newly added regular lines on an L5 system requires only
replacing dummy plug-in modules with active modules in the Less-3
bay and adding a few control leads to the new line bays. No new intra-
bay wiring changes or connections are involved.

A new series of coaxial switches has been developed using diodes as
the switching elements. These switches were designed specifically to
comply with the L5 system bandwidth and modulation performance
standards.

A reliability analysis of estimated failure rates, mean time to failure,
and service outage was made for the Ld coaxial system. This included
estimates relating to the L5 line, main-station, and switching equip-
ment. These studies were then projected to estimate average outage
times for various types of switching sections through the use of sys-
tem models. In addition, the probability of cable damage has been
analyzed and is included in the overall system outage predictions. An
outage is a service loss, ie., a failure not remedied by protection
switching.

The results of this study show that the use of average switching
intervals of 120 miles, total system length of 4000 miles, and one pro-
tection coaxial line for 10 regular lines will result in adequately small
outage times. With this arrangement, service outage time because of
individual line failures is significantly less than outage time caused by
massive failures such as a man-made fracture of the buried cable.
Modification of switching section length and spare-to-regular ratio will
not reduce the latter outage time, which is controlling.

A broadband transmission system such as L5 is composed of two
basic elements: the office or main station equipment and the repeatered
line equipment. The main station equipment combines and separates
the system message content and performs functions such as power
feed, equalization, line switching, and fault location. The line equip-

ment provides the transmission equipment required to connect main
stations together. The line-protection switching system provides the
backup protection to guarantee transmission between main stations
when one of the regular lines fails. This is accomplished in L5 by
utilizing the LPss-3 equipment to control and cross-connect the main-
station line-connecting equipment to the standby lines provided in the
transmission medium. (In the coaxial cable, one unit coaxial is pro-
vided for standby or protection use in each direction of transmission,
since L5 repeaters are unidirectional.)

Figure 1 is a diagram of an L5 line-switching section. It has been
simplified to show only one regular line and the standby line for each
direction of transmission. The protection equipment is located within
the main stations at each end of the switching section (up to a maxi-
mum of 150 miles apart). Within the station, the protection equipment
is divided into two basic locations. The line failure detectors and
coaxial switches are located in the individual line bays, while the
common switching equipment is in the Lpss-38 bay. The inputs to the
detector circuits and the access of the interstation signaling commands
are derived from the line-connecting circuits.

The detector circuits are located in the switch initiator, Fig. 2.
These include both transmitting and receiving detectors. The receiving
detectors monitor the 42.880-MHz line pilot and the total system
average power. If the pilot deviates more than 5 dB from nominal for
more than 2.5 ms, a switch request is initiated (whether a switch takes
place depends on many conditions and is discussed in detail later). If
the average power exceeds its threshold, a switch request is issued,
followed 50 ms later by a termination request (as before, the results of
these requests depend on other conditions and are covered in detail
later).

Transmitting detectors monitor the 42.880-MHz pilot. Each time
that pilot exceeds the +5-dB limit, a 50-ms signal is generated in the
LPss-3 transmitting switch-control circuits, preventing a line switch
from taking place during that interval. The assumption is that the
failure has occurred in a previous switching section (the probability of
such is very high) and that section’s protection-switching facility will
remove the apparent problem. If both transmit and receive detectors
in a switching section persist in the failed state beyond a reasonable
switching interval, a switch is completed in that section to obtain the
benefit of the independent standby line temperature pilot source.

AA AA REGULAR AA AA
ee
H HYBRID TRANSFORMER MCS MESSAGE CUTOFF SWITCH
S| SWITCH INITIATOR LTS LINE TERMINATE SWITCH
TLS TRANSMIT LINE SWITCH DIR DIRECTOR SWITCH

Two basic types of coaxial switches are used in the protection of L5.
One is the director switch, a 1 X 11 solid-state switch without any
control of its own; it is a slave to the switches with which it operates.
One director is located at each end of the standby line (Fig. 1). Its
common port is connected to the line equipment, and its 11 other ports
connect to the line-connecting circuits via line switches. With no pro-
tection switch in force, the standby line-connecting circuits are con-
nected to the standby line. With a protection switch in force, the
directors route the message from the transmitting line-connecting
circuit for the failed line, through the standby line, and on to the
receiving line-connecting circuit for the regular line.

The remaining line switches are combinations of the basic solid-
state T structure. This structure uses two series elements and a shunt
element. In the pass condition, the series elements provide low loss and
low distortion, while the shunt element has high loss. The stop condi-
tion is just the opposite, with the series elements providing high inser-
tion loss to the signal and the shunt element acting as a short to ground
for any signal passing the first series element. The line switches are
composed of from two to four T’s. In Fig. 1, the T’s have been simpli-
fied to show only their normally released state. (The straight line across
the path indicates the pass condition of a T, while the X indicates the
stop condition.) The transmitting and receiving line switches connect

the line-connecting circuits to the standby line via the director switches.
As pointed out previously, the director switches are slaves to the line
switches. In fact, the solid-state elements of the director switches are
the final elements of the T’s of the line switch to which they connect.
Control of the director is by a de current from the line switch to the
director over the center conductor of the coaxial cable carrying the
message between them. Two other switches are used in protection
switching, one located at each end of the switching section. The trans-
mitting end of each line, both regular and standby, contains a message
cutoff switch. The function of this switch is to remove the message
load from the line, under controlled conditions, to permit line measure-
ment for special purposes—equalization and, if necessary under high
noise conditions, fault location. The line-terminating switch is located
at the receiving end of the section, immediately following the receiving
line switch. Its function (detailed in Section 4.9) is to prevent the
propagation of system overload conditions.

High noise immunity and low power dissipation are more important
characteristics than speed of operation in Lpss-3 circuitry. With this
in mind, saturated logic integrated circuits have been used in Lpss-3
designs. Timing circuits and NAND logic functions are provided by a
family of diode-transistor logic (pTL) circuits. Resistor-transistor logic
(RTL) circuits are used in the signaling system to provide Nor and
EXCLUSIVE OR logic in addition to clocked flip-flops for parallel-to-serial
and serial-to-parallel conversion. Set-reset flip-flops from both families
are used as memory elements throughout the system.

There are two varieties of L5 offices from an Lrss-3 point of view:
switching power-feed stations and terminal or terminal main stations.
The physical arrangement of equipment in the Lpss-3 bay is the same
for either type station, but the manner in which Lpss-3 interfaces with
the line equipment differs. The Lpss-3 bay is mounted in the same
aisle as its associated line bays. The reasons for choosing this approach
are twofold: it was possible to design one universal arrangement that
fits all applications, and the maintenance operation is simplified by
always keeping functionally associated equipment in the same aisle
alignment. In the terminal and terminal main stations, the Lrss-3
equipment is associated with a line section. In the switching power-feed
station, the Lrss-8 equipment is associated with one direction of

transmission in two line sections because the line bays are through-
transmission units. No message administration is performed at these
stations, so the optimum line arrangement is to have the receiving and
transmitting circuits all in the same bay.

Protection switching operations require coordinated actions at
diverse physical locations. Transfer of service from a regular to a
standby line, for example, requires properly sequenced operation of
line switches separated by up to 150 miles. This interstation coordina-
tion is maintained by the Less-3 signaling system, a self-contained sub-
system that generates and detects digitally encoded carrier signals.
The L5 system itself is the medium for transmittal of these signals;
the associated frequency band is blocked at each switching main
station to eliminate signaling interference among switching sections.

Thirty-nine digital code words are in active use within the system;
each has a specific purpose and consists of seven bits (e.g., 1100101).
The digital rate is 2 kilobits per second. When signaling is not taking
place, an alternating 1-0 pattern is transmitted to allow immediate
alarming whenever interstation signaling continuity is lost. The seven-
bit code word is preceded by two successive 1’s to mark the beginning
of the code word.

The transmitter generates an FsK signal with a 1 bit corresponding
to energy at 68.76 MHz and a 0 bit corresponding to energy at 68.78
MHz. The receiver separates and independently detects the two
channels. With no transmission or detection errors, the two channel
outputs are complementary.

Error-free detection is of prime importance. With this in mind, the
39 code words have been chosen to have even parity to allow detection
of errors resulting from the permutation of a single bit. Additional error
detection is made possible through complementarity checking of the
two channels for each bit of the code word. A transmission medium
disturbance must simultaneously permute a 0 to a 1 in one channel
and a 1 to a 0 in the other channel in at least two bit positions to cause
an erroneous receiver output.

Error detection at the receiver causes the incoming word to be re-
jected; no attempt is made at error correction. To prevent switching

system lock-up under this condition, the transmitter automatically
retransmits the code word every 15 ms until evidence is received that
the command has been properly decoded at the remote location. The
sequence is as follows: An originating command is transmitted to the
remote station to cause the change of state of a line switch. With
this action completed, the remote signaling transmitter generates an
answering command that is routed back to the originating location.
The proper decoding of the answering command stops the retrans-
mission of the originating command. At the remote location, cessation
of the incoming originating command is taken as evidence that the
decoding of the answering command has been successful, and the
retransmission of the answering command is stopped. Every signaling
operation consists of a round-trip operation as described above to
verify the completion of each originating command.

Special provision has been made to allow in-service exercising of
the signaling system to verify proper operation. Local or remote (via
telemetry) manual action causes the transmitter to generate a TEST
command, which is decoded at the far end of the switching section.
A TEST RECEIVED command is then returned, completing the round
trip and causing the signaling to return to idle. Failure of any involved
circuitry along the route will cause a TEST FAIL lamp to illuminate at
the originating point. This test can be performed at any time, even
with a protection switch in effect, and checks most of the circuitry
involved with signaling.

Hach set of up to 11 coaxial lines in a given transmission direction
has an associated signaling transmitter and an associated signaling
receiver. At the transmitting end, a switch signal distribution unit
controls the application of the switch signals to the transmitting
coaxial lines. The signals are normally introduced on to all outgoing
regular lines but, as explained in Section IV, the signals are sometimes
temporarily introduced on to only the standby line or one regular line.
The signaling receiver is electrically connected to only one line by
the signaling receiver switch unit, which normally provides connection
to the lowest-numbered ‘good’ regular line. In this case, ‘‘good”’
implies the line is in service (e.g., not manually switched out of service)
and not failed. Under certain conditions, the receiver is temporarily
connected to the standby line.

In this section, we examine the operations of Lpss-3 in detail through
the use of flow charts. Line switching, termination, and message cutoff
operations are discussed after additional preliminary concepts are
established.

A switching section consists of line switches and associated switching-
control circuits at both the transmitting and receiving ends of the
coaxial lines. The receiving Lpss-3 circuitry has primary control of all
switching actions in that switching section. The transmitting LPss-3
circuitry merely responds to directions from the receiving circuitry.
Switching activity, manual or automatic, is never initiated at the
transmitting end.

In upss-3 jargon, the term, “priority,” is associated with internal
LPss-3 operations, which establish the right for an operation to take
place. There are two types of priority. A failed regular coaxial line
attempts to establish switching priority as a first step in executing
an automatic switch. Any operation requiring the use of the signaling
system must obtain signaling priority before being given access to
the signaling transmitter. In both cases, priority avoids the system
confusion that would otherwise result when independent, nearly
simultaneous operations take place. The priority circuits in LPss-3
(Fig. 3) are N-input, N-output logic circuits designed so that a driven in-

Fig. 8—Priority circuit. Only one output (corresponding to a driven input) may
be operated regardless of the input time sequencing. Inhibit prevents new outputs,
cancel inhibits new or established outputs.

put will result in the corresponding output going to the operated state.
With multiple driven inputs, there will be only one operated output.

If the inhibit input is activated, no new output can become acti-
vated ; if the cancel input is activated, any previously activated output
is cancelled. Consider, for example, the case in which regular line No. 5
fails at approximately the same time as line No. 2. This will result in
the No. 2 and No. 5 inputs of the switching priority circuit being
driven. Depending on the input sequence, either the No. 2 or the No. 5
output will be driven, but both can never co-exist, even for short
intervals. Thus, only one line will obtain clearance to establish a
switch. At certain times (e.g., standby line failed) Lpss-3 will refuse
to grant new switching priority, but will not defeat established priority.
This causes the inhibit to be driven. Both new and established priority
may be defeated by the cancel input. Lock normal (Section 4.8) is
one such condition.

The major steps in accomplishing an automatic transfer of service
from a regular line to the standby line are diagrammed in Fig. 4. The
line bay switch initiator detects a line failure and automatically causes
an LPpss-3 switch request for that line. Switching priority will be
established if Lpss-3 can provide switching. Action will stop if another
line has switching priority, if the standby line is failed, or if lock
normal is in effect. Signaling priority is established next. If another
operation is in the process of using the signaling transmitter, another
delay is encountered until the transmitter is available. At that time,
a specific digital code word called the 1p (there are 10 such code words,
one for each regular line) is sent to the transmit end of the switching
section over all regular lines in that direction. The 1p is decoded by
the signaling receiver, causing the appropriate transmit switch to
operate. An Lpss-3 indication Tso (transmit switch operated) is
illuminated. The message is now being introduced to both the regular
and standby lines. Signaling priority is sought at the transmit end so
that the answering command verifier may be sent back to the receiving
end. This one command is used in common by all 10 lines. Receipt of
the verifier is taken as evidence that the transmit switch has operated
properly. The receive switch is operated, completing the switching
sequence. An Lpss-3 indication osa (out of service automatic) is
activated.

The route of the verifier command from transmitting to receiving
ends of the switching section deserves special attention. The verifier is

introduced into the transmit line-connecting circuit of the line to be
switched. The verifier then propagates through the operated transmit
switch on to the standby line. Meanwhile, at the same time that the
ID was sent out from the receiving end, the signaling receiver was
connected to the standby line in anticipation of the verifier returning
from the transmit end of that line. This routing gives an added measure
of confidence in the serviceability of the path to which the message is
to be transferred. If the verifier is not successful in traversing this path,
the receive switch will not operate.

Elapsed time from failure to switch complete is typically less than
12 milliseconds:

2.2 ms Signaling, verifier (zero length system)
0.8 ms Verifier propagation (150 miles)
0.3 ms Delay at receiving end

Completion of an automatic switch results in a minor office alarm,
since switching protection has been automatically used. If, for any
reason, a switch request does not result in a completed automatic
switch within 1 second, an Lpss-3 TF (total fail) indication and ac-
companying major office alarm result, since service is presumably lost.

In many cases, the 42.880-MHz pilot that is detected for switch
initiation traverses more than one switching section. The switch
blocking feature prevents several tandem sections from switching on a
single failure.

The temperature pilot level in the regular transmitting line-con-
necting circuits is detected by the switch initiator. When the pilot
exceeds limits, the initiator so informs the Lrss-3 transmitting switch
control circuitry. The transmitting switch is then inhibited from opera-
tion for the next 50 milliseconds. The receiving switch initiator circuits
at the next office will also detect the failure, causing an 1p to be sent
back to the transmit end. Since a block is in effect in the transmit
switch control circuitry, the transmit switch will not operate. Under
this condition, the BLOCK ON command is sent back to the receive end
of the switching section instead of the verifier. At the receive end, the
apparently failed line is deprived of switching priority for 100 ms.
During this interval, the switching system is free to execute other
operations. After the 100-ms interval, the involved line is allowed to
seek switching priority if it is still failed. Note that the section with
the actual failure is allowed to switch, since its transmitting detector
experiences no failure. Subsequent switching sections undergo a 12-ms
failure (typical) at both transmitting and recciving detectors and try
to switch, but are blocked. After the 100-ms lockout, the lines are no
longer failed because of the completed switch in the failed section, and
no further action results. If the switch does not complete in the failed
section, the next section will switch after the 100-ms interval.

When a failed line that has an automatic switch in force returns to
a nonfailed condition as indicated by the switch initiator, an automatic
switch release sequence starts. Figure 5 summarizes the operation.

For the first 30 seconds after the line has returned to normal, no
action takes place. If the line momentarily fails during this interval,
the full 30-second count is restarted. After the line has been nonfailed
for the full interval, the receiving switch is released. This transfers
service back to the regular line, since the operated transmit switch did
not remove the message from the regular line, but merely caused a
dual feed of message on the regular and standby lines. The remaining
steps are taken to clear the switching system for the next operation.
Signaling priority is obtained to send the RELEASE command to the

transmit end of the section, causing the transmitting switch to be
released. The release verifier is then sent back to the receiving switch
control circuitry to verify the successful release of the transmitting
switch. If the release verifier fails to return within 5 seconds of the
receiving switch being released, the transmit switch release fail (TsRF)
indicator is activated, and the minor office alarm is sounded in the
receiving office. With this condition in effect, the system will not
attempt to establish another protection switch, since a transmitting
switch may still be operated. The rsrr condition is cleared by the
successful completion of a manual release (Section 4.7).

Manual switching capability is provided by Lrss-3 so that nonfailed
regular L5 lines may be taken out of service for equalization, main-
tenance, or measurement. The steps taken to establish a manual
switch are identical to those summarized in Fig. 4 for automatic
switching, except that the action is started manually rather than
automatically, the resulting indication is osm (out of service manual),
not osa, and no office alarms result.

In general, manual operations on Lpss-3 override automatic opera-
tions. A manual switch may be executed on one line while another has
an automatic switch in force. The manual switch initiation causes
the established automatic switch to be released before the manual
switch is executed. This capability allows operations personnel to
control which of several failed lines is to be switched to the standby
line. If the manual switching procedure is executed on a line that has
an automatic switch in force, the control of the protection switch is
made manual (osa extinguishes, osm lights), and the line switch will
not release automatically after the failure clears.

The release of a manual switch is identical to the automatic release
sequence of Fig. 5 and Section 4.5, except that the action must be
started manually and the 30-second delay is bypassed. If a manual
switch on a nonfailed line is released with another regular line failed,
an automatic switch on the failed line will result. The release of a
manually switched line that is failed results in a transfer of control
to the automatic mode (osm extinguishes, osa lights), but does not
cause a release. A manual release may be performed at any time, not
only when a manual switch is in effect. This allows operations personnel
to clear the TsRF condition, should it occur (Section 4.5).

The lock normal condition is manually initiated to prevent the use
of the standby line for service. This feature is used while the standby
line is being equalized, for example. The essence of lock normal is that
switching is inhibited. Restoration lock normal also prevents switching,
but is intended for use when the standby line is being used to carry
service from a failed facility that is not normally associated with that
standby line. Unlike lock normal, restoration lock normal is activated
and released from the restoration patch bay and not from the Lpss-3
bay.

Lock normal may be activated at any time that neither a manual
switch nor restoration lock normal is in effect. Restoration lock normal
is inhibited by a manual switch, lock normal, or a standby line termina-
tion (Section 4.9).

Activation of either lock normal or restoration lock normal results
in the release of any automatic line switch that may be in force.
Release of lock normal or restoration lock normal allows the failed
line to again complete an automatic switch.

The line termination capability of Lpss-3 provides a check against
the propagation of overloads on the Ld lines. Each line has a terminate
switch located on the output side of the receiving switches (Fig. 1).
Each regular line terminate switch may be set in the manual or auto-
matic mode. Under the normal automatic control, any terminate
switch that is connected to an overloaded line by the receiving switch
matrix is operated, thereby removing the overload from subsequent
system components. As previously discussed, an overload on a regular
line will cause an attempt to switch that line out of service. The
automatic line termination will result only if the switching action (or
lack of it) results in a persistent overload condition at the output of
the receiving switch matrix. When the manual termination mode is
selected, the particular terminate switch involved may be manually
operated at any time that the terminate switch is connected to a
failed line, where the failure may be due to either pilot deviation or
system overload. Automatic terminations result when switching action -
cannot stop an overload, while manual terminations are enabled when
switching action cannot remedy a failure, either pilot or overload.

The L5 manual equalization procedure is accomplished out of service,
with the message removed from the line facility. A message cutoff

switch is therefore located in each transmit line connect panel. Since
untimely operation of this switch while the particular line is in service
would cause a service loss, activation of a regular line message cutoff
switch is enabled only when the line is manually switched or manually
terminated. The normal procedure for removal of the message from a
regular line for equalization or measurement is to manually switch
the line, then operate the message cutoff control. Enablement of the
message cutoff feature for a manual termination is intended to allow
the removal of an overload condition from a line to facilitate measure-
ment of that line. Message cutoff capability for the standby line is also
provided. Enablement of the feature is caused by lock normal or
standby terminate.

Whenever the message cutoff feature is in force, none of the pre-
conditions for message cutoff can be released. For example, if a regular
line is manually switched and the message cutoff is activated, the line
would be isolated at both transmitting and receiving ends. If the
manual switch is inadvertently released, service would be lost. To
prevent this, the manual release is inhibited while the message cutoff
and manual switch conditions exist on the same regular line.

The steps taken for a regular line message cutoff operation are shown
in Fig. 6. Provided one precondition is met when the control is acti-
vated, signaling priority is established and the appropriate command
is sent to the transmitting end. The command is decoded, causing the
operation of the message cutoff switch and the illumination of an
indicator Mcso (message cutoff switch operated). Signaling priority is
then sought to return the answering signal (message received) to the
controlling receiving end of the switching section. When this command
is decoded, the Mcco (message cutoff control operate) lamp for the
involved line is illuminated, and the release of a manual switch or a
manual termination on that line is inhibited.

Release of a message cutoff condition is also executed from the
recelving end of the switching section. The message cutoff release opera-
tion is never inhibited and may, in fact, be exercised without a message
cutoff in effect.

Switching activity often has a direct effect on the presence or
absence of pilots on the L5 system. The most critical pilot is the
42.880-MHz temperature pilot. Resupply of the temperature pilot is
always provided when that pilot is disrupted by switching, specifically
for any regular line message cutoff and for a regular line termination

at offices where the pilot is not blocked and reinserted in the line-
connecting circuit. The temperature pilot is always blocked and re-
inserted on the standby line on the line side of the message cutoff
switch. This eliminates the need for standby line temperature pilot
reinsertion. The other three E3 equalization pilots are often passed
through the office. Without special provision, these pilots would be lost
in the transmitting switching section on the standby line each time a
receiving section switch is completed. Because of the relatively high
activity of such occurrences, the three E3 pilots on the standby line are
reinserted whenever disrupted by receiving switching activity. Except
for the 42.880-MHz temperature pilot, E3 pilots are not reinserted on
the regular lines, since a line termination is the only means by which

they are disrupted. The combination of the lack of necessity of con-
stant presence of the three E3 pilots and the relative infrequency of
line terminations allows this mode of operation.

One of the 10 regular lines in each direction of transmission may be
designated the key line. The key line is different from all others in that
a failure on the key line causes the release of an automatic switch on
another line, so that the key line may be switched out of service auto-
matically. The key line reacts as do the other lines in all other ways.
The key line will not automatically overtake a manual switch, can
itself be overtaken by manual action, and will not attempt to switch
to a failed standby line. The circuit module with the key line feature is
provided only if specifically ordered by the customer. When provided,
the capability may be easily disabled by using a switch on the module
or transferred from one line to another by simple module interchanging.
Figure 7 illustrates the key-line-switching process. If switching priority
is available when the key line fails, the line switch to the standby line is
executed in the conventional manner. With switching priority un-
available, a release operation will be executed if the standby is not
failed, no manual switch or lock normal is in effect, and another line
has completed a line switch. When the release is complete, the key
line is allowed to establish a line switch. Typical time for a key line
switch with another line previously switched is 21 ms from time of
failure to key line switch complete.

The terss-3 bay is illustrated in Fig. 8; one bay is required per
coaxial cable end at each office.

Powering equipment is provided in the form of regulated de-to-de
converters, fuse panels, and power alarm circuits. The converters
provide regulated outputs of +25, +12, and +6 volts de using the
—24 volts de office supply as a source. The various logic circuits
throughout the bay have decentralized voltage regulators to provide
the proper voltage levels for the integrated circuits.

Signaling circuitry is housed in four shelves; terminal strip and
signaling receiver switch unit, switch signal distribution unit, signaling
transmitter, and signaling receiver. These components contain all the
high-frequency circuitry associated with the generation and distribu-

tion of the signaling commands and also contain modular logic circuits
for the control of signaling.

The indicator and control panel provides a centralized input-output
capability for routine switching activity. Section 5.2 provides additional
discussion of this important panel and associated operations.

Per-line circuits are located on the three shelves directly below the
indicator and control panel. The lowest of these three shelves is as-
sociated with receiving operations, the middle shelf is associated with
transmitting operations, and the top shelf provides interfacing for
both receiving and transmitting circuits with the indicator and control
panel, the office alarms, and the telemetry systems. Whenever a new
regular line pair is added to an existing route, a new working module is
added to each of these shelves. Positions for lines not yet equipped are
filled by special modules that allow proper switching operation for the
equipped lines. Both the transmitting and receiving per-line modules
have logic disable circuits with the associated control key and indicator
on the module face plate. Activation of this feature prevents Lpss-3
from responding to the normal stimuli that cause automatic line-
protection switching. The intended use is for cases in which automatic
switching is to be prevented on a per-line basis for switch initiator
maintenance or repair and for lines equipped but not yet in service.

Common control circuitry is located in the bottom panel in Lrss-3.
The coaxial line to be connected to the signaling receiver is selected
by circuitry on the left-hand module. A set of 11 lamps provides visual
indication of which line is being accessed. Other functions performed
by the common control include switching priority, signaling priority,
receive switch timing, release control, standby line receiving control,
signaling retransmission, and signaling system test control.

The indicator and control panel (Fig. 9) is the focal point of LPss-3
manual switching activity and visual indications. The controls and
indications are organized in rows and columns by functions. The top
eight rows are associated with receiving functions, and the next two
rows are associated with transmitting functions. The next-to-last row
provides for signaling system control and visual displays. The bottom
row is a set of lamp test keys that allow a rapid check of the lamps in
the panel.

The columns contain per-line functions, with the exception of the
signaling system row and the left column, which provide common
functions. The second column is associated with the standby line,
while the next 10 identical columns are associated with the regular
lines. Each position on the panel contains a key, a lamp, or a key and
a lamp. The key designations are stamped on the panel to the left of
the key position, with the exception of LP rst (lamp test), which is
below the keys. The designations on the plastic covers are for the
lamps under the covers. To illustrate this, the top position in the
third column is the sEL L MAN key; the position also contains three
distinct indicators, SLM, osa, and osm. The upper left position contains
the rus key but no indicator, while the fourth position in the left
column contains the TsRF indication but no control key. The control
identifications are not repeated for the regular lines. Table I defines
the indicator and control panel abbreviations.

The indicators provide office personnel with information regarding
the present state of the switching system, while the keys allow the
state of line switching, termination, and message cutoff to be manually
controlled, as discussed in Section IV. Disconnect keys and correspond-
ing indicators are provided for receiving, transmitting, and message
cutoff line switches (e.g., TRMTG L SW Disc and TLsp—third row from
the bottom). When a disconnect key is activated, the associated in-
dicator is illuminated and the corresponding line switch is forced to its
normal nonoperated state. This allows Lpss-3 maintenance to be
carried out without concern for loss of service resulting from improper
line switch operation under abnormal switch control bay conditions,
such as having circuit modules removed. A message register at the top
of each per-line column indicates the number of completed line switches
for each regular line and the number of failures for the standby line.
These registers are nonresettable, so that the change in readings over
a time span is an accurate measure of switching activity.

Two control panel design features reduce the probability of service
loss because of accidental manual operations. Particularly sensitive
controls, such as the switch disconnect keys, are mechanically in-
hibited from accidental operation. In addition, manual operations that
could interrupt service require simultaneous operation of the involved
key and a master key, or the sequential operation of two keys, depend-
ing upon the operation.

MCCO message cutoff control operated
MCSD message cutoff switch disconnect
MCSO message cutoff switch operated
MSTR master

STOA standby terminate operate automatic
STOM standby terminate operate manual
ST OVRD standby terminate override
TERM terminate

The signaling system control and indication appearances are in the
next-to-last row. The first three positions are associated with the
signaling transmitter. A normal operation of the transmitter results
in a momentary flash of the two diagnostic lamps, TRMTR sT and
REG LOAD. An alarm lamp is associated with each oscillator, one for
each of the two signaling channels. An alarm lamp is also provided for

the clock signal that times the pulsed high-frequency output of the
transmitter.

The fourth position contains the signaling system test key and the
TEST FAIL lamp. Depression of the key causes the signaling test opera-
tion to start; if the operation has not completed in 1 second, the
TEST FAIL lamp is illuminated.

Positions 5, 6, and 7 in the signaling row are associated with the
signaling receiver. The alarm lamp com FAIL indicates that the receiver
was unsuccessful in decoding the last command. COMPL FAIL and PTY
FAIL indicate a command lacking those properties. Loss of timing
ability in the receiver is indicated by CLK FAIL. RCVR ST and DR ENAB
are the diagnostics for the receiver that flash momentarily with each
normal operation. The last two positions indicate failures of either
signaling channel and provide indication and control of which channel
is being decoded by the receiver. The channel not being decoded is used
for the complement check.

The upss-3 switching system provides protection against service
loss because of line failures and protection against overload propagation
from any source. Line maintenance is aided through manual switching
controls, including service transfer and message cutoff capabilities.
One switch control bay provides switching capability for 10 regular
lines and one standby line, both transmitting and receiving.

The trss-3 bay is modular in design and utilizes a pom signaling
system to maintain interstation switching coordination. Several design
features, both electrical and mechanical, minimize the risk of in-
advertent service loss from untimely or accidental control activation.

Copyright © 1974 American Telephone and Telegraph Company
Tue Beut System TEecunicat JOURNAL
Vol. 53, No. 10, December 1974
Printed in U.S.A.

A centralized and automated transmission surveillance system has been
developed for the LS Coaxial-Carrier Transmission System as a means
for attaining desired transmission reliability. Additional benefits are
extensive and include accurate data-processing capabilities plus substantial
cost and time economies. The system consists of two basic measuring and
control facilities: (2) a transmission surveillance center, located at a desig-
nated main station, originates all control operations and accumulates and
processes all measured data through use of a small computer, and (12)
transmission surveillance auxiliaries, located at all other main stations,
perform measurement functions as directed and return the resulting raw
data to the controlling transmission surveillance center. Digitally operated
test equipment makes desired measurements under local or remote pro-
grammed control, and the E2 Status Reporting and Control System provides
interstation transmission of commands and data through time-sharing of
a four-wire data transmission link.

Verification of the transmission integrity of a complex, broadband
network such as the L5 Coaxial-Carrier Transmission System requires
many measurements at different locations and considerable processing
of the measured data. To minimize the manpower and time require-
ments for these maintenance operations while providing a high degree
of system reliability, a centralized, automated transmission surveil-
lance system has been developed and forms an integral part of the L5
network.

The transmission surveillance system (Tss) takes advantage of a
technology evolving from the availability of inexpensive, flexible, small
central processors (minicomputers) and programmable test sets. A
surveillance network, consisting of a computer-controlled center

located at a strategic L5 main station and a set of remotely controllable
auxiliaries located at other stations and having digitally operated test
sets with automatic access to key test points, provides the following
advantages on a large segment of an Ld route:

(c) Transmission performance overview that is not possible with
local individual station maintenance.
(22) Early warning of slowly developing troubles.
(iit) Rapid localization of trouble by automatic techniques.
(tv) Efficient use of manpower through computer control of routine
tasks.
(v) Accurate, flexible records as a result of computer processing
and teletypewriter printout.

A four-wire data-transmission system transmits all remote control
commands and retrieves remotely measured data in digital form. This
data transfer capability is realized by utilizing a new feature included
in the E2 Status Reporting and Control System.

The L5 system also has a built-in fault-location capability for re-
motely identifying a defective repeater in a coaxial line. It consists of
an oscillator unit associated with each repeater and a logic unit located
in the manhole. Commands from an adjacent main station activate an
oscillator unit, and signals injected at the repeater input and output
ports are monitored at the receiving end of the line to verify proper
repeater operation.

The transmission surveillance system (Tss) for the L5 carrier system
provides computer-controlled transmission-measuring capability utiliz-
ing technology evolving in the field of digitally controlled transmission-
measuring equipment. A typical Tss, shown in simplified block schema-
tic form in Fig. 1, serves a segment of several hundred route miles in
an L5 system. The main functional units are as follows:

(z) One transmission surveillance center (Tsc), the focal point of
each surveillance system, originates all automatic operations.

(it) Several transmission surveillance auxiliaries (Tsa’s), preferably
not over 10 to 12 per system, are controlled remotely by the
Tsc and are located at all main stations other than the Tsc site.

(zi) A modular, coaxial, switched access network is associated with
each Tsc and TSA.

(iv) A fault-location facility provides test oscillators at each re-
peater, remote-control circuits, and power for operating the
oscillators.

(v) A data-transfer facility is used for transmission of remote-
control commands and measured data, and is provided by the
E2 Status Reporting and Control System on a time-shared basis
with other services such as alarm surveillance.

The Tsc is generally located at a well-manned terminal main station
having significance in the overall maintenance and operation of the L5
system area covered by the tss. The Tsc functions as a nerve center for
originating and processing automatic measurements on L5 line and
jumbogroup multiplex (smx) equipment. Upon diagnosing a trouble
condition, TSC personnel may request maintenance action at other
distant main stations, which may normally be unmanned or partially

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manned. The tsc for the initial L5 carrier installation (Fig. 2) is at
the Xenia, Ohio, station, which is on the main backbone route and has
three sidelegs and mx equipment. In this installation, the Tsc monitors
13 other stations and 815 miles of repeatered line. Placing the tTsc
at a station with the largest number of sidelegs and mx signal-process-
ing equipment maximizes the “local-category”’ control and measure-
ments. This hastens surveillance operations by minimizing the amount
of data-link time sharing required for remote control through the E2
Status Reporting and Control System.

(t) Power-feed main.
(iz) Switching power-feed main.
(vt) Terminal/terminal main.
2.4 Transmission measuring system

Recently developed 90-type digitally controlled test equipment?
associated with the Tsc and each TsaA measures transmission at the

various stations in a transmission surveillance system. The equip-
ment consists of a signal generator providing an adjustable sine-
wave output in the amplitude range of —99.9 to 0 dBm and fre-
quency range of 10 kHz to 100 MHz. The selective detector measures
the amplitudes of received signals in the range of —119 to 0 dBm and
10 kHz to 100 MHz. Most transmission measurements are made auto-
matically under computer control in the surveillance system. A digital
control unit (Dcu) associated with the 90-type equipment operates
the signal generator and selective detector in this mode. Four 16-bit
binary words set the output signal amplitude and frequency of a signal
generator. Five 16-bit words set the frequency, sensitivity, bandwidth
(250 or 2500 Hz), and noise-distortion mode of a selective detector.
An analog-to-digital converter in the pcu transforms the analog
measurements of a selective detector to digital form suitable for input
to the computer or to a data-transmission facility such as the E2
Status Reporting and Control System.

During automatic surveillance operation, an E2 Status Reporting
and Control System associated with the Ld system transmits remote-
control commands from a Tsc to the various Tsa’s and raw measured
data from the tTsa’s to the computer in the tsc. The E2 system, as
arranged for L5 use, provides interstation communication under the
following categories or modes:

(t) Alarm polling.
(17) Status reporting.
(177) Remote switching.
(iv) Data transfer or remote callup.

Alarm polling, the principal E2 function, continues automatically until
interrupted by a request for one of the other functions. Data transfer
required for Tss operation is interleaved on a time-shared basis with
the other E2 operations. If not inhibited periodically, the remote
callup processing of most Tss programs would interrupt the alarm-
polling cycle longer than the permissible alarm-updating period. There-
fore, the E2 system suspends callup operation every 30 seconds and
polls all remote stations for alarms. Callup resumes after a complete
alarm-polling cycle, which takes two to four seconds, depending on
where the sequence starts.
The E2 system consists of the following units:

(z) A central station, which initiates, supervises, and controls most
of the E2 system operations.

(tit) Remote call-up units (RcU’s) (one provided as an integral part
of each remote station having general-purpose data-transfer
capability).

(iv) A four-wire data link for interconnecting the remote stations
and central stations. ;
The manual-type E2 central station provided for the initial L5 installa-
tion is located at Williamstown, Ky. (Fig. 2). It accommodates one
data link, which may interconnect a maximum of 16 remote stations.
At least one remote station is provided at each of the 14 Ld stations
for alarm reporting and Tss control purposes.

During alarm polling, the central station automatically interrogates
each remote station in succession and registers the station location and
category of an alarm should one occur. When an E2-central operator
desires detailed status information about a station, he manually ini-
tiates the status reporting operations. The frequency and duration of
the alarm-polling interruptions for this purpose depend on the number
of statuses assigned to each alarm indication and the amount of infor-
mation needed. Remote switching is used in Tss operations to control
the switching of remote line sections for out-of-service measurements.®
Each remote switch command interrupts the polling cycle for approxi-
mately 0.3 second.

During any of the first three E2 system operating modes, the central
station and the addressed remote station communicate only with each
other. The rcv provides the remote-to-remote communications or data
transfer capability needed by a Tsc for sending commands to a distant
TSA and retrieving remotely measured data. The E2 central station
supervises the callup operation, which involves communication among
remote stations and between the remote and central stations. The
RCU accepts and delivers commands and data in bit-parallel, word-serial
form, with each word containing 16 information-bearing bits. As used
with the L5 carrier, the E2 system transmits bits serially through the
data link at a rate of 600 bits per second.

Figure 3 shows the functional arrangement of the principal parts in
a Tsc. A desk-type console (Fig. 4) contains all the items except two.
The standard teletypewriter console is located at the left. The switched
access equipment is mounted in the surveillance distribution bay,
which is at the left rear in this particular installation.

Vo CONTROL OSCILLATOR = =
7” FACILITIES» DIGITAL
CONTROL FUNCTION
UNIT AND TO LINE
= ROUTE LEVEL OF
WRITER, AUXILIARY BETECrOe LEVELS OF -©Q SWITCHED
DIGITAL SWITCHED ACCESS
CONTROL ACCESS NETWORK
UNIT NETWORK

TAPE COMPUTER
UNIT TO FAULT-
LOCATION
OSCILLATOR
CONTROL
PUNCHED COMPUTER- TO E2
TAPE TELEMETRY TELEMETRY
READER INTERFACE SYSTEM

The small general-purpose digital computer (minicomputer) in the
TSC serves as the central processor in a Tss. The extensive computational
and control capabilities in it provide much flexibility for scheduling
and sequencing measurements, processing data, and presenting results.
Among the salient features of the computer are core memory for
8192 16-bit binary words, a comprehensive program-interrupt system,
interface circuits for peripheral equipment, and an internal power

The tTsc operator uses the teletypewriter for entering commands
and program parameters. The teletypewriter also prints measured
results or processed information derived from measured results as
directed by the programs.

The punched-tape reader provides direct computer loading of
program material from punched tape. Depending on the type and
operational phase of a program, it may be used when initializing the
system, when adding information on magnetic tape, when running
special programs, or during diagnostic procedures.

Small magnetic tape cassettes, having a maximum storage of 180,000
computer words each, are used in the digital magnetic-tape unit for
convenient storage of programs and data. An operator loads tape con-
tents into the computer by typing a command at the teletypewriter.

The digitally programmable transmission-measuring system makes
measurements at the Tsc location when directed by the computer. The
switched access network described in Section V automatically connects
the oscillator and/or selective detector to the desired measuring points.
Binary-coded instructions pass through the auxiliary digital control
unit (AUX pcU) and are held in memory in the pcv to control test set
parameters such as frequency, output power, sensitivity, and band-
width. An analog-to-digital converter in the pcu encodes the measure-
ments into digital form suitable for input to the computer.

Logic in the aux pcu transfers and steers control commands from
the computer or manual control circuit to test equipment, fault
location, or switched access control circuits. Flip-flop memory holds
the fault location or switched access commands until the command
selection is changed or released.

Front-panel pushbutton keys and associated programmed logic in
the manual control provide manual backup of local switched-access
and fault-location oscillator control. Numeric readouts display the
selections made with these keys. Other keys permit generation of any
16-bit binary word for testing or limited operation of the transmission
measuring system. Presently, manual control is possible only on the
local-office operated equipment. Subsequently, remote manual control
capability will be provided for operating fault-location control logic
at adjacent stations and one beyond the adjacent stations.

The computer-telemetry interface circuit and a software program
driver adapt one 16-bit computer input/output (I/O) channel to the
control format of the E2 Status Reporting and Control System. The
time multiplexing in the interface circuit and software programming
enable the computer channel to receive 16-bit information-bearing
words from the E2 system and, in addition, to monitor six E2 system
control leads. The interface circuit transfers outgoing 16-bit words
from the computer without modification, but does stretch the outgoing
strobe pulse (device command) to the duration required by the E2
system.

The tsa has all the functional units of a Tsc except the computer
and associated I/O peripherals (Fig. 5). A bay framework mounts all
equipment except the transmission-measuring system, which is as-
sembled in a rolling console and plugged into the bay. The console
may be disconnected temporarily from the Tsa and used for general-
purpose measurements anywhere in the office during visits by main-
tenance personnel. The tsa bay also contains portions of the switched
access network.

DIGITAL FUNCTION
CONTROL AND
UNIT ROUTE TO LINE
LEVELS OF ENTRED
AUXILIARY SELECTIVE Lanter NeCESG
DIGITAL DETECTOR A NETWORK

The tsa operates the local switched access network, fault-location
oscillators, and transmission-measuring equipment similarly to the
tsc. However, all automatic commands originate at the computer in
the Tsc and are delivered to the tsa through the E2 data link. When
directed by an operational program, the Tsa sends raw measured data
in digital form through the E2 data link back to the computer in the
tsc. The local switched access network and locally powered fault-
location oscillators may be operated manually by means of the Tsa
manual control, which is the same as that in the Tsc.

Signal paths between the measuring equipment and desired Ld
system test points are established under computer or local-manual
control through a three-level, dual array of multiport, coaxial, ferreed-
type switches (Fig. 6). The transmitting array conveys outgoing
signals from the oscillator in a TSA or TSC, and the receiving array con-

veys the signals from certain designated measuring points to the
selective detector. Interconnection with the L5 line-protection switch-
ing system safeguards working channels from strong test tones.? A
switched-access-network transmitting path is not completed to an L5
line unless the line is out of service.

The circuit and equipment groupings of three-level switching
(function, route, and line) offer much flexibility and accessing capacity
and permit modular growth. The basic arrangement of Fig. 6 yields a
maximum of 528 transmitting line-level ports and 1584 receiving line-
level ports, not including those used for network calibration. Only 110
transmitting ports and 330 receiving ports are used for switched access
to L5 line facilities in a maximally equipped Tsc location (10 routes).
Most of the remaining capacity will be used for accessing jumbogroup
multiplex equipment if the station has much of it.

A port on the function switch determines the class of an accessed
measuring point, such as a particular place in the L5 lines or the mmx
equipment. For example, port 1 of the receiving function switch
always gets signals from a test point designated receiving line test
(RCVG LINE TST), regardless of the selected route or line. A port on the
route switch determines the cable entrance, and a port on the line
switch determines the coaxial line.

The route- and function-level switches, with an associated power
enabler and decoder circuit for controlling all switches, are mounted in
the tsa bay at Tsa locations and at a Tsc in the nearby distribution
bay. A line access bay associated with each L5 transmission bay
(transmit-receive) lineup contains the line-level switches, which make
connections to the L5 line test points. A transmit-receive bay lineup
is associated with each cable entrance, which is designated as a route
for Tss administration and control purposes. A station without sidelegs
and located along a through backbone system has two of these routes
defined above, which result from the two cable entrances. Each sideleg
contributes one route to a station, since it has one cable entrance.

The switched access network is controlled either automatically or
manually by registering specific binary-coded commands in the aux
peu memory, which outputs this information as continuous de signals.

The power enabler and decoder circuit decodes and converts these de
signals to decimal arrangements with current capacity and voltage
necessary to energize the switch windings. Switch-selection control
progresses through the levels as follows:

(z) A signal from the aux pcv enables the function switch.
(iz) Operating a function-switch crosspoint enables the route switch
connected to that crosspoint.
(iz) Operating a route-switch crosspoint enables the line switch
connected to that crosspoint.

Interlock logic prevents simultaneous closure of more than one cross-
point at the same switching level to assure adequate crosstalk isolation
in the access network.

Several circuit features and operating procedures combine to mini-
mize the contribution of the switched access network to measuring
errors. For receiving measurements, the losses are effectively calibrated
out in the paths from the test access points in the transmission bays
to the selective detector input. The implementation is as follows:

(z) All B cables are made equal length between the 12 X 1 line-
level switch and the associated receiving measuring points in
the transmission bays (Fig. 6).

(t7) The loss-frequency slope from the level-control output through
the C cable to the 12 X 1 line-level switch input is made equal
to that of the B cables.

(iit) As part of the measuring program, a reference value is obtained
at each frequency by sending a signal from the oscillator,
through the level-control unit to the 12 X 1 line-level switch,
and then down through the route and function switches to the
selective detector.

(zv) The computer programming takes the reference into account
in determining the absolute amplitude of the received signal.

(v) The attenuation in the signal-splitting arrangement at the
level-control output brings the calibration-signal amplitudes
near those of frequently measured signals to minimize or
eliminate attenuator ranging in the selective detector.

For transmitted test signals, the circuit paths from the level-control
unit to the transmitting point, where signals are applied to the line, are
equalized to a nominal 50-dB flat loss. The accumulation of small errors

in the switched access network, together with the inherent accuracy of
the transmission-measuring equipment, should yield an overall mea-
suring accuracy of better than 0.2 dB for received signals. Straightaway
measurements should have slightly less accuracy because of switching
and measuring equipment involvement at two locations.

Location of failed or degraded repeaters along an L5 coaxial line is
based on measurement of tones transmitted through the line from a
four-oscillator fault-location unit associated with each repeater (Fig.
7a). During fault location, de power and control commands are sent
through interstitial wires in the cable from the nearest Tsc or Tsa to
the manhole locations to energize one oscillator group at a time. Two
simultaneously closed coaxial switches in an energized oscillator unit
connect the four different test frequencies to the input and output of
the repeater, as shown in Fig. 7a. The separations between the two
low frequencies and between the two high frequencies permit measure-
ment of each individual tone with a selective detector in a distant Tsc
or TSA.

Before the oscillators are installed in a repeater manhole, the out-
puts are adjusted so that signals leave the location as follows:

(z) The two low-frequency signals have nominally equal amplitudes
if the low-frequency gain of the repeater is normal.

(iz) The two high-frequency signals have nominally equal ampli-
tudes, but not the same as the two low frequencies, if the high-
frequency gain of the repeater is normal.

Therefore, the signals received from a basic repeater indicate a gain
abnormality if the low-frequency pair or the high-frequency pair
differs in amplitude. Signals from a regulating repeater require differ-
ent interpretation because the gain varies with seasonal temperature
changes. Equal amplitudes should be expected in the received signal
pairs only when the cable temperature is at the seasonal mean.®

A rectifier power supply and control unit associated with each route,
or cable entrance, at a Tsc or TSA station directly control one fault-
location unit at a time through interstitial wires in the coaxial cable
(Fig. 8). In straightaway runs, direct control extends halfway to the
adjacent station, which in turn controls the far half of the power

am
| APPARATUS
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15
INTERSTITIAL
WIRES

section. The 48-location capability (maximum) of the fault-location
power arrangement permits one-end control in a significant fraction
of the sidelegs and end links. Voltage drop in the interstitial wires and
the permissible sending-end voltage limit the powering and control
distance.

(t) Upon receiving an appropriate two-word sequence from the
computer or manual control at a Tsc location, or from the E2
system or manual control at a tsa location, the aux pcu
registers the selection (route, line, subsection, and manhole).

(72) The aux pcu sends continuous de voltages through office
wiring to operate a fault-location-oscillator power-control
circuit.

(tit) The operated power-control circuit makes connections to an
associated power supply, applies constant current to one in-
terstitial wire (+J), applies binary-coded combinations of
plus and minus voltages to eight interstitial wires (A to H),
and establishes a connection to one of six subsection wires
(SS1 to S86) that provides a return path for the oscillator-load
and control-signal currents.

With one possible exception, each subsection wire serves a group of
eight consecutive locations. The last or highest numbered wire serves
the most distant group, which may have fewer than eight locations.
Most rss control words use the four most significant bits (NV, R, S, and
T’) of the 16 for strobe steering or addressing. Fault-location-oscillator
turn-on also uses a fifth bit (/) for this purpose. The first fault-location
control word conveys the route selection via bits A to D; and the second
word conveys the selections of line via bits A to H, manhole via F to H,
and subsection via J to L. A binary-to-decimal decoder provides a de
signal on one of 10 outputs to enable the power-control circuit asso-
ciated with the desired route.

Transistor logic in the enabled power-control circuit operates minia-
ture relays, which apply the appropriate control voltages to outgoing
wires A to H and complete connections from the power supply to the
+J constant current lead and the desired subsection lead, SS1 to SS6.

The constant-voltage control signals applied to leads A to H are
referenced to the subsection leads at the sending end. Positive voltage
signifies a logical 1 and negative voltage a logical 0.

At each controlled location, a combinational logic circuit (381A gate)
bridged across the interstitial control wires decodes the binary-coded
combinations of positive and negative voltages on these leads and
performs desired switching (Fig. 7b). The circuit consists of a tree of
many simple transistor gates and transistor switches. A mechanical
rotary switch assigns one of eight binary codes appearing on wires F,
G, and H to each manhole in the subsection. The load- and line-selec-
tion logic part of the 31A gate responds to voltages on the A to E leads
and selects a desired oscillator load. Coaxial windings in series with the
oscillator loads enable directing the outputs to a desired line. Of the
32 binary combinations available with the five A to E bits, 22 only are
used for oscillator turn-on and switching, corresponding to the maxi-
mum number of coaxials in a cable. Inhibit logic in the power-control
circuit at the station prevents sending out an unused code, which could
operate the 31A gate ambiguously.

The low current drains in the 31A gate decoding circuits permit
individual control of fault-location oscillators at many different loca-
tions over a few wires of relatively small diameter. The turned-off
locations draw almost negligible current. Since the coded control in-
formation is applied continuously while a load is turned on, the inter-
stitial wires need not be loaded or delay equalized as for pulse signals.
Use of memoryless logic contributes to reliability by minimizing the
possibility of a sustained lockup at a remote point during a malfunction
of the control system.

Commercial 117-volt, 60-cycle power energizes the entire Tss. Should
this source fail, an “‘essential’’ supply such as a local engine- or turbine-
driven alternator carries the load after a few seconds of interruption.
Since the Tss is off-line equipment not in the service transmission paths,
the five-second or longer start-up time permitted in an “essential”
category supply is tolerable.

Solid-state rectifier supplies provide the nominal 5- and 25-volt de
needed to operate the office logic and control circuits in the tss. Fuses
in the distribution paths to the various circuits provide overcurrent
protection and convenient power-disconnect capability for main-
tenance. A blown fuse operates an alarm relay, which activates the

local office alarms and furnishes an indication to the E2 Status Re-
porting and Control System for transmission to the E2 central.

The tTsc equipment is assembled in a desk-type console having a
writing area and a turret-type structure with three vertical panel
regions just beyond the writing area (Fig. 4). Frequently used units
have front-panel access. The left turret has the computer, the center
turret the manual control panel, and the right turret the transmission-
measuring system. The punched-tape reader is accessible in the left
front, behind a hinged door under the writing area. The teletypewriter
console is conveniently located to the left of the Tsc operator position.
Infrequently accessed units such as logic circuits and power supplies
are mounted in the rear of the console behind hinged doors. Slides on
the logic unit shelves facilitate access for maintenance.

A surveillance distribution bay located near the Tsc console contains
the function- and route-level coaxial switches for the switched access
network, the associated control circuitry, and terminal blocks for
connecting numerous station control wires to the Tsc and line access
bays.

The tsa bay contains essentially all the equipment of both the Tsc
console and the surveillance distribution bay except the computer, the
punched-tape reader, and the transmission-measuring system. Person-
nel normally stand while operating Tsa equipment locally. The same
type of transmission-measuring system as that in a TSc is provided in
a rolling console.

One line-access bay is associated with each route or cable entrance.
It is located in the associated lineup of L5 transmit-receive bays and
contains principally fault-location power and control equipment, the .
line-level switches and level-control unit of the switched access net-
work, and the 31A gate for controlling fault-location oscillators in the
station.

Except for the 31A gate, which uses discrete solid-state components,
the various logic control circuits throughout the Tss contain mostly
commercial integrated circuits assembled on plug-in printed wiring
boards. Most mounting frameworks and panels are fabricated sheet
aluminum.

Software has been developed to exploit the computational and
control facilities of the Tss computer (Section 3.1). Features are pro-
vided to minimize operator interaction and to process and present

results in an easy-to-use format. Discussion of the software has been
divided into the following categories.

(7) Application programs.
(177) Data-base generation and administration.
(tv) Diagnostic system.

Computer programming is done in two languages, Assembly and
FORTRAN. Assembly language is used primarily to write software
drivers (routines to control peripheral devices), where the pseudo-
machine-language code is needed to set up and complete input/output
operations. The higher-level FoRTRAN language is used for applications
programs (complete routines for each surveillance activity), where
advantage is taken of its speed and flexibility in program development.

A teletypewriter is used to enter system commands and program
parameters. The teletypewriter is vendor-modified to allow bit-parallel
transfer of information and independent control of the input, print,
and punch functions.

The fault-location subsystem, switched access network, and test
set are controlled directly by the computer at Tsc locations. The com-
puter remotely controls identical equipment at Tsa locations via the
K2 telemetry facilities.

To relieve the Tsc operator of much punched-tape handling, a
cassette-type digital magnetic tape unit is also interfaced with the
computer; and programs for surveillance system operations are stored
on magnetic tape. The magnetic tape unit can read or write on a data
track or an address track and access tape files quickly. Tape positioning
addresses are prerecorded on the address track of each cassette in
proportion to the number of spindle revolutions. To access a file, the
tape unit counts spindle revolutions as the file approaches at high speed
and then reads the address track only during the last few revolutions
to verify that the desired tape position has been reached.

A 64-word program, used to load other programs into the computer
via the high-speed punched-tape reader, resides in a protected area of
computer memory. This program is used, for example, to load the Tss
bootstrap program, which, in turn, loads the operating system from
magnetic tape.

Basic Binary Loader
Tss Bootstrap
Directive Processor
Input/Output Control
Drivers for:
Teletypewriter
Paper Tape Reader
Magnetic Tape Unit
Test Set
Fault Location Oscillator
Switched Access Network
E2 Telemetry
Formatters for:
Teletypewriter
Magnetic Tape Unit
Arithmetic Routines

Generators for:
Pilot Measurement
Fault Location
Line Measurement
Jmx Measurement
tss Diagnostic

Computer Diagnostic for:
Computer Instructions
Computer Memory
Hardware Functions

catable loader, and magnetic-tape storage programs, used to prepare
and manufacture those programs that are a part of the Tsc.

The Tss operating system processes a small set of operator directives
(Table II); and, to economize magnetic-tape space and reading time,
it contains a set of subroutines expected to be used by all applications

programs.
Table I! Operator directives
Command Function
LOAD (NO.) To recall and start an applications program stored on
magnetic tape.
BEGIN To restart the program currently in computer core.
RUN To continue a program after a programmed pause for
manual operations, or
To retry certain input/output operations resulting in
error messages.
REWIND To rewind the magnetic tape cassette to clear leader.

Applications programs are stored on magnetic tape and can be re-
called by file number into the computer for execution upon an operator-
initiated Loap (file number) command. At the termination of a pro-
gram or after a programmed pause, the operator may request another
program to be loaded and automatically started. He may restart the
program currently in the computer core with the BEGIN command; or,
in the case of a programmed pause for manual operations, he may cause
the current program to continue with the RUN command. Also, using
the RUN command, he may request retries on certain input/output
operations that result in error messages. All commands for these opera-
tions are given by the operator at the teletypewriter.

Vendor-supplied, interrupt-compatible software drivers for the
teletypewriter and magnetic tape and an input/output control pro-
gram for policing requests to these devices are required to perform the
operator directives. Software drivers have been developed by Bell
Laboratories to control the E2 telemetry system, the fault-location
subsystem, the switched access network, and the programmable test
sets.

Vendor-supplied library routines such as a formatter for teletype-
writer operations and arithmetic subroutines are included in the
operating system because of frequent use by applications programs.
Also included are Bell Laboratories developed rortTRAN-callable rou-
tines for teletypewriter plotting of measurement results and for con-
trolling magnetic-tape operations. The last-mentioned routine permits
reference measurement files and data-base files to be stored on magnetic
tape and/or recalled under program control.

All the above routines are stored on magnetic tape as file number 2,
which is automatically recalled by loading a short paper tape referred
to as the Tss bootstrap. The teletypewriter prints the message *WHAT?
when the computer is ready to accept an operator directive. At that
time, the operator can request, for example, that a specific application
program be loaded from magnetic tape and executed. File number 1 is
reserved for a tape address directory, which contains the tape position
of up to 67 other files for application programs, reference-measurement
files, and data-base files.

Use of a programmable computer permits quick and easy imple-
mentation of various maintenance activities. In general, an applica-
tions program is a complete sequence of computer instructions designed
to monitor and evaluate a particular function of the L5 transmission

The surveillance library serves as a memory jog for the operator.
It prints a list of program names and numbers and optionally will print
detailed instructions regarding a specific program.

After starting the automated fault-location program, the operator
enters the set of cable routes (one or more power-feed sections), man-
holes, and the lines containing the repeaters to be interrogated. The
computer recalls a data base from magnetic tape and checks for the
presence of requested repeaters.

For each section of coaxial line requested, the computer initiates the
following: (2) calibration of office trunks and coaxial switches in the
line-access network between the RCVG LINE TsT point (Figs. 6 and 9)
and the receiver, (77) access of the test point, (777) turn-on of the oscil-
lators for one repeater at a time, and (v) measurement of the received
set of tones. Repeater gain deviations, as derived from tone measure-
ments, are compared with allowed limits and are printed only for out-
of-limit repeaters unless the operator specifies, when starting the
program, that all measured and derived data be printed. As each section
of line is completed, the program releases the switched access network
and the fault-location oscillators.

Figure 10 is a sample printout depicting a repeater in trouble. The
teletypewriter bell and special characters in the printout provide
audible and visual indications of anomalies and out-of-limit conditions.
The operator may optionally obtain a teletypewriter plot of the
received tone amplitudes as a function of distance along the cable.

In a composite message signal, pilot tones may be monitored by the
pilot measurement program. The operator may select from the set of
line, jumbogroup, mastergroup, and supergroup pilots. Pilots can
automatically be measured at both ends of all smx sections or optionally
on a set of lines between two stations. The computer recalls a data
base from magnetic tape and checks for allowable combinations of
transmitting and receiving stations.

For each section requested, calibration of office trunks and switches
is performed at both stations. At the transmitting end of a section,
pilots are accessed and measured via the TRMTG LINE TsT point (Fig. 9)

and, at the receiving end, via the RcvG LINE TsT point. The difference
in pilot amplitudes between the two stations is a measure of the gain
characteristics of the line. Out-of-limit conditions or, optionally, all
pilot amplitudes and differences are printed as a function of frequency.
This printout can serve as an early indication of the need to re-equalize
a line.

The line measurement program is used to determine the gain-
frequency characteristic of a coaxial line with better accuracy and
finer granularity than with pilot measurements. A sequence of tones
is injected at the EQL sIG IN point in the station at the transmitting
end of the desired line and measured at the RcvG@ LINE TST point in
another station. Using the line-protection switching system, the opera-
tor must first take the line out of service and operate the message cut-
off switch.

After taking the line out of service, the operator selects the trans-
mitting and receiving stations, the line, and the set of frequencies, and

decides whether or not a teletypewriter plot of the gain characteristic
is required. Careful study of the printout can be used to evaluate the
need for or results of equalizer adjustment. Measurements can be
stored on magnetic tape for future comparison with historical data.

Using the smx measurement program, the operator can access the
remote test switch for each jumbogroup and calibrate office trunks and
coaxial switches. The program then measures carrier or jumbogroup
and mastergroup pilots at 11 test points per jumbogroup. The data
are compared with limits and with historical data. The historical data
are updated on magnetic tape, and a printout of out-of-limit conditions
is made.

The general-purpose loss measurement program operates the trans-
mission-measuring system in a TSC or a distant TSA under computer
control to observe directly connected equipment or facilities in the
selected station. Loss vs. frequency characteristics are printed and,
optionally, plotted.

The operator can select the station address of the test set and all
programmable functions of the test set, including the transmit ampli-
tude and set of frequencies. Measurement averaging and reuse of
calibration measurements are available.

Several application programs use data bases that are prepared by
the data-base-generator program and stored on magnetic tape. The
data bases determine the arrangement and extent of surveillance
equipment to be controlled by a center Jocation, as well as testing
limits, etc. Separate data-base files are created for the automated fault
location and Jmx measurement programs, whereas the pilot and line
measurement programs share a data-base file.

Each data-base-generator program is stored on magnetic tape and
recalled by the operator using the same procedure as for other applica-
tions programs. The program simply asks questions that the operator
must answer. For many questions, the operator may elect to use the
preprogrammed answers. Questions are answered to define and cor-
relate manhole designations, route numbers, coaxial lines equipped,
telemetry addresses, the allowable range of measurement levels, test
points available, etc. When all questions are answered, a separate data-
base file containing this information is automatically created on

When new or additional equipment must be accounted for in the
data-base file, area engineers provide the craft operator with completed
forms for updating those files.

A separate set of programs, stored on a second magnetic tape, is
dedicated to checking transmission-surveillance-related equipment for
proper operation. These diagnostics are called into the computer using
the Tss operating system and, if no errors are detected, control is
returned to the operating system. Errors are indicated by a computer
halt or message on the teletypewriter.

Separate programs are used to test the teletypewriter, high-speed
punched-tape reader, digital control unit, and aux pcu. The magnetic
tape unit diagnostic is loaded from punched tape.

The computer diagnostic control program supervises the running
of a large set of vendor-supplied computer diagnostics. There are
diagnostics for testing all instruction types, core memory, and other
computer hardware. Each computer diagnostic has an overlay as-
sociated with it to allow nonstop testing and multiple execution of
each diagnostic. The control program interacts with the operator to
determine the diagnostics to be run.

The Tss diagnostic program provides an indication of the health of
the entire surveillance system by exercising all the hardware at the
tsc and the Tsa’s. To narrow down possible trouble causes, each new
phase of testing involves a minimum of previously untested equipment.
When a problem is detected, more detailed testing is performed in the
area affected. To determine what facilities are equipped in each office,
the program shares a data base with the automated fault location
program.

The degree to which the diagnostic system localizes troubles is
established principally by the number and significance of the places
where the statuses of transmission and logic signals can be determined
automatically and remotely. The Tss automated diagnostic procedures
generally localize a fault to a subsystem small enough for knowledge-
able craft personnel to find the actual fault quickly.

Since the transmission surveillance system provides the potential
for maintaining the L5 system on a network basis rather than as a

series of independent stations, many somewhat complex and inter-
related factors must be considered in the evolution of an overall main-
tenance plan to effectively realize this advantage. To begin with, a Tss
controlled by a single tsc should be assigned to a portion of L5 route
on a basis compatible with the overall administration of the system.
The association of surveillance operations and alarm reporting, as
provided by the E2 system, is another important consideration. A
third consideration is that of defining the boundary between adjacent
surveillance systems along a route. Finally, in developing an overall
maintenance plan, intervals for performing routine tasks must be
determined by carefully considering such factors as system size,
measurement speed, need for update, and output requirements. Of
course, considerable blocks of free time must be left for performing
demand-type measurements to locate a trouble or to characterize
performance, as a result of an external stimulus such as a line switch,
E2 alarm, or a line-equalization adjustment.

The rules for the layout of a transmission surveillance system are
determined as much by administrative and management aspects as by
any hardware constraints. Presently, the E2 data facility has a capacity
of 16 remote stations, which limits a Tss to 15 Tsa stations and one con-
trolling Tsc station. The resulting system bound of this number of
stations, likely near 800 to 900 route miles, probably approaches the
limit that can be monitored effectively by a single rsc. Other con-
siderations include: (¢) AT&T-Long Lines area boundaries, (77)
planned E2 alarm reporting and other maintenance arrangements,
and (iz) location of strategic stations along the route.

Once a TSS arrangement has been established within a region, it is
imperative that a single E2 data facility interconnect all the stations
served by the Tss to provide data-transfer capability to and from the
tsc. Generally, this data facility will be time-shared with normal E2
alarm polling operations.

Judicious tsc placement enhances the effectiveness of a Tss. Several
factors should be considered in selecting the station. It should (2) be
on the associated L5 route, (77) be a major, manned facility, (277) have
significance in the area maintenance plan, and (zv) be a junction point
with signal processing (multiplex). Examination of the Tss area and
consideration of these points should lead to an appropriate Tsc place-
ment. Another highly desirable goal is to colocate the Tsc and E2
central, if at all possible, since this would centralize all major mainte-
nance operations. It is important that the Tsc location be manned 24

hours a day. Although it is possible to control the tsc computer over a
data set/rry link from another office, perhaps off the route, it is not
considered desirable unless craft personnel thoroughly familiar with
Ld operations are available to interpret performance data.

The criteria for obtaining a viable tss layout have been given
(Section 10.1), and, if availability of computer software is assumed, the
role of the surveillance system in the overall maintenance scheme for
the L5 system can be defined. As previously discussed, the Tss performs
transmission measurements for the purposes of maintaining a high level
of performance and locating troubles. These operations fall into two
categories:

The demand measurements are unscheduled and the need is gen-
erally spontaneous, resulting from a trouble condition. The routine
operations, where the words transmission surveillance apply, are sched-
uled to realize continued monitoring of performance and to spot any
deteriorating conditions. Procedures and associated software for an
initial L5 maintenance plan have been based on experience gained
during the L5 field trial, in the early stages of the initial system turn-up,
and in discussions with AT&T personnel. This plan, when implemented,
will assist in maintaining a high quality of service on the L5 system.
In simple terms, the plan consists of:

(7) Pilot measurements : Transmitted and received line and master-
group pilots will be measured daily between all signal-processing
points (terminal stations) to characterize the transmission per-
formance of the coaxial lines.

(iz) Fault location runs: All line repeaters will be checked weekly,
and out-of-limit conditions and regulating repeater gain
changes will be printed out.

(ziz) Jumbogroup multiplex measurements: Test points in all mmx
equipment will be monitored on a weekly basis and out-of-
limit conditions will be printed out.

(iv) Line gain/frequency measurements: Detailed measurements,
on an out-of-service basis, of the frequency characteristic of a
coaxial line will be made following any repair or realignment
operation to verify proper performance.

_(v) Tss diagnostics: Proper operation of the surveillance system
itself will be routinely checked on a weekly basis. This includes
an overall test at each station to pinpoint any problems in the
access network, control circuits, or test equipment as well as
diagnostic tests of the computer and associated peripherals at
the TSsc.

According to calculations for the 815-route-mile initial system be-
tween Lillyville, Pa., and Hillsboro, Mo., these routines should use
about 50 percent of the total hours in a seven-day week when all 22
tubes are in service. The calculations have assumed that only a few
pilot measurements are needed beyond those in the daily overall check
between terminal stations to associate a trouble with a particular
power-feed section. Methods are being investigated to reduce the
measurement times before systems reach full capacity so as to free the
tsc for troubleshooting operations when necessary. Furthermore, the
presently provided facilities are probably just the beginning of auto-
mated transmission-measuring capability. As experience is gained with
centralized and automated maintenance and more sophisticated test
equipment becomes available, the role of the Tss will grow—perhaps
into the areas of acceptance testing, trouble shooting of individual
equipment units, and interfacing with other evolving maintenance
systems.

Realization of the transmission surveillance system for the L5
coaxial system has required the coordinated efforts of many persons.
The authors hereby acknowledge the contributions of the following
immediate associates: P. M. Berard, M. A. Leveille, and J. P. Russo
for circuit design and performance verification ; R. A. Noel for diagnos-
tic software design; M. A. Plante for application planning; and W. P.
Frawley and C. J. Rimas for physical design.

1. F. C. Kelcourse and F. J. Herr, “L5 System: Overall Description and System
Design,”’ B.S.T.J., this issue, pp. 1901-19383.

. N. H. Christiansen, “New Instruments Simplify Carrier System Measurements,”’
Bell Laboratories Record, 48, No. 8 (September 1970), pp. 232-238.

. J. H. Green and R. W. Sanders, “L5 System: Line-Protection Switching,” B.S.T.J.,
this issue, pp. 2011-2034.

. E. H. Angell, Y.-S. Cho, K. P. Kretsch, and M. M. Luniewicz, ‘“L5 System:
Repeatered Line,’ B. ST. J., this issue, pp. 1935-1985.

Copyright © 1974 American Telephone and Telegraph Company
TuE BELL System TECHNICAL JOURNAL
Vol. 53, No. 10, December 1974
Printed in U.S.A.

The jumbogroup multiplex (sux) translates basic jumbogroup signals
to and from the L& line spectrum. Each of three jumbogroups provides
8600 4-kHz voice circuits. In addition to frequency translation, the sux
provides regulation and equalization on a sumnbogneup basis and auto-
matic one-for-one protection.

Performance objectives are discussed and interpreted Gah the perspec-
tive of their anfluence on the design philosophy and approach used and
the resultant physical realization. All subsystems are presented in detail.
Particular mention is made of the double-balanced diode-ring modulator
developed for sux application.

While gross performance objectives for analog multiplex equipment
are available, the detailed statement of design objectives is part of the
development task. The basic guiding principle is that the resultant
system, the aggregate of line and terminal equipment, must provide
transmission paths of quality and reliability appropriate to the Bell
System communication network, and it must do so at a cost sufficiently
lower than other alternatives to warrant development by Bell Labora-
tories, manufacture by Western Electric, and purchase by Long Lines
and the operating telephone companies.

The basic function of the jumbogroup multiplex (4x) terminal is to
translate basic jumbogroup signals to and from the Ld line spectrum.
Each of the three jumbogroups is composed of six mastergroups.
These signals must not be corrupted or distorted by the smx in a
manner or degree that would compromise the quality of transmission.
The equipment must be reliable and maintainable. Derivable from
these considerations are a host of performance objectives, including
those discussed in the following sections.

The overall objective on noise, including thermal noise and inter-
modulation, for a 4000-mile, transcontinental, L5 system is 40 dBrnc0.?
Of the total, 39.4 dBrnc0 is allocated to the L5 line, and the remaining
31.2 dBrncO is allocated to the aggregate of all terminal equipment.
While the L5 line is, in a sense, a totally new design entity, such is not
the case in the associated terminal equipment. The jmx has to share
the terminal noise allocation with an existing hierarchy of equipment,
including channel, group, supergroup, and mastergroup banks.
Consider a representative transcontinental connection including two
channel banks, five supergroup multiplex (LMx) terminals, seven
mastergroup multiplex (mmx) terminals, and eight mmx terminals.
Assuming 18 dBrncO performance for the tmx and mmx terminals
and 10 dBrncO for the channel banks, a maximal allocation of 18
dBrnc0 results for the ymx. Allowing 1 dB for misalignment and other
vagaries, including the possibility of other connections more demanding
than the one cited, the ymx design objective was set at 17 dBrncO, with
the realization that an even lower number would be desirable. In
essence, it appeared that at least initially the design philosophy should
be to obtain the lowest noise possible. |

To put the 17-dBrncO objective into perspective, it is helpful to
view the smx challenge relative to the most comparable multiplex
available, the mmx-2.2 In an optimally designed multiplex terminal,
the controlling noise sources are the modulators and the associated
amplifiers following the modulators. Normally, the intermodulation
distortion of the modulators is greater than that of the other trans-
mission apparatus, so that lowest signal levels are found at modulator
outputs. Since levels are low at such points, the noise figure of the
following amplifier is of greatest importance. Levels are chosen to
minimize the combined noise from these critical nodes. Since the smx
uses multiple steps of modulation and demodulation, while mmx-2
uses single steps, the smx has twice as many critical nodes in each
transmission path as the mmx. Furthermore, the bandwidth of the
jumbogroup signal is six times that of the mastergroup signal. For
third-order intermodulation distortion alone, the number of products
the ymx must contend with is 16 dB greater than in the mmx. In a
sense, JMX must be many decibels better than mMmx-2 to yield com-
parable noise performance.

Based on studies of typical system configurations, with the multi-
plicity of occasions for crosstalk, an objective of 85 dB equal-level

coupling loss (ELcL) has evolved for analog multiplex equipment. This
objective was assumed for the Jmx.

The sources of crosstalk in multiplex equipment are many and varied,
and include many modes not encountered elsewhere. Crosstalk may
occur between different jumbogroup signals, between different portions
of the same jumbogroup, between signals at different stages in the
modulation-demodulation process, and in an intelligible or noise-like
fashion. Coupling may exist between signals, between carriers, between
pilots and carriers, etc. While an exhaustive discussion of crosstalk
is inappropriate here, one typical illustrative example will be
mentioned.

Given a carrier with a tone 4 kHz removed from the carrier at a
level 79 dB lower than the carrier level, intelligible adjacent channel
crosstalk appears in almost every channel at a level approximately
85 dB below the level of the interfered channel. Because of this mode
of crosstalk, spectral purity of mmx carriers is important. Since Jmx
carriers are as high in frequency as 91.648 MHz, filtering, shielding,
and grounding are critical. Common ground impedances of very small
magnitude are sufficient to cause unacceptable crosstalk performance.

Tones falling in a voice channel are particularly annoying. Often,
tones generated from several sources add coherently (voltage addition).
Because of the multiplicity of sources of tones, the design objective
for mx was that no tone falling in the jumbogroup passband should
have a level exceeding —70 dBm0. This becomes particularly challeng-
ing when, in the sox, the source of the tone may be a 15-dBm, 91.648-
MHz carrier, and transmission levels are as low as —43 dB. The
resulting implications on isolation of separate paths, carrier balance,
and filtering of carrier leak signals are significant.

The transcontinental objective on frequency offset is less than 2 Hz.
This implies an accuracy requirement of one part in 10° on the effective
carrier frequencies used in the gmx.

The overall misalignment objective for a transcontinental L5
connection is less than +4 dB. This requires that the mx passband
be flat to +0.2 dB. In addition, since the mx may carry digital
signals, delay distortion must be limited.

The L5 line,’ with its automatic line-protection switching system,‘
is highly reliable. If terminal equipment is not to add noticeably to the
mean outage time for a transcontinental connection, based on the
system configuration considered earlier, the objective for each smx
path (transmit or receive) becomes less than 0.14 minute per year.
This corresponds to an equivalent failure rate of about 200 Fit’s
(mean time to failure is 500 years), assuming that the mean time for
repair is one hour.

While it is difficult to become quantitative relative to maintain-
ability, several observations of a qualitative nature may be made.
Each piece of transmission equipment can handle as many channels
as a whole L4 system. The equipment may be housed in unmanned
main stations. Many modes of failure are of a subtle nature. It there-
fore seems desirable to use automatic, remote diagnostics extensively.
Routine measurements at the bay should be held to a minimum. Most
troubleshooting on site should require no removal of modules prior to
trouble isolation. Status indicators should be provided for the con-
trolled and trouble states of the equipment. In short, the equipment
should be easy to maintain and difficult to operate incorrectly.

The cost of the smx on a per-channel basis may be reasonably viewed
from two perspectives: (7) its contribution to the total terminal cost
for the terminal equipment required to take a voice signal from voice
frequency to basic jumbogroup frequency and back, including signal-
ing, and (22) its contribution to the cost of an L5 system for a system
length of a few hundred miles or more. It has been estimated that the
per-channel cost of the mx amounts to less than 2 percent of either
the terminal or line cost; therefore, the mx cost would have very
little influence on the total cost of providing a voice channel.

An overall block diagram of the transmitting arrangement, typical
of any jumbogroup, is shown in Fig. 1. Basic jumbogroup signals,
composed of six mastergroup signals, and a 5.888-MHz jumbogroup
pilot are fed to the smx from the basic jumbogroup trunk bay (Bser).°

These signals, contained between 0.564 and 17.548 MHz, first enter
the ymx at the transmitting jack field. Two input ports are provided.
The first or regular input is that through which the signal normally
passes. The second or spare input is provided for emergency patching
purposes and is normally terminated. These two inputs are combined
in an input hybrid transformer and then connected to a test pad that
provides a low-loss (0.6 dB) through path and a high-loss (30 dB)
bridging path. The high-loss or low-level output is fed to a splitting
transformer, and provides a local and a remote test point, both of
which are isolated from the transmission path by about 33 dB of loss.
Local test points are accessed via jacks mounted at the front of the
bay. Remote test points are provided to allow access for automated,
centralized test equipment. The low-loss pad output is fed to a splitting
hybrid that provides signals to the redundant A and B paths of the
transmitting side.

The basic jumbogroup signal then passes through the restoration
access switches which, when operated, are capable of providing an
appearance of the basic jumbogroup signal and access to the input of
the associated modulator at the restoration patch bay. These switches
are connected in a manner that allows monitoring of the interbay
cabling when the switches are in the unoperated condition.

The basic jumbogroup signal then goes to the modulators, where it is
filtered and translated to any one of three jumbogroup line assign-
ments. Multiple steps of modulation are used, and the required
carriers are provided by the associated carrier supply. Carrier supplies
are redundantly provided for each jumbogroup. Each carrier supply
provides the carrier signals for an associated modulator and de-
modulator. Carriers are generated by appropriately mixing internally
generated signals with signals provided by the sync-distribution
circuit. There are two sync-distribution circuits per bay, each of which
uses redundantly provided jumbogroup frequency supply (sFs)’ signals
to generate a number of highly stable reference signals for trans-
mission to all of the A or B carrier supplies.

The jumbogroup signal, at line frequency, is then fed through the
output restoration switch to the cable equalizer. The output restora-
tion switch, connected in a manner similar to that of the input
restoration switch, provides access to the modulator output. The
cable equalizer, located in the transmitting line-interface unit, compen-
sates for the loss of the cable connecting the mmx to the L5 line bay.®
Levels are adjusted in the transmitting line interface to provide a
stepped preemphasis for transmission over the repeatered L5 line.

The cable equalizer output passes through a test pad which provides
a high-loss access for local and remote testing and a low-loss path to
the transmitting protection switch. The transmitting switch accepts
the A and B output signals as inputs, and provides a working and idle
output. The state of the switch, set by the switch-control circuitry,
determines which input will be connected to the working and idle
outputs. Both outputs are fed to unequal-ratio hybrid transformers
which provide a high-loss (7 dB) output for monitoring purposes,
and a low-loss (1 dB) output that is connected to the transmitting
jack field. The working output signal is then cabled to the Ld line bay,
and the idle output, available for emergency patching through spare
cabling, is normally terminated.

On the receiving side (Fig. 2), the complete L5 spectrum is re-
dundantly cabled to the receiving jack field of each jumbogroup
equipment. The signal first passes through a test pad that provides
high-loss access for testing and a low-loss connection to the associated
cable equalizer. The cable equalizer compensates for the loss of the
cable between the L5 line bay and the smx. The signal is then fed to an
input restoration switch which is capable of providing an appearance
of the L5 line signal and access to the demodulator input at the restora-
tion patch bay.

The received signal then goes to the demodulator where the desired
jumbogroup is selected and translated to basic jumbogroup frequency.
After passing through an output restoration switch, which can be
used to gain access to the demodulator output for the restoration
patch bay, the signal is fed through a test pad to the basic jumbogroup
equalizer, a manually adjustable, multistage equalizer that will
provide* the capability of correcting for misalignment accrued over
many miles of the repeatered L5 line on a jumbogroup basis. The
signal then goes to the regulator which monitors the level of the
received jumbogroup pilot and modifies its gain accordingly.

The regulator output is fed through another test pad to the receiving
protection switch. Working and idle outputs are connected to unequal-
ratio hybrid transformers that provide access points for automatic
monitoring in addition to output signals connected to the receiving

* Space and power have been provided for the jumbogroup equalizer in the ymx
bay. Determination of the appropriate characteristics of this equalizer awaits evalua-
tion of misalignments for working L5 systems.

patch field. Additional equal-ratio hybrid transformers are used to
provide dual appearances of the working and idle output signals. One
of the working output signals is cabled to the Bsat. The remaining
ports, available for emergency patching, are normally terminated.

The modulator and demodulator subsystems for all three jumbo-
groups are shown in Fig. 3; transmission levels are as indicated. In
each case, the subsystems used in the A and B sides are identical.

Three types of amplifiers are used in the modulator and demodulator
subsystems. The 9-dB fixed-gain amplifiers and the 12 + 2-dB adjust-
able amplifiers are transmission quality amplifiers with a controlled
transmission band running from 0.5 to 100 MHz. They are realized
in the hybrid integrated circuit (HIc) technology. The 9-dB amplifier
is designed for low-level application and has a 5-dB noise figure. The
- 12-dB amplifier has been designed to handle higher signal levels
without introducing appreciable intermodulation distortion. Both
amplifiers are multistage, major-loop feedback designs with 75-ohm
input and output impedances. The 15-dB carrier drive amplifier is
designed to provide a 15-dBm carrier signal for the modulator. In
addition to providing highly linear gain for carriers up to 91.648
MHz, this amplifier provides at least 40 dB of reverse isolation to
control crosstalk that might otherwise be established through the
carrier drive path. A multistage local-feedback design with printed-
circuit realization is used. All amplifiers use feed-through filters for
battery connections.

The 17A modulators are double-balanced diode-ring modulators
using Schottky barrier diodes and 75:300-ohm center-tapped trans-
formers.’ These are highly linear modulators providing at least 40-
dB carrier and signal balance. Nominal port impedances are 75 ohms.
A single design is used throughout the gmx. More will be said about
the 17A modulator in Section 3.16.

The filters* have been designed on a system basis; i.e., the loss
requirements have been specified in an interactive manner in an
attempt to reach a global optimum without placing undue stress on
any one design. Often the attenuation required for some undesired
sideband can be obtained more easily through the combined effect
of two or more filters than in a single filter. The systems of filters were
specified to provide combined attenuation to undesired modulation

Fig. 3—Modulator and demodulator subsystems for jumbogroups 1, 2, and 3. (a) Jumbogroup 1 modulator.
(b) Jumbogroup 2 modulator. (c) Jumbogroup 3 modulator.

Fig. 3 (cont.)—(d) Jumbogroup 1 demodulator. (e) Jumbogroup 2 demodulator. (f) Jumbogroup 3 demodulator.

products and carrier leak sufficient to keep them at least 85 dB lower
than the level of the desired sideband. All filters have nominal input
and output passband impedances of 75 ohms.

Ideally, the basic jumbogroup signal fed to the smx should contain
no energy above 17.548 MHz. If some disturbance is present above
the desired input spectrum, it could corrupt the message spectrum
either by being superimposed as a result of linear transmission through
the modulator (signal leak) or by being translated into the message
band via the modulation process. Accordingly, after the input signal is
raised to the appropriate transmission level via the input amplifier,
it is passed through a low-pass filter to reject any spurious high-
frequency disturbance. The basic jumbogroup spectrum is then
translated in frequency using a 42.496-MHz carrier. The signal is
amplified and its level is adjusted to a more optimal value for a subse-
quent step of modulation. Next, a bandpass filter selects the lower
sideband between 29.948 and 41.932 MHz. The signal is modulated
once more, using a 45.056-M Hz carrier. After amplification, the lower
sideband located at 3.124 to 20.108 MHz is selected by a second
bandpass filter. The modulator subsystem output level is then achieved
through the use of an adjustable 12-dB amplifier, a fixed-loss pad,
and a test pad that introduces a 0.6-dB loss in the signal path and
provides a bridged test point with 30-dB isolation.

The net frequency shift provided by the jumbogroup 1 modulator
is 2.56 MHz. Since the input spectrum overlaps the output spectrum
for jumbogroup 1, multiple steps of modulation are mandatory. Signal
leak prohibits using one step of modulation (or demodulation) for
jumbogroup 1.

Jumbogroup 2 is processed in a manner similar to that for jumbo-
group 1. The first step of modulation uses a 70.144-MHz carrier. The
first bandpass filter selects the lower sideband between 52.596 and
69.580 MHz. The second carrier frequency is 91.648 MHz, and the
jumbogroup 2 signal occupies the band of 22.068 to 39.052 MHz. The
net shift for jumbogroup 2 is 21.504 MHz.

It is often more desirable to use the lower sideband, since many of
the undesired byproducts of modulation corrupt the upper sideband.
In jumbogroup 2, using a single step of modulation and selecting the
lower sideband would cause the most difficult filtering—attenuation

of carrier leak and rejection of the upper sideband—to be done in the
neighborhood of 40 MHz. Using two stages of modulation greatly
facilitates this task by shifting the chore to the region of 20 MHz.

Jumbogroup 3 uses three steps of modulation. The first two steps
are identical to those of jumbogroup 1. The third step of modulation
uses a 40.448-MHz carrier. The third bandpass filter selects the upper
sideband between 43.572 and 60.556 MHz. The net frequency shift
is 43.008 MHz.

With either one or two steps of modulation, the attenuation of
carrier leak and the rejection of the nearby sideband would require
filters of questionable realizability for jumbogroup 3. Thus, three steps
of modulation were necessary. The first two steps may be viewed as
providing a baseband signal with significant separation between the
lowest signal frequency and de. The result is that adjacent sidebands
are separated by about 6 MHz after the third step of modulation.
This, coupled with the use of the upper sideband (in this case, no
spurious modulation products overlapped the upper sideband), greatly
facilitates filter synthesis.

Following a slight level adjustment to place the signal at a more
optimal level for the first step of demodulation, the jumbogroup 1
signal is selected from the L5 line signal by the first bandpass filter
whose passband extends from 3.124 to 20.108 MHz. Using a 45.056-
MHz carrier, the first step of demodulation shifts the input spectrum
up in frequency. Following amplification and level adjustment, the
second bandpass filter selects the lower sideband signal between
24.948 and 41.932 MHz. The filter output goes through a 9-dB ampli-
fier-pad combination to provide isolation between the filter and the
following modulator. This signal passes through a second stage of
demodulation using a 42.496-MHz carrier. The signal is amplified,
and the lower sideband signal at basic jumbogroup frequency is
selected by the output low-pass filter. The basic jumbogroup signal is
further amplified, then fed to a test pad that provides a local bridged
test point.

The jumbogroup 1 demodulator is similar in most aspects to the
jumbogroup 1 modulator. It uses the same carriers and uses filters
with identical passbands (although the detailed nature of the rejection
bands is not always the same).

Modulators of the type used in the smx are bilateral. That is, a
signal applied to the output port will be translated in frequency and
appear at the input port in the same manner that signals applied to
the input are translated and appear at the output. Consider a modu-
lator with conversion loss C' terminated on its output by a load whose
return loss, with respect to the average modulator output impedance,
is R,, and terminated on its input by a source whose output return
loss, relative to the average modulator input impedance, is R;. Signals
appearing at the modulator output will have some energy reflected
back through the modulator, and some of this energy will, in turn, be
reflected from the source output through the modulator once more.
The result will be an echo-like signal superimposed on the original
signal at the modulator output. The reflected signal will be attenuated
relative to the original signal by 2C + Rk, + AR; dB.

The modulator input and output impedances are complex and
periodically time-varying, and are not capable of being carefully
controlled and manipulated without significant decrease in conversion
efficiency. In practice, the average input impedance is controlled to
some nominal value. For the smx, this nominal value is 75 ohms.
Similarly, amplifiers, pads, and in-band filter impedances are designed
to be 75 ohms. Filters are designed to pass in-band energy and reject,
principally through reflection, out-of-band energy. Consequently,
filters typically have rejection-band terminal impedances correspond-
ing to open or short circuits.

Assume that a modulator is driven from a filter and operates into
an amplifier. If the amplifier input impedance has an average return
loss of R, = 18 dB, relative to the modulator output impedance, the
filter out-of-band return loss is R; = 0 dB, and the conversion loss
is C = 6 dB, then the reflected signal will be suppressed 30 dB relative
to the signal with which it adds, assuming that the signal that ap-
peared at the filter input was an out-of-band signal. The interfering
signal will be at the same frequency as the interfered signal and will
add on a voltage basis, depending on the phase relationship between
the two signals. If the angle is constant across the band, a slight level
shift will occur. If the angle is not constant, but changes slowly with
frequency, passband distortion, gradually changing with frequency,
will be introduced. This can be compensated for through the use of
simple deviation equalizers designed to attend to the aggregate
distortion introduced by filters and other apparatus. If, however, the
angle of the reflected signal changes rapidly, as it would if the angle
of the filter output impedance changed in the region of resonance of a

crystal, then sharp, unequalizable notches could be introduced into
the jumbogroup passband. The reflected energy could cause variations
in passband loss as large as +0.27 dB for the relative levels indicated
above. This was, in fact, experienced in initial measurements of
jumbogroups 1, 2, and 38 receiving equipments, and required the
introduction of 9-dB pad-amplifier combinations to improve the return
loss of driving impedances for selected modulators, as indicated in
Fig. 3.

The jumbogroup 2 demodulator is functionally identical to that of
jumbogroup 1. It uses the same carriers and has filters with the same
passbands as the jumbogroup 2 modulator.

The jumbogroup 3 signal is demodulated in a single step; thus, all
three jumbogroups traverse four modulators during the modulation/
demodulation process. After level adjustment, the jumbogroup 3
signal is selected from the Ld line spectrum by a bandpass filter with a
passband between 43.572 and 60.556 MHz. The signal then goes
through a 9-dB amplifier and pad to provide isolation between the
filter and the following modulator. The jumbogroup 38 signal is then
demodulated with a 43.008-MHz carrier and, following amplification
and level adjustment, is sent through a low-pass filter to isolate the
basic jumbogroup signal. Following additional amplification, the signal
passes through a test pad that provides local test access.

A major and critical aspect of the design of an Fpm terminal involves
the ordering of apparatus and the settings of internal levels in the
modulator and demodulator subsystems. The output of a double-
balanced diode-ring modulator contains the dominant upper and lower
sidebands, carrier leak, and a multitude of other signals.? The energy
in either sideband is of a magnitude comparable to that of the energy
in the carrier leak. If the modulator output is fed through a bandpass
filter into an amplifier, then the undesired sideband and the carrier
leak can be attenuated so that the total signal power carried by the
following amplifier is greatly reduced. This tends to reduce the inter-
modulation distortion generated by the amplifier. It also tends to
degrade the effective noise figure of the amplifier by an amount
equivalent to the filter passband loss. Furthermore, should the signal

next be fed to a subsequent stage of modulation, the next modulator is
presented with a broad band of noise covering not only the message
band, but the image band as well. If the order of the amplifier and
filter is reversed, then the thermal noise problem is mitigated at the
expense of increased intermodulation distortion in the amplifier. The
more beneficial arrangement depends upon a number of parameters,
including filter loss, amplifier noise and intermodulation distortion,
modulator intermodulation distortion, and carrier leak. While pre-
liminary analysis of the gross characteristics of the various trans-
mission elements is helpful in establishing a starting point, the optimal
subsystem configuration can only be obtained through extensive, if
not exhaustive, examination of the various possible configurations.
This has been done for all modulator-demodulator subsystems using
noise-loading techniques.’°

It has been found that the setting of internal levels is equally
important. Signal and carrier levels also were optimized using noise
loading.

Jumbogroup signals are translated to their L5 line-frequency
allocation through multiple steps of modulation. The net frequency
translation they experience may be viewed as having been achieved
through a single modulation step using an equivalent carrier frequency
as follows. Consider the first step of modulation to have used a carrier
of frequency fi. Following the first step of modulation, a signal of
frequency f, would be translated to fi — f,. The second step of modu-
lation, using carrier frequency fe, would translate the signal to
fe -—fitf.. The net translation is f. = fe — fi. In jumbogroups 1
and 2, where two steps of modulation are used, the above applies.
For jumbogroup 3, a third modulation step, using a third carrier
frequency f3, is employed. In this case, the equivalent carrier is
fe =f3s+fe — fi. To demodulate the jumbogroup signals accurately,
only the equivalent carrier frequencies need be generated at the
receiver. This principle is used extensively in the mx. The equivalent
carrier frequencies are 2.560, 21.504, and 43.008 MHz for jumbogroups
1, 2, and 3, respectively. The frequency stability of these frequencies
is determined solely by the stability of the jumbogroup frequency
supply (sFs).’

The srs generates three reference signals at frequencies of 20.480,
2.560, and 1.024 MHz. The second and third reference signals are
obtained from the first through division of a 20.480-MHz signal by 8

and 20, respectively. These signals are further processed by the sync-
distribution circuits to yield the 2.560- and 21.504-MHz signals which
are fed to the carrier supplies. The carrier supplies, using the ap-
propriate input signals and locally generated signals, generate the
carriers for modulation and demodulation. The functional separation
between the srs and sync-distribution circuit is somewhat arbitrary.
The counters that provide the division by 8 and 20 operations could
have been located in the sync-distribution circuit, or much if not all
of the sync-distribution circuit could have been housed in the Jrs.
Since it may be feeding up to 20 smx bays, failure of the srs could be
catastrophic; therefore, the yrs should be as uncomplicated as possible,
with every effort made to maximize its reliability. The only exception
to this rule was the inclusion of the count-down circuitry in the Jrs.
These counters are implemented using high-speed emitter-coupled
logic. It was expected that locating such circuitry in the smx could
cause the generation of high-level spurious tones for which appropriate
shielding and filtering might prove unachievable.

The sync-distribution circuit is shown in Fig. 4; amplitudes are
indicated in dBm. Three reference signals at 1.024, 2.560, and 20.480
MHz are cabled to the input from the srs. The 2.560-MHz signal is
amplified, filtered, and passed through splitting hybrids to provide
three —5-dBm output signals, which are then cabled to the appropriate
(A or B) carrier supplies, where they are used as needed. The 20.480-
and 1.024-MHz signals are amplified, filtered, and mixed in a 17A
modulator to provide an output at 21.504 MHz. This signal is filtered,
amplified, and passed through splitting hybrids to provide three
—5-dBm output signals, which are cabled to the appropriate carrier
supply inputs. Level adjustments are provided as required. Local
test points are provided for the combined three-tone input signal, for
both output signals, and for both intermediate signals through 30-dB
bridging test pads. All these test points appear at jacks mounted in the
face plate of the sync-distribution circuit drawer.

The gmx bay is arranged to process three jumbogroup signals in any
arrangement; that is, it can process three jumbogroup | signals, one
each of jumbogroups 1, 2, and 3, etc. There is no fixed assignment of
the equipment to limit the flexibility with which the smx is used. Since
it is not known a priort which jumbogroup any given position will

handle, the sync-distribution outputs are fed to all locations. Those
which are required are used. Those not required are automatically
terminated when the carrier supply drawers are installed.

The jumbogroup 1 carrier supply terminates the 21.504-MHz input
signal, as shown in Fig. 5; amplitudes are indicated in dBm. This
carrier supply contains a temperature-compensated crystal oscillator
that generates a —15-dBm, 42.496-MHz signal whose frequency
stability is typically good to one part in 10°. As with all de-powered
apparatus used in the mx, this oscillator uses feed-through filters on
the battery leads to prevent coupling between circuits through common
de paths. The oscillator output is amplified and passed through a
splitting hybrid. One of the hybrid outputs passes through a fixed
pad, a level adjustment, a phase-adjust network, a low-current 14-dB
amplifier, a crystal bandpass filter, a 15-dB amplifier, a bridging test
pad, and a splitting hybrid whose outputs each experience further
noncrystal filtering to yield two 0-dBm output signals.

The phase-adjust network is provided to facilitate the matching
of the A and B side signals from the modulators (demodulators)
prior to manual operation of the protection switch. The match is made
to minimize the mean-squared error between the output signals to
achieve hitless switching. This will be covered in more detail later.

As indicated earlier, the spectral purity of the carrier signals and
isolation between carrier ports are of special importance. It may
appear desirable to reverse the order of the crystal filter and the
15-dB amplifier which follows it. The amplifier introduces both thermal
noise and harmonic distortion. However, the signal level into the
crystal filter would then be on the order of 10 dBm. The reliability
of a crystal degrades significantly when it is exposed to levels exceeding
0 dBm. Accordingly, the arrangement shown was used. While the
crystal filter provides sufficient rejection in the immediate vicinity of
the carrier, achieving sufficient rejection elsewhere requires further
filtering. This is accomplished through the use of noncrystal tc
filters placed in both output legs. This output arrangement provides
a net isolation between carrier supply outputs equivalent to the trans-
hybrid loss of the transformer plus twice the output filter loss.

The second output of the input splitting hybrid is mixed with the
2.560-M Hz input in a 17A modulator to yield a product at 45.056 MHz.
This signal is similarly amplified, crystal filtered, further amplified,
and passed through a 30-dB bridging test pad. The low-level output is
fed to a 6-dB splitting pad that provides two test appearances, one
local and one remote. The higher-level output is fed to a splitting

hybrid, where its outputs are further filtered to yield two 0-dBm
carrier signals.

The carrier signals are cabled to the appropriate modulators and
demodulators as indicated. On the transmitting side, the first modu-
lator uses the 42.496-MHz carrier, while the second uses the 45.056-
MHz carrier. Each of these carriers is only as stable as the stability
of the local oscillator. However, the effective carrier frequency, which
is the difference between the actual carriers used, is orders of magni-
tude better since it depends only on the stability of the srs. For
purposes of accurate demodulation, the stability of the local oscillator
is not important. Other considerations—namely, the matching between
filter rejection peaks and the location of the tones to be rejected—
require that the actual carrier frequencies do not wander too far from
their nominal values. The carriers must be at least as stable as a few
parts in 10°,

Functionally, the jumbogroup 2 carrier supply is very similar to
that for jumbogroup 1, as shown in Fig. 6. In this case, the 2.560-MHz
input is terminated and the 21.504-M Hz input is mixed with the locally
generated 70.144-MHz signal. For jumbogroup 2, the phase-adjust
network is located in the path of the input reference signal prior to
mixing with the locally generated signal, while for jumbogroup 1 the
phase-adjust network was located in the through path of the locally
generated signal. Functionally, either location is acceptable. The
relative positions shown were chosen because they facilitated network
realization.

Jumbogroup 3 requires three carriers for modulation and one
carrier for demodulation; the first two steps of modulation for jumbo-
group 3 are identical to those for jumbogroup 1. As shown in Fig. 7,
the 42.496- and 45.056-M Hz carriers are generated in the same way as
for jumbogroup 1. Since only one appearance of each signal is required,
no splitting hybrids are needed at the outputs. Following the second
step of modulation, the jumbogroup signal has a net translation of
2.560 MHz. The third step of modulation and the one step of demodu-
lation must both use carriers dependent solely on the srs. The desired
net translation for jumbogroup 3 is 43.008 MHz. This requires the
generation of a 40.448-MHz carrier for the third step of modulation,
and a 43.008-MHz carrier for demodulation. Both sync-distribution

signals are used. The 43.008-MHz carrier is obtained by doubling the
21.504-MHz input. The 43.008-MHz signal is then mixed with the
2.560-MHz input to form the 40.448-MHz carrier. Once again, all net
translations depend only upon the stability of the mrs. Two phase-
adjust networks are required for jumbogroup 3, one for modulation
and one for demodulation.

The jumbogroup regulator, shown in Fig. 8, compensates for level
variations that might be introduced in the jumbogroup passband by the
Ld line or terminal equipment between the remote transmitting BJaT
and the regulator input. A bridged-T variolosser, using indirectly
heated thermistors in the shunt and bridging arms, provides a level
control of +5 dB dynamic range. Control currents are automatically
adjusted in a complementary manner in accordance with the level of |
the received 5.888-MHz jumbogroup pilot. In the event that the
jumbogroup pilot is lost, the regulator automatically assumes a mid-
range setting. An operational amplifier, connected as an integrator,
provides high de gain in the control loop. This results in a residual
error of less than 0.01 dB over the full regulation range.

To facilitate maintenance, the jumbogroup regulator is equipped
with a manual reset feature. Operation of the reset switch locks the
regulator in the midrange state. This feature is particularly valuable
when noise loading tests are being made. An indicator lamp is lighted
when the regulator is manually placed in the reset state.

For each jumbogroup, both transmitting and receiving, A and B
paths are provided. These paths emanate from a splitting hybrid and
terminate on a 2 X 2 protection switch. It is the principal function of
the protection-switching eircuitry to monitor the integrity of the A
and B paths, to assure that a good signal is fed from the protection
switch when possible, and to activate alarms as appropriate. The
circuitry also provides for the control of restoration switches and the
manual operation of the protection switch in a hitless manner.

As shown in Fig. 9, both working and idle output jumbogroup pilot
signals are monitored by the protection-switching circuitry. The
switch control detector emits a de signal proportional to the level of
the jumbogroup pilot. On the transmitting side, the pilot frequencies
are 8.448, 27.392, and 48.896 MHz for jumbogroups 1, 2, and 38,
respectively; receiving, all jumbogroup pilots are at 5.888 MHz. If
the idle signal is lost for more than 0.1 second, a minor alarm condition
is indicated. If the working output is lost and the idle output is present,
the switch changes state. If this causes a good signal to be present at
the working output, and the signal is not present at the idle output,
a minor alarm is indicated. If both outputs are lost simultaneously for
more than 0.1 second, the switch does not change state, but a major
alarm is indicated. Alarms are indicated both in real time and with
memory. Switching is nonrevertive; that is, once a service-providing
condition has been established, the switch will remain in that state;
neither A nor B output is preferred. This arrangement avoids un-
necessary hits on jumbogroups that might be heavily loaded with
digital signals.

In the event that the working output appears lost while the idle
signal is present and operation of the protection switch fails to correct
the situation, no further switching occurs. Such a condition would most
likely be due to failure of the circuitry monitoring the working output.
A major alarm would be indicated.

Manual control of the protection switch is provided. If no trouble
condition is indicated, manual operation of the appropriate control
switch will cause the protection switch to change state. To facilitate

this operation in a hitless manner, a variable-sensitivity monitor of the
degree of match is provided. The idle and working outputs are sub-
tracted in a special hybrid transformer, yielding an error signal. This
error is amplified and detected, and the resultant de signal is fed to a
meter. The meter reading is indicative of the energy in the error signal.
Given that the gain of the A and B sides are closely matched, the
meter reading is indicative of the phase difference between the A and
B sides. Adjustment of the phase network in an associated carrier
supply corresponds to varying the phase, constant with frequency, of
one side relative to the other. By altering the phase network adjust-
ment to null the meter reading, the best possible phase match is
achieved. Since this procedure uses the actual message loading present
at the time, it results in the best match over the most meaningful
portion of the jumbogroup passband. Inability to reach a very low
meter reading is indicative of gain mismatch. This can be corrected
by making gain adjustments as appropriate, through the use of the
bridged test jacks provided on all modules.

Once A and B paths have been matched, the protection switch is
operated manually. In this mode of operation, the switch contacts are
sequenced in a make-before-break manner to avoid the momentary
open that might otherwise result.

If the manual operation would cause the protection switch to select
an apparently failed side, the control logic will not normally respond
to the manual command. In this instance, an additional, recessed
button must be depressed by the craftsperson to allow him to override
the inhibit feature of the control logic.

Restoration access switches are operated via the switch-control
logic. Upon command for restoration switch operation, the switch-
control logic locks the protection switch in its existing state. It then
operates those restoration switches associated with the unused side.
When the control logic is in this state, an indicator lamp is lighted
at the bay.

Frequency translation is achieved in the ymx through the use of a
double-balanced diode-ring modulator, the 17A. The important
performance parameters of such a modulator include conversion loss,
passband distortion, intermodulation distortion, carrier balance, and
signal balance.

Conversion loss, which gives rise to the difference in level between
the input signal and the desired output signal, has a theoretical mini-

mum value of 3.9 dB as a result of the partitioning of input energy
into the multitude of sidebands generated by the modulation process.
Additional loss is introduced principally through diode dissipation,
transformer loss, and whatever additional padding is introduced
owing to other considerations. While conversion loss can be offset by
adding appropriate gain in the amplifiers preceding and following the
modulator, the tendency of increasing conversion loss ordinarily is to
result in increased noise, either because levels are increased internal
to the modulator, or because the effective noise figure of the following
amplifier is degraded, or both.

Passband distortion is introduced principally through transformer
roll-off and circuit parasitics. Mutual coupling decreases with de-
creasing frequency, and shunt capacitance becomes dominant as
frequency is increased. Although transformer design can be optimized
for a specific frequency range, the design of a single transformer for
the multitude of input and output spectra encountered in ymx modu-
lators is particularly challenging. While passband distortion is usually
equalizable, prudent design requires that such distortion be minimized
at its source.

Intermodulation distortion is influenced by diode nonlinearities,
diode match, transformer and circuit balance, carrier waveform, and
signal level.? Intermodulation distortion may fall in-band or out-of-
band, relative to the desired sideband. Unlike most other parameters
of interest, in-band intermodulation distortion cannot be mitigated
once it is generated. Consequently, this type of distortion must be
controlled at its source.

Ideally, in a double-balanced modulator, neither input signal nor
carrier should appear at the output. Circuit imperfections preclude
realization of these objectives. Carrier balance and signal balance
depend on diode match, transformer balance, and circuit layout.
Typically, the carrier input level is 70 to 80 dB higher than the level
of the output signal. These relative levels are chosen because they
result in optimal intermodulation performance. Furthermore, Jmx
carriers run as high in frequency as 91.648 MHz. Consequently,
sizable levels of carrier signals may appear at the output. Since they
may subsequently fall within a message spectrum, such carrier leak
must be attenuated. While carrier leak is theoretically filterable,
rejection peaks well in excess of 100 dB are required. Furthermore,
unfiltered carrier leak may significantly increase the total power that
an amplifier directly following a modulator must handle. This can give

rise to further nonlinear distortion in the amplifier. Finally, any noise
entering the carrier port from the carrier-drive amplifier will appear
at the output to the extent that carrier balance allows. Should a
portion of this noise overlap the output message spectrum, it will
contaminate the signal in an unfilterable manner. Accordingly, good
carrier balance is critical, second only in importance to intermodulation
distortion.

Signal leak, by design of the modulation process (i.e., the selection
of modulation steps and carrier frequencies), is always filterable.
Nevertheless, signal leak at times gives rise to extremely demanding
distrimination objectives.

The 17A modulator is shown in Fig. 10. The center-tapped trans-
formers have a 75:300-ohm impedance ratio with a 7:7 + 7 turns
ratio. These transformers introduce less than a 0.05-dB passband
shape over any jumbogroup band. Transformer balance exceeds
55 dB.

Shottky barrier diodes are used. They provide essentially instan-
taneous switching at all jmx carrier frequencies, and appear to be
purely resistive up to 350 MHz. Their I-V characteristics are extremely
linear in the region running from 2 mA to 20 mA. In this region, their
resistance is about 40 ohms. Diodes are selected to yield matched
quads. Each quad has less than 15 mV mismatch at 2 mA and at 20 mA.

The 1-dB pads at both signal and carrier inputs were found experi-
mentally to result in improved intermodulation distortion, carrier and
signal balance, passband distortion, and return loss.

Circuit layout was optimized experimentally. Topological symmetry
was found to be critical. This was achieved, for example, by using four
manually inserted crossovers as opposed to partial use of printed-
circuit land connection.

The 17A modulator has a conversion loss of 6.2 dB, passband
distortion not exceeding 0.1 dB, less than 0 dBrnc0 of noise at optimal
signal levels, and, when driven by its associated carrier-drive amplifier,
signal and carrier balance of at least 40 dB.

The smx is housed in a shop-wired, shop-tested, 11-foot 6-inch,
unitized double bay. It is arranged to contain all the equipment to
process three jumbogroup signals. Each jumbogroup position is in-
dependent of other positions and contains transmitting and receiving
jack fields, transmitting and receiving switch-control logic units, and
redundant modulator and demodulator subsystems, carrier supplies,
regulators, equalizers, transmitting and receiving line interface units,
and detectors. Common equipment includes redundant sync-distri-
bution circuits at the lower left of the bay, and a summary alarm
circuit at the upper right. Sixteen dc-to-de converters are located at
the top of the bay, just above the fuse panels. Alarm cutoff and lamp
buttons are at the lower right just above the writing shelf. Redundant
battery feeds and srs inputs are provided. Several photographs of the
JMX equipment are contained in Ref. 11 of this issue.

No at-the-bay routine maintenance is planned for the mx. Rather,
each jumbogroup is automatically and remotely monitored by the
transmission surveillance system” associated with L5. Eleven critical
test points are provided for each jumbogroup position. These include
the transmitting input, the transmitting A and B outputs, the receiving
A and B inputs and outputs, the A and B regulator inputs, and two
carrier supply test points. These eleven access points are concentrated
in a remotely controllable 1 X 12 switch. Through this arrangement,
detailed analysis of smx performance is possible with little intervention
by a craftsperson.

All the above-mentioned test points, in addition to others, are also
provided at the bay. It is possible to verify the integrity of any trans-
mission-related module without removing it from the bay, through the
use of the test points provided.

Verification of the integrity of the many status and alarm indicators
is facilitated through the lamp-test feature. Depressing the three
lamp-test buttons allows the location of any failed lamp in seconds.

The gmx has undergone extensive field evaluation. Both functionally
and operationally, the equipment operated superbly. The poorest
jumbogroup noise performance obtained was 12 dBrnc0.

The poorest equal-level coupling loss obtained was 90 dB. This was
measured between A and B sides of the same transmitting or receiving
jumbogroup, and was due to coupling through the subtracting hybrid
transformer of the circuitry used to evaluate the degree of match
between the associated A and B sides. This is a well-controlled coupling,
not expected to vary from system to system. Most other forms of
coupling were so low as to be unmeasurable.

Extensive testing of regulation and protection switching showed
these features to operate without fault.

Ten gmx bays were constructed by Western Electric for use in the
Lillyville, Pennsylvania-Hillsboro, Missouri route. This equipment
has been providing commercial service since January 1, 1974.

Taking into consideration the bandwidth of the signal it processes
and the frequencies at which it operates, the sx is the quietest analog
multiplex terminal ever designed. The success of the design is a tribute
to the technical excellence and total dedication of its design team
which included J. S. Young, T. B. Merrick, G. W. Kattke, B. B. Garg,
S. J. Davis, A. G. Favale, and W. D. Radwill.

1. F. C. Kelcourse and F. J. Herr, “L5 System: Overall Description and System
Design,”’ B.S8.T.J., this issue, pp. 1901-1933.

2. W.G. Albert, J. B. Evans, Jr., T. J. Haley, T. B. Merrick, and T. H. Simmonds,
Jr., ‘“L4 System: Terminal Arrangements,” B.S.T.J., 48, No. 4 (April 1969),
pp. 993-1040.

3. E. H. Angell, Y.-S. Cho, K. P. Kretsch, and M. M. Luniewicz, “L5 System:
Repeatered Line,” B.S.T.J., this issue, pp. 1935-1985.

4. J. H. Green and R. W. Sanders, “L5 System: Line-Protection Switching,”’
B.S.T.J., this issue, pp. 2011-2034.

5. R. E. Maurer, ‘“Terminal and Main Station Functions,’’ IEEE Transactions on
Communications, COM-22, No. 2 (February 1974), pp. 226-229.

6. R. K. Bates and D. J. Zorn, “L5 System: Signal Administration and Inter-
connection,” B.S.T.J., this issue, pp. 2129-2145.

7. J. F. Barry, S. Narayanan, and J. F. Oberst, “L5 System: Jumbogroup Fre-
quency Supply,” B.S.T.J., this issue, pp. 2109-2127.

8. J. L. Garrison, A. Olsen, Jr., and T. H. Simmonds, Jr., ‘“L5 System: Transmis-
sion Networks and Magnetic Components,” B.S.T.J., this issue, pp. 2203-2248.

9. R. BE. Maurer, ‘Analysis of Intermodulation Distortion in Double Balanced
Diode Ring Modulators,’”’ IEEE 1973 Int. Conf. Commun., Conf. Record, II,
June 11-13, 1973, pp. 51.21-51.26.

10. B. B. Garg, “FDM Synthesis Using Noise Loading,’? IEEE 1974 Int. Conf.
Commun., Conf. Record, June 17-19, 1974, pp. 7F-1 to 7F-3.

12. J. L. Thomas, R. E. Anderson, and P. J. Baun, ‘‘L5 System: Centralized Trans-
mission Surveillance,” B.S.T.J., this issue, pp. 2035-2064.

Copyright © 1974 American Telephone and Telegraph Company
Tue Bey System TECHNICAL JOURNAL
Vol. 53, No. 10, December 1974
Printed in U.S.A.

The 39A precision oscillator is designed to operate at 40 watts from a
24-V +10-percent power source. The reference frequency can be set digitally
to 6.12 MHz +8 X 10 over a tuning range of +4 X 107". The output
signal 1s a 1-mW sinewave into a 100-ohm load. This performance is
assured over a temperature range of 0 to 60 degrees Celsius. Oscillator out-
line dimensions for length, width, and height are 35.8, 22.1, and 21.8 cm
respectively.

The 39A oscillator was developed to meet the precise frequency-
control requirements of the L5 jumbogroup frequency supply. The need
for frequency control as exemplified by the 39A oscillator is explained
in the paper on the jumbogroup frequency supply in this issue. This
5.12-MHz precision oscillator is characterized by long-term stability
of <1 X 10- per day, and by short-term stability of <1 X 10-8 for
a 1-millisecond sampling time or <2 X 10~" for a 10-second sampling
time. External frequency control is accomplished utilizing a compatible
digital word applied to the oscillator’s 14-bit, TTL, precision, digital-to-
analog converter. Setting range of the converter is >+4 X 107’,
which gives a least-significant-bit resolution of 5 X 107"; the con-
verter section incorporates a linearizing network to reduce the fre-
quency-set deviation from a straight-line function to <+5 X 107"
over a +4 X 107° tuning range, and to a maximum variation of
tuning sensitivity of +30 percent over the entire range of +4 X 107’,
with less than +1 X 10~° departure from a smooth curve.

Il. GENERAL DESCRIPTION
The 39A oscillator can be separated into three major subassemblies:

circuit, a tuned feedback amplifier detector circuit (aac)
functioning to provide constant low-level crystal-current
drive, and an output amplifier instituting sufficient load isola-
tion for the output signal.

A thermal and electrical noise-isolation subassembly which is
composed of temperature-control circuits, power-supply regula-
tors, and redundant electrostatic shielding amply protecting
all major circuitry.

The digital frequency-control subassembly with a precision
digital-to-analog converter and a frequency-linearizing net-
work for remote steering of output frequency.

crystal-oscillator circuit in the 39A oscillator is the modified
commonly used in precision oscillators for many years.)?

varactor diode in the crystal network is used to control the operating
frequency of the network. Earlier circuit designs have varied the
oscillator-transistor de operating point to control the loop gain, but
here low-noise performance is achieved by fixing the de operating point
of the oscillator transistor and adjusting its positive loop gain. Oscilla-
tor loop gain is controlled by varying the forward bias of variolossers
CR; and CR4,, which controls the degeneration of this transistor stage.

The acc amplifier circuit is shown in Fig. 2. This is a three-stage
direct-coupled feedback amplifier selected for low noise, simplified
tuning, and having a minimum number of components. Passband
tuning is accomplished by an uc network in the feedback path. This
amplifier operates with 38-dB in-band gain and has a 3-dB bandwidth
of about 200 kHz. The rectified output from diodes CR: and CRez is
used to control the oscillator gain for a crystal-operating current of
100 to 200 wA.

The output signal of the acc amplifier is coupled to the final amplifier
circuit by a variable attenuator which is used to adjust the oscillator
output level to 1 mW into a 100-ohm load. The design of the output
amplifier is essentially the same as the acc amplifier.

The crystal oscillator, aac amplifier, and output amplifier combine
to make up the oscillator subassembly shown in Fig. 3. The unit at

this stage in fabrication can be production tested as a functional sub-
assembly. These tests include a thorough mechanical inspection, a
test of acc and output-level, and a check on harmonic distortion.
The long-term frequency of this precision oscillator is determined
primarily by changes of the crystal unit and directly associated Lc
components. In this respect, these components have been selected for
minimum long-term drift. Short-term frequency variations (times of
less than 0.1 second) can be attributed to transistor noise and small
fluctuations in crystal current. The effects of crystal-current variations
have been reported’ and are typically 1 X 10-" per 0.02 dB for current
levels in the range of 100 to 200 wA at a frequency of 5.12 MHz.

The 5.12-MHz crystal units*:® used in the 39A oscillator operate at
76°C minimum to 81°C. At this operating temperature, the frequency
dependence on temperature is approximately 7 X 10-°/°C if the operat-

ing temperature of the crystal is <-£0.25°C from the frequency turn-
ing point. In addition to the effects of long-term temperature variations,
short-term changes in temperature cause perturbations in frequency
due to thermal gradients within the crystal. A 1 X 10-" frequency
change results from a 4 X 10-°°C/hour rate of change in crystal
temperature.

To reduce frequency variation effects due to ambient temperature
changes, the oscillator subassembly is housed in a two-stage tempera-
ture-stabilized oven. This two-stage oven has an ambient ratio of
> 105:1 resulting in short-term temperature variations being measured
as <l X 10-*°C per hour. The outer oven reduces ambient effects on
the temperature-control and voltage-regulator circuits to <5 X 10-2°C.
Two identical voltage regulators are used in the oscillator to reduce

the effects of power-supply variation on reference voltages, RF circuits,
and control circuits. Figure 4 shows the environmental isolation
subassembly.

I'requency control is accomplished by a varactor diode in the
crystal circuit. The precision voltage supplying this diode is derived
from the precision digital frequency-control subassembly shown in
Fig. 5. This subassembly consists of a 14-bit digital gate network,
with strobing option, fabricated from standard Tru logic. The network
contains latching circuits, relay drivers, and relays (5 V dc) controlling
a constant-impedance voltage divider. Figure 6 gives the basic circuit
diagram for this digital-to-analog converter.

Figure 7 is a simplified schematic of the divider excluding the control
logic. Resistors Rvi, Rye, and Ruz (or Rae if required) are selected by
computer program at time of manufacture to precisely give the
required nonlinear binary-voltage allowing a linear binary-frequency
function of oscillator output. The loading resistor, Rumi or Re, de-

termines the network output voltage function and is selected for each
oscillator. A typical set of output-voltage characteristics for this net-
work are shown in Fig. 8 for values of Rai or Rue from 20,000 ohms
to © plotted as output voltage versus ratio of the programmed binary
divider.

Figure 9 shows the tuning characteristics of a typical 39A oscillator
over the frequency range from F,, (1 — 4 X 10-8) to F, (1 + 4 X 107-8)
and plotted relative to the frequency setting. The mean slope achieved
over this range is within a few percent of the objective, with slope
uniformity within about +10 percent over the range. Over the entire

Fig. 8—Typical digital-to-analog converter output-voltage characteristic.

tuning range of +4 X 107’, slope variations are no more than +30
percent, and deviation from a smooth curve is no greater than
+1 X 107.

The circuit also provides an LED indicator for each input bit, which
allows a visual check of the frequency-setting word. This, in addition
to an electrical output provision for automatically sensing the tuning
status, gives a convenient check on the complete functioning of the
digital-to-analog converter.

The overall electromechanical design makes maximum use of current
instrument-fabrication methods. All printed-circuit boards are of a
plug-in configuration simplifying assembly and testing by a large
factor. Hand wiring has been held to a minimum throughout the entire
assembly. Polyurethane foam both isolates and supports the inner and
outer oven assemblies. RFI or constant K bandpass filtering has been
incorporated between major subassemblies where deemed necessary.

Figure 10 illustrates the entire oscillator disassembled to show inner
and outer ovens plus the inner and outer card frames. The front cover
is mechanically attached to the digital-to-analog converter. The two
connectors seen on the precision digital-to-analog converter circuit-
board serve to provide a major part of the wiring interconnecting the
rear input panel and the oscillator circuitry. The right front vertical
edge of the external housing is designed to allow viewing of the LED

indicator. All connections are made through the rear oscillator panel
to the digital-word inputs, strobe pulses, 5- and 24-V de power inputs
and to the sense and rF outputs.

The 39A oscillator has been developed as a secondary frequency
standard and is characterized by long-term stability of better than
1 X 10~°/day, and short-term stability of better than 1 X 10-8 for a
1-millisecond sampling time and 2 X 10-" for a 10-second sampling
time. The total range of digital-to-analog frequency control is
+4 X 10-7 from a nominal 5.12 MHz having slope variations < +30
percent with a maximum deviation of any point from a smooth curve
of less than 1 X 10-'°. Midrange frequency control has a nominal
straight-line tuning characteristic of 5 X 10-"/bit, with an average
slope tolerance of +10 percent and maximum deviation for a smooth
curve of less than +5 X 10-". This oscillator has been designed to
minimize the effects of thermal, mechanical, and electrical noise to a
high degree and is fabricated using quality hardware and reliable
components.

1. W. L. Smith, ‘Miniature Transistorized Crystal-Controlled Precision Oscillators,’
IRE Trans. on Instrumentation, J-9, No. 2 (September 1960), pp. 141-148.

2. H. S. Pustarfi, “An Improved 5 MHz Reference Oscillator for Time and Fre-
quency Standard Applications,’ IEEE Trans. on Instrumentation and Mea-
surement, [M-15, No. 4 (December 1966), pp. 196-202.

3. T. C. Anderson and F. G. Merrill, ‘“Crystal-Controlled Primary Frequency
Standards: Latest Advances for Long-Term Stability,” IRE Trans. on In-
strumentation, J-9, No. 2 (September 1960), pp. 136-140.

4. A. W. Warner, “Ultraprecise Quartz Crystal Frequency Standards,” IRE Trans.
on Instrumentation, [-7, Nos. 3 & 4 (December 1958), pp. 185-188.

5. A. W. Warner, ‘Design and Performance of Ultraprecise 2.5 MC Quartz Crystal
Units,” B.S.T.J., 39, No. 5 (September 1960), pp. 1193-1218.

Copyright © 1974 American Telephone and Telegraph Company
Tue Bey System TEcHNICAL JOURNAL
Vol. 53, No. 10, December 1974
Printed in U.S.A.

The jumbogroup frequency supply (srs) provides accurate and reliable
synchronization signals to L& jumbogroup multiplex carrier supplies.
The srs output-signal frequency is held to within several parts in 10" of
an input reference frequency. This extraordinary accuracy is achieved by
digital frequency control of a precision quartz oscillator. The excellent
reliability required when the srs serves over 100,000 two-way voice circuits
as achteved by redundant power and signal feeds, one-for-one protection
of key circutts, and automatic protection switching.

In addition to its L& application, the srs will also serve as a regional
frequency supply in a new synchronizing frequency distribution network
that will upgrade the entire Bell System transmission plant.

It is the goal of the Bell System to maintain frequency offset in any
transcontinental circuit to within +2 Hz. This requirement is imposed
by the most sensitive message signals, such as those in program-
ming, telegraph signals, and data signals. In an analog amplitude-
modulated system, such a frequency offset can be introduced into the
message signal if the transmitting and receiving multiplex carriers are
not synchronized. In a single-sideband suppressed carrier multiplex
format as used in jumbogroup multiplex! terminals (Jmx), the multi-
plexing carriers are not transmitted. Thus, precision frequencies must
be provided for the multiplex terminal at each geographical location.

The top channel frequency of the long-haul, analog, frequency-
division multiplex system has been steadily increasing. With the in-
crease in top channel frequency (F), the frequency precision required
of the multiplex carriers increases. This is because the frequency offset

(AF) is limited to 2 Hz, whereas the ratio AF’/F decreases as F in-
creases. In the L5 coaxial system, the top channel frequency is at 60
MHz, and the 2-Hz requirement results in a ratio of 3.3 parts in 108
(3.3/108). Because a typical signal may traverse through several ymx
terminals, carrier frequencies must be synchronized with an accuracy
of about 1 part in 10%.

The frequency precision demanded by the L5 system could not be
provided by the synchronizing network existing at the time L5 design
work was initiated. Several alternatives were studied. One alternative
was to provide a cesium frequency standard at each multiplex location.
The cost of the cesium standard, the need for providing at least two
such standards for reliability at each location, and the need for training
personnel to maintain them made this option unattractive. An alterna-
tive was to provide a primary frequency standard at a central location
and transmit reference-frequency signals to different locations. Such a
primary frequency standard, called the Bell System Reference Fre-
quency Standard? (BsrFs), has been designed. It comprises three
cesium standards and provides highly accurate reference signals
(better than one part in 101).

At the receiving location, several alternatives were open. One option
was to use a precision voltage-controlled crystal oscillator, whose fre-
quency drift could be manually corrected periodically. The manual
control implied trained personnel and periodic maintenance. Moreover,
frequency-control circuitry is required for alarming and switching
purposes in case of oscillator failure. Thus, it was decided to control
and monitor frequency automatically, and the precision frequency
supply so designed (the jumbogroup frequency supply, or Fs) is the
subject of this paper.

The reference-frequency signal, even though highly accurate, may
suffer short term impairment like noise, switching transients, etc. It
may even be lost completely due to transmission difficulties. Because
of the large number of circuits (more than 100,000) dependent upon
one JFs, it is imperative that precision frequencies be supplied to the
JMx even when the reference signal is lost or impaired. To accomplish
this objective, a precision crystal oscillator capable of accurately free-
running for a short period of time is employed in the srs. The precision
oscillator is loosely coupled to the reference signal such that it can
automatically correct itself when the unimpaired reference signal is
present and disconnect from the reference signal if the reference signal
is impaired or lost. A digital memory is provided in the digital fre-
quency-control circuit so that the oscillator can remain at the correct

frequency even when the reference signal is disconnected. The precision
oscillator and the precision digital frequency-control circuits are housed
in a unit called the jumbogroup frequency generator (JFG).

An important consideration in the design of the srs is reliability.
More than 100,000 two-way voice circuits may depend upon one JFs.
To ensure reliability, automatic protection switching is provided for
its key circuits. Several visual alarms are provided so that preventive
action can be taken before protection switching is needed. The working
of the frequency-control circuitry can be checked without any inter-
ruption of service. Redundant power feeds connect each srs to the
office battery and two cables are provided to each mx.

Human engineering has been given careful consideration. A precision
test unit to measure accurately the difference between two signals has
been incorporated in the srs. This will enable the craftsperson to
troubleshoot the bay if necessary. A manual patching arrangement and
manual override have been provided to ensure that the craftsperson
can take full control of the bay if needed.

The srs furnishes three precision frequencies to the JmMx carrier
supply, where further processing occurs to generate all the carriers
needed for frequency multiplexing. The three frequencies are at
20.48 MHz, 2.56 MHz, and 1.024 MHz. An overall block diagram of
the srs is shown in Fig. 1. It consists of three Jra@’s arranged in a master-

slave configuration, two frequency dividers, switch and alarm circuitry,
two output buses, and a precision frequency-measuring test unit.

The master sra directly receives the 20.48-MHz reference signal.
In some locations, the reference signal will be at 2.048 MHz; a fre-
quency converter has been designed to convert 2.048 MHz to 20.48
MHz. The master Jra compares its precision-oscillator frequency with
the incoming reference signal and corrects its oscillator based on the
long-term average of the frequency difference. The master src output
signal is at 20.48 MHz and is fed to the regular and standby sra’s.

The incorporation of three sr@’s in a JFs ensures detection of a JFe@
failure even when the reference signal is noisy or absent. Such a detec-
tion would quickly initiate alarm and switching circuits to take ap-
propriate action. If only two sre’s were provided in a srs, the failed
JF@ could not be isolated if the failure occurred when the reference
signal was absent or noisy.

The regular and standby sre’s use the output of the master sre as
their reference. In this master-slave arrangement, the input signal to
the regular and standby sre’s should be good all the time unless the
master sFG fails. In contrast, the master sra reference signal, while
highly accurate when present, may be absent or noisy at times. For
this reason the master sFa does not produce frequency alarms unless
the frequency offset exceeds 1/10® and lasts at least two hours; the
regular and standby sra@’s produce frequency alarms if the offset
exceeds 2/10° and lasts for several hundred milliseconds (see the sre
description in Section III).

As shown in Fig. 1, the output of the regular yra (20.48 MHz) is fed
to the regular frequency divider; similarly, the standby sre feeds the
standby frequency divider. The function of the frequency divider is to
generate 2.56-MHz and 1.024-MHz signals and combine all three
frequencies (20.48 MHz, 2.56 MHz, and 1.024 MHz) together for
transmission on one coaxial cable. The output of the regular divider
is transmitted to the two output buses through two coaxial switches.
The coaxial switches are operated by the alarm-and-switch-control
unit which processes information from the three jre’s and the level
detectors. The output bus has 32 taps, and because of reliability con-
siderations one JFs may serve up to 22 Jmx’s. Each smx is connected
by two cables, one from bus A and the other from bus B. Thus, the
regular divider feeds both sides of the sux. The output of the standby
divider is fed to the two coaxial switches. If a regular sra@ or frequency
divider fails, the alarm and switch control unit will operate the coaxial
switches so that the standby sre and divider will supply signals to

both sides of the ymx. A phase comparator is provided to compare the
phases of signals from the regular and standby dividers; this enables
the craftsperson to manually transfer output signals from the regular
gra to the standby sre with minimal disturbance.

Several features have been included in the srs to facilitate trouble-
shooting and maintenance. A precision frequency-measuring test unit
has been designed and incorporated as a part of the srs. A patch panel
has been included to provide a convenient and non-service-affecting
way to rearrange the remaining JFe’s if a ura fails. Several test points
and a level meter have been provided. The coaxial switches can be
patched around without affecting service; the alarm and switch control
circuit could then be exercised for routine checkup of the coaxial
switches and control circuitry.

The srs is shown in Fig. 2 in a double 9-foot bay. A 7-foot-bay ar-
rangement is also available. At the top of each side is a distribution
bus; these buses provide 32 redundant (64 total) outputs. The three
grq@’s are on the left side with the master on top and below it the
regular and standby jrae’s with the ura patch panel between them.
Each sre has its alarm display panel above its oscillator panel on which
the 14 bits of the oscillator frequency-control word are displayed. On
the right side, from top to bottom, are shown: three sets of de-de con-
verters, test unit and associated dc-dc converters, regular frequency
divider, phase comparator, standby frequency divider, and alarm and
switch control unit.

Most of the srs circuitry consists of digital integrated circuits (1c’s)
and various discrete components mounted on two- or four-layer

printed-wiring boards. Additional apparatus such as thin-film, hybrid,
integrated-circuit amplifiers and crystal filters are also employed.

The wide use of digital 1c’s in the Jrs made it possible to achieve
sophisticated control and alarm functions that would be impractical
with discrete components. Western Electric-manufactured 101-type
emitter-coupled logic (EcL) was chosen for reasons of speed and
availability. Since only small-scale integration (sst) members of this
logic family were available during the srs design phase, a large number
of “‘chips’’ was necessary to implement all srs functions—about 1100
logic chips in a full yrs. A typical two-sided printed-wiring board is
shown in Fig. 3.

The sre is discussed in detail in the next section. The frequency
divider, protection switching, and test unit are discussed in the suc-
ceeding sections.

The sFa produces an accurate 20.48-MHz sinusoidal output signal
with a AF/F normally less than 1/10" relative to its input signal. This
small AF/F is achieved by combining an extremely stable crystal
oscillator with precision frequency-control circuitry. Frequency ac-
curacy is provided by reference to a signal whose ultimate source is
the BSRFs.

The oscillator is the 39A digitally controlled crystal oscillator.’ It
has a drift rate less than 1/10" per day, low enough to operate without
correction for several weeks and still meet srs frequency-accuracy
requirements. This oscillator’s frequency is controlled by a 14-parallel-
bit binary-control word developed by the sre frequency-control circuit.
The control word is changed when comparison of the JFe¢ input and
output signals indicates a need to correct the oscillator frequency.

Figure 4 is a block diagram of the src showing the relationship be-
tween oscillator, input, output, and frequency-control circuits. The
39A digitally controlled crystal oscillator (pcxo) has a 5.12-MHz
output which is multiplied to 20.48 MHz in the sre output circuit. The
20.48-MHz input signal, after being filtered and amplified in the sra
input circuit, is compared with the 20.48-MHz output signal in the
frequency-control-and-alarm circuit, which produces the frequency-
control word and frequency alarms. The logic-interface circuit processes
frequency, level, and voltage alarms generated within the sre for
transmission to the srs alarm-and-switch-control circuit and for dis-

play on the sre itself. This alarm processing is controlled by a status
input from the srs alarm and switch control.

The major consideration in designing the Jra was the character of
the signal supplied to the srs as a frequency reference. As mentioned
earlier, this signal could be intermittent, noisy, or absent due to trans-
mission difficulties. Furthermore, at the time of Jra development, the
proposed reference-signal-transmission path contained phase-locked
oscillators that could introduce frequency offsets as great as several
parts in 10° if they lost phase-lock. These phase-locked oscillators were
not used in the final Bsrrs distribution network. Because of these
reference transmission difficulties, the JrF@ was designed to rely on
only one characteristic of the reference signal—its average frequency
as measured during times of good transmission.

The low drift rate of the 389A pcxo allows a sre to ignore the refer-
ence during times of poor transmission. Although this drift rate is low,
it is not zero and cannot be ignored. In the sra the pcxo frequency-
control word is held constant until the average of a large number
(256 minimum) of short (16 seconds), valid frequency comparisons

between input and output signals indicates a frequency offset greater
than 1/10'°. Then the control word is changed to move the pcxo
frequency 2/10" in a direction opposite to the offset. This frequency-
control operation can be separated into three functions: frequency-
offset detection, averaging, and frequency correction.

Figure 5 shows, in general terms, the basic portion of the circuit by
which the ura detects small values of AF/F between two frequencies
A and P. In the first portion of this circuit, divider D and the two
mixer-bandpass filter circuits develop a frequency A/D + AF. The
final portion of the circuit consists of a frequency divider N whose
output controls a gate, allowing the frequency A to be counted for one
gate period, resulting in the count C. C, is the count when AF = O.
The count AC contains the AF/F information obtained in the time T.

Implementation of this circuit involves the selection of values for
D and N and the number of stages in the counter. To maximize the
AC/T ratio, D is made as large as possible within the constraints im-
posed by the bandpass filter requirements and the short-term frequency
stability of the signals. In the ura, D = 320; hence, A/D = 64 kHz.
N is selected to provide the necessary precision (AC) in a reasonable
time (7). N is also considered in the counter design. A counter large
enough to count to C would be larger than necessary. In the sre fre-
quency-control circuit the largest AF /F that must be measured by the
counter is 1/108; this limits AC. If one selects N and the number of

counter stages so that after multiple overflowings C, leaves a remaining
count of all zeros, the number of counter stages need only be sufficient
to detect the sense and magnitude of AC. In the sra, N = 2”° and the
counter has 19 stages. This results in a circuit capable of detecting
sense and magnitude of frequency offsets in the 1/10" to 2/10 range.

Figure 6 shows the application of this circuit in the src. During the
16.384-second gate period provided by cate 1, the count signal (the
JFG output frequency) is counted. At the end of that period, AC is
examined and if it indicates a AF/F > 1/10", high or low, a pulse is sent
to the high or low input of the accumMULATOR. The INHIBIT 2 signal will
interrupt this process and reset the circuit should other, quicker-
acting circuits determine that the frequency comparison is not ac-
ceptable for frequency control.

Averaging of the results of the 16.384-second frequency comparisons
is done in the accumuLatTor. This circuit is a nine-stage up-down
binary counter. Starting from its midrange, or reset, position, this
counter counts up or down depending on the sense of AF/F. When this
counter has received 256 more pulses in one direction than in the other,
it sends two pulses to the memory by way of the FREQ CONT MODE
SELECTOR.

The normal position of the FREQ CONT MODE SELECTOR will connect
the ACCUMULATOR to the Memory. The memory provides the 14-bit
frequency-control word to the 39A pcxo. In the 39A a D/A converter
plus varactor diode circuit provide a linear relationship between
oscillator frequency and digital control word. The least significant bit
(usB) of the control word corresponds to AF/F = 5/10"; so the 14-bit
word provides a frequency-control range of +4/10’. The mMmEmMoRyY
keeps the LtsB at 0 and increases the higher bits by one for each pulse
it receives. Thus, the two pulses from the ACCUMULATOR result in a
2/10 frequency change.

The maximum correction rate of the frequency-control circuit is
2/10!° per 256 frequency comparisons made at a rate of one every
20.48 seconds; this converts to 3.4/10° per day.

Figure 6 shows a number of frequency alarms being generated. The
circuits generating these alarms are interconnected to utilize the inverse
relation between the value of AF'/F and the time needed to detect it.

This is the coarsest frequency alarm and utilizes the 30-Hz band-
width of the 64-kHz bandpass filter. At the output of this filter a AF
of 15 Hz or greater causes a 3-dB or greater drop in signal level. This
drop corresponds to a AF /F of 7.5/10" or greater. A level-detector
circuit, DET, detects this and produces the 1/10° aum and iNursIT 1
signals. (The 1/10° designation is nominal.)

GATE 2 is called the frequency-alarm gate, and the circuits generating
it and using it comprise the frequency-alarm circuit. This circuit uses
the frequency offset detection scheme shown in Fig. 5. N = 2 and the
counter has 12 stages; so the circuit has a range from 1/10° to 2.5/108.
The 1nHIBIT 1 signal interrupts and resets this circuit which makes
a AF/F measurement every 320 ms. The main function of this circuit
is to produce the 1/108 nz atm and 1/108 Lo aLm and shut down the
frequency-control circuit when AF/F > 1/108.

Since the counter used in the frequency-control circuit can detect
1/10® offsets, it is also used to produce the 1/108 nr aum and 1/108
LO ALM. This provides protection in case of frequency-alarm-circuit
failure.

The 2/10° nr and to alarms are produced in the frequency-control
circuit. These alarms alert maintenance personnel to problems before
they affect service.

A frequency drift less than 3.4/10° per day can be compensated for
in the gra. An oscillator with a drift rate greater than 3.4/10° per day
will eventually produce a frequency alarm. An oscillator with a lower
drift rate will not cause such an alarm and no action would be taken
until the oscillator reached the end of its control range and had to be
replaced. The pRiFT ALARM forestalls this. The DRIFT ALARM circuit
notes each frequency-correction pulse from the ACCUMULATOR, and
generates the DRIFT ALM signal when the memory has received 64 more
pulses in one direction than in the other. The DRIFT ALM signal notifies
maintenance personnel, who manually reset the circuit and record the
alarm. The record of prirT ALM times and oscillator frequency-control
words indicates oscillator drift rate.

If the oscillator is at the end of its control range; i.e., its control
word is all ones or all zeros, the MEMorRy produces the END OF RANGE
ALM and inhibits itself from further changes. The oscillator must then
be replaced. However, long before this happens, maintenance personnel
should have predicted it from the drift-alarm records.

Most functions within the Jr@ are automatic, but installation and
maintenance operations must be performed manually. The sre design
has tried to facilitate these operations.

At time of installation, or following a loss of power, the oscillator
frequency might need to be changed so much that the 3.4/10° per day
rate would be far too slow. Shown in Figs. 6 and 7 are three features
to provide more rapid frequency correction. First, there is the MEM
RST switch which can put the frequency control word in its midrange
position. Second, there are the 1/10" up and pown switches which
change the oscillator frequency by 1/10" with each operation. Third,
there is the speed-up circuit which uses the 320-ms period of the

frequency-alarm circuit to correct the oscillator at a 6/10 per second
rate. To operate the speed-up circuit, the FREQ CONT MODE switch is
moved from the Nor to SPEED-UP position and then the SPEED-UP
pushbutton switch is operated. The frequency-alarm circuit automat-
ically stops sending pulses to the MEMoRyY when the AF’/F is reduced
below 1/10°. Figure 7 shows that all these controls are located behind
covers to forestall their misuse. Note the angle bracket on the cover
of the FREQ CONT MODE switch; this bracket forces one to return the
switch to its Nor position before the cover can be closed.

Intermittent alarms have always been the bane of maintenance
personnel. In the src, alarm indications are held until they can be
noted or until their cause has been corrected, whichever comes later.
Figure 7 shows the Atm rsT switch which will extinguish an alarm lamp
if the trouble has gone. If the trouble still exists, the lamp cannot be
extinguished. An exception to this lock-up is that all alarms associated
with the srs reference signal do not lock up.

A failure of the frequency-alarm circuit could go unnoticed. The sre
provides a simple way of checking this. Operation of the FREQ ALM
TEST switches, HI or LO, artificially shortens or widens the 256-ms gate
by 1.6 ms, causing the effect of a 1/108 ur or xo offset. If the circuit
detects this, a green lamp lights; if it does not, the red rain lamp lights.
No other indications are generated in this operation. But if one
desires to check the interaction of this alarm circuit and the srs alarm
and switching circuits, operation of the yrs ALM & Sw TEST switch
(see Fig. 7) simultaneously with the H1 or Lo switch will allow 1/10°
alarms to be transmitted from the sre to the srs. Note that these
operations do not disturb the sra@ output frequency.

As shown in Fig. 1, the sFs contains two frequency dividers. Called
the regular divider and standby divider, these units are connected to
the outputs of the regular yra and standby sre, respectively. Each
frequency divider provides as an output 1.024-MHz, 2.56-MHz, and
20.48-M Hz signals multiplexed onto one coaxial cable.

Figure 8 is a block diagram of the basic circuit showing that phase
control is provided for all frequencies. The 20.48-MHz phase-adjust
networks provide a 180-degree control range. The fine phase adjust-

ment for the 1.024-MHz and 2.56-MHz output signals is done at 20.48
MHz to avoid component problems associated with lower-frequency-
phase-adjust networks. The 180-degree phase adjust at 20.48 MHz
yields 22.5 degrees at 2.56 MHz and 9 degrees at 1.024 MHz. Coarse-
phase-adjust circuits are included in the two dividers to increase the
range of phase adjustment at the two lower frequencies. Each divider
skips a single input pulse for each push of its associated pushbutton,
thus producing a step change in its output phase. From each divider,
the test outputs, shown in Fig. 8, are connected to the phase-compara-
tor panel, which provides one meter for each frequency. A meter null
is an indication of phase coincidence. All three meters must be nulled
to indicate that the signals from the two dividers are in phase. In
adjusting phase to produce these nulls, the coarse-phase adjust of the
in-service divider should not be used.

Protection switching plays a vital role in ensuring the continuity
of sFs output signals. Figure 9 shows the switching arrangement; the
switches are fabricated with coaxial, dry-reed contacts. The normally
closed contacts of these switches are magnetically biased so there is no
need for switch coil current while the switch is released. Since the
switch is normally released, a loss of switch power will normally not
affect the srs output signal. As shown in Fig. 9, with the switches

released, the regular divider feeds detectors A and D and output
buses A and B, while the standby divider feeds detectors B and C.
When the switches are operated, the standby divider feeds detectors
A and D and output buses A and B, while the regular divider feeds
detectors B and C.

The switch-control circuit energizes (to operate) or deenergizes
(to release) the switch coils in a ‘‘make before break’’ sequence. This
feature has little importance if the switching is being done automat-
ically due to a signal fault. However, if the switching is being done
manually for any nonemergency reason, the switch input signals can
be adjusted to be of equal phase and level; this combined with the
‘“‘make before break” feature minimizes perturbations of the JFs out-
put signal caused by nonemergency switch operation.

Manual control is provided in the srs alarm-and-switch-control
panel by means of pushbuttons. Accidental manual switching is pre-
vented by requiring multiple simultaneous pushbutton operations for
switching. It is possible to override automatic-switch-control signals
manually.

The alarms received from the four level detectors (Fig. 9) and the
JFq’s are processed by the switch-control circuit, which sends a com-

mand to the switches if an automatic switch is warranted. The switch
will operate if detector A or D indicates an unusual change in signal
level or if the regular sre indicates it is impaired sufficiently to require
being switched out of service. Should detector B or C indicate an un-
usual change in signal level or the standby sre indicate it is impaired,
switch operation will be inhibited.

If a sre failure occurs, the sFs may have to operate for some time
with only two sra@’s. Patching facilities are provided for restructuring
the sra interconnections (Fig. 10). For example, if the master sre
fails, the standby sra temporarily serves the master sFa@ function. The
alarm-and-switch-control circuit must be modified to recognize the new
arrangement. This modification is achieved by selecting a position on
a rotary switch corresponding to the new arrangement, and then
operating an associated pushbutton switch. Because the srs can func-
tion for some time with only two sFe’s, the Jrc is spared on a regional
rather than a per-office basis. For most of the other units in the srs,
spares are included either in the srs itself or in each office.

A jack-and-patch-plug arrangement has been provided in the alarm-
and-switch-control panel so that the srs output signals can be routed
around the coaxial switches. Using this arrangement to take the
switches out of service produces small phase and amplitude ‘‘hits” on

Fig. 10—Jumbogroup frequency supply operation using two jumbogroup fre-
quency generators.

the JFs output signals. When the patching operation has been com-
pleted, the signal through the patched path is within 2 degrees and
0.2 dB of what it had been through the switch path.

A rotary-switch-and-power-meter arrangement has been provided
on the alarm-and-switch-control panel to enable the craftsperson to
read the level of each frequency (1.024, 2.56, and 20.48 MHz) at
detectors A and B shown in Fig. 9.

The JFrs incorporates a test unit to measure small frequency offsets,
since a typical office does not have precision-frequency-measuring
equipment.

The test unit can compare two frequencies of approximately 20.48
MHz, and display on numeric indicators the fractional frequency offset
in parts in 10" or 10". The test unit has two modes of operation, either
of which can be selected by a switch on the front of the panel. The
6 sEc, or normal, mode measures frequency differences of parts in
10°. The 30 sEc, or extended precision, mode measures frequency
differences of parts in 10". The frequency-offset information is dis-
played on LED numeric readouts.

Signals to be compared must have a level greater than —30 dBm
into 75 ohms, and the frequency difference must be less than one part
in 10°. If either of these conditions is not met, the OUT OF RANGE
indicator will light, and the test unit will not operate. The counTING
indicator is lit while the test unit is counting. When the couNTING
indicator goes off, information is transferred from the counter to the
numeric readouts. After approximately six seconds, a new count is
taken.

The internal functions of the test unit can be divided into three
sections: gate generator, counter, and display. The gate generator
provides a gate whose period is a function of AF, the frequency differ-
ence between the two input signals. In the counter, the duration of the
gate is measured by counting proportional to the fractional frequency
offset (AF/F). This is similar to the frequency-offset-detection tech-
nique used in the src. The information in the detection counters is
transferred to a digital display unit.

The first sFs production unit was shipped from the factory in June
1973. There are presently more than ten srs bays installed at L5 main

stations. The operational performance of these units has been excellent,
and they have met all of their design objectives.

The availability of these accurate and reliable frequency supplies
has stimulated the modernization of the entire Bell System synchroni-
zation network.‘ Beginning in 1974, the United States will be divided
into approximately twenty synchronization regions, and telephone
offices in each region will receive their synchronization signals from a
regional frequency supply (RFs). Certain L5 main stations will become
regional syne centers, with the L5 srs serving as the rFs and providing
64-kHz and 512-kHz sync signals to telephone offices in its region. In
areas where no Ld facility exists, a special two-sFre version of the Jrs
will serve as the rFs. Thus, the entire frequency-division multiplex
plant, including both radio and coaxial transmission systems, will soon
be synchronized through srs equipment.

In addition to serving L5 and the Bell System synchronization net-
work, the srs has sufficient frequency precision to serve future systems
having ten times the bandwidth of L5.

The authors acknowledge the many contributions of R. E. Benjamin
to the srs development; he has been involved with the srs since its
inception. Several others contributed to the development of the sFs,
in particular, W. W. Brown, L. J. Finelli, D. A. Lane, W. G. Albert,
D. M. Pierce, and W. C. Schmidt. Special mention is due W. L. Smith
and H. 8. Pustarfi whose expertise in the field of precision frequency
sources led to the development of the 39A oscillator.

2. J. F. Oberst, ‘“Keeping Bell System Frequencies on the Beam,” Bell Laboratories
Record, 52, No. 3 (March 1974), pp. 84-89.

3. A. F. Flint and H. S. Pustarfi, “L5 System: 39A Precision Oscillator,’’ B.S.T.J.,
this issue, pp. 2097-2107.

4. R. E. Powers, “Reference Frequency Transmission Over Bell System Radio and
Coaxial Facilities,’ Proc. of 28th Annual Frequency Control Symposium,
Fort Monmouth, N.J., U.S. Army Electronics Laboratories, May 29-31, 1974.

Copyright © 1974 American Telephone and Telegraph Company
Tue BELL SystEM TECHNICAL JOURNAL
Vol. 58, No. 10, December 1974
Printed in U.S.A.

Each L5 coaxial line is capable of carrying three 8600-channel basic
jumbogroup signals that are translated to and from the Lé line spectrum
(3.124 to 60.556 MHz) through jumbogroup multiplex equipment. The
LS line signal also includes several line pilots and switching, maintenance,
and reference signals. Administration of all these signals ts performed by
the L5 line connecting circuits, which vary in complexity with the type of
matin station they serve.

In addition to the circuits required to handle the various components
of the Ld line spectrum, signal administration is also required at basic
gumbogroup frequencies, before the jumbogroup multiplexing step, to
allow interconnection to lower-order multiplex or other systems using the
basic jumbogroup frequency format of 0.564 to 17.548 MHz. To perform
this function, the basic jumbogroup trunk bay was developed and provides
interconnection flexibility hitherto unavailable for direct connections to
other long-haul systems, such as L4, LMp, Tp, or TH radio, or LS systems
of another route.

The L5 coaxial line is a transmission facility with a message capacity
of three jumbogroups. These jumbogroups are placed on the line in the
frequency format shown in Fig. 1. Each jumbogroup begins as a
basic jumbogroup signal, formed by the mastergroup multiplex-2
(mMx-2) frequency-division multiplex or the basic jumbogroup trunk
equipment. This basic jumbogroup consists of six 600-channel basic
mastergroups and has a frequency assignment identical to that of the
L4 line assignment.! Each of three 3600-channel basic jumbogroups
is translated to the L5 line spectrum through the jumbogroup multiplex

(gmx) frequency-division multiplex equipment? to form the 10,800-
channel line signal.

several pilots used for the dynamic equalization and temperature
regulation of the line,’ switching signals required for the control of the
line-protection switching system-3 (LPss-3),4 fault-locating signals
used for line maintenance tests in the transmission surveillance system
(rss),5 and a reference frequency signal used by the jumbogroup
frequency supply (srs).° Several rules and administrative functions
must be applied to the message and other signals forming the L5 line
spectrum which vary under specific situations. The line-connecting
circuits described in Section II perform all of these functions.

In the evolution of the various long-haul systems such as the L-
carrier and TD and TH radio systems, the need for more direct inter-
connection between these systems in large channel blocks became
increasingly evident. Prior to the development of L5, the predominance
of intersystem message interconnection was done on a basic master-
group basis,* which requires the use of costly mmx terminals and
mastergroup connectors. The need for a simpler and less expensive
means of intersystem interconnection was recognized early in the L5
development, especially in view of the large 180-mastergroup capacity
of this new system.

This need was met with a new bay, designated the basic Jumbogroup
trunk bay (BseT), which allows interconnection of single- or multi-
mastergroup signals in the basic jumbogroup spectrum; i.e., before the
jumbogroup multiplexing step to the L5 spectrum through sux
equipment. Interconnections may be made directly to radio systems
(using 3A wire line entrance links), L4 systems, L-carrier mastergroup
digital (LMp) terminals, mmx-2 terminals, or other L5 systems. The
BJGT circuits are covered in detail in Section ITI.

As mentioned in the introduction, the function of the line-connecting
circuit is to process the Ld line signal in accordance with circuit require-
ments and certain administrative rules. Although many options are
required to handle the various conditions that arise, there are only
three basic line-connecting arrangements, one for power-feed main
(PFM) stations, another for switching power-feed main (sPrM) stations
and, finally, one to cover terminal stations or terminal main (TM)
stations.!

All line-connecting equipment is located in the line transmit-receive
bay. With the exception of line-connecting equipment, transmit-

*The basic mastergroup is the “U600” output of the L-type multiplex terminal
(Ref. 7).
t The distinction among the four types of main stations is covered in Ref. 8.

—_———_—_———_—
TO LINE
RCVG LINE CONN | TRMTG LINE CONN i eaonn
DETR FROM
(MAJOR ALM) FLO
TO
TO LPSS-3 PILOT TO
TSC DETR RESUP TSC
FROM SW
TRMTG PIL | |
FROM TST GEN TRMTG
RCVG LINE TST
EQUALIZER
/ mco
“o'une + o
TO LINE
JFS PILGEN | INTERCONN
[ DETR LH +] i
(IDLE ALM) TO
TRMTG
EQUALIZER
EQL SIG
! ‘ J
FROM | FROM
To | stBy | RSTN LH | a FROM
LPSS-3 | LINE TRMTG §fsc
DETR SW
TO TO LINE PILOT
RSTN INTERCONN ae TO - GENERATOR
DETR moar eet le BT Ie \\rrom STBY LINE ec CKT
(INPUT ALM) TO JMX TO ANOTHER FROMANOTHER ‘MX Poe gs fo
TERMINAL L5 ROUTE L5 ROUTE TERMINAL a aL Baa
PS ee a See ae SS ee ]
| LINE INTERCONN DETR CKT Tine INTERCONNECTING CKT (THRU OR BRANCH)! PILOTS | | ule
From (INPUT ALM To AEF FREO | Reve
DETR LINE ] SIGNAL
BRIDGE INTERCONN | FROM
POINTS LPSS-3
SWITCHES | Sanat
ee | ie Sins SF Sees Ie ee De reel ]

receive bays are virtually identical for all types of stations.*® The
line-connecting circuits are simplest at Prm stations where no signal
processing is performed, and all message continues through the office.
A pad and hybrid, for fault-location oscillator (FLO) access to the line,
are the only apparatus in the line-connecting circuit. At sprm stations,
all message is again connected through; however, switching apparatus,
filtering, and additional access to the transmission surveillance
center (Tsc) are added to the line-connecting circuits for system
administration.

Terminal or Tm stations require similar basic equipment units in
the line-connecting circuits, but the terminal station options within
that equipment are simpler. Each Tm station transmit-receive bay
requires six basic circuits to perform the line-connecting functions: the
receiving line-connecting circuit, transmitting line-connecting circuit,
pilot generator circuit, line-interconnecting circuits, line-interconnect-
ing detector circuit, and, optionally, the line-branching circuit. These
circuits are shown in the block diagram found in Fig. 2, and are
discussed more thoroughly below.

The signal administration role of the line-connecting circuits in T
stations may be reduced to six major functions:

(t) Message Administration: Provides routing of the message
signal for any combination of the three jumbogroups.

(it) Pilot Administration: Introduces equalizing and temperature
pilots to the line and provides pilot blocking when required.

(i172) Switching Administration: Provides line switches, introduces
switching signals to the line, and provides detection access and
signal blocking when required.

(tv) Transmission Surveillance Administration: Provides access to
the line at strategic points for Tsc analysis, and furnishes signal
blocking when required.

(v) Reference Frequency Signal Administration: Introduces a
reference frequency signal to the line, and provides signal
blocking when required and a distribution network for circuits
requiring the use of a reference frequency.

(vt) Restoration Access: Provides access to the regular transmitting
and receiving lines for restoring L5 over other facilities and
access to the standby transmitting and receiving lines for
restoring other facilities over L5.

The simplest of the Tm station line-connecting circuits is required
when all message passes through the office; i.e., no jumbogroups are
branched to other L5 lines or are dropped to mx equipment. In such a
case, the line signal entering the line-connecting circuit from the
receiving equalizer (Fig. 2) would only connect (via splitting hybrids)
to the A and B Turv cxtT modules at the output of the receiving line-
connecting circuit. No message blocking filters are required in the
THRU CKT modules for this application.*

In the example shown in Fig. 2, the message administration is
arranged to split the jumbogroup signals into three directions: one
jumbogroup connected through, one branched, and one dropped to
jmx. Assuming JGl is connected through, s@2 branched, and Je@3
dropped, blocking filters would be assigned to block sa2 and Ja3 in
the THRU CKT and Jel and Ja3 in the BRANCH cKT. No blocking filters
are required in the drop circuit, since the smx circuitry selects that
jumbogroup for which it is equipped.

Any combination of jumbogroups can be passed in the through,
branched, or dropped path. The blocking is accomplished through the
use of five arrangements of filter designs developed for L5 use. The
requirements for jumbogroup blocking are outlined in Fig. 3.* In all but
jumbogroup 2 blocking, single high-pass or low-pass filter designs are
used ; however, in the case of jumbogroup 2, filters d and e are paralleled
with a “‘split-apart’’ filter at the input and output of the paralleled
pair. The advantage of split-apart filters, as compared to hybrids,
allows better return loss with lower in-band loss.

A minimum of 80-dB out-of-band discrimination for each blocking
arrangement is required. Return loss is approximately 26 dB (75 ohms).
Maximum insertion loss is less than 4.0 dB.

Ideally, it is desirable to maintain continuous-line pilot continuity
throughout a frogging® section (approximately 800 miles). Two factors,
however, preclude this possibility. When jumbogroup blocking filters
are required at TM stations, one or more line pilots are attenuated to
some degree because they fall within the attenuation region of the
filters. In addition, the stability of each line pilot is adversely affected
as the number increases of regulating repeaters or dynamic equalizers
through which they must pass. Therefore, the pilots are blocked and
reinserted at specific intervals.’

Since the administration of individual pilot blocking and reinsertion
would be unwieldy, all pilots are blocked when any pilot or jumbogroup
signal is blocked. Thus, rules for pilot blocking were established such

* A more detailed description of filter design techniques for L5 is given in Ref. 10.

that all pilots are blocked when either (7) jumbogroup blocking is
provided or (77) pilots will otherwise pass through more than four E3

equalizers.
Minimum requirements for the pilot band-elimination filters are as

The line-connecting circuits provide several functions for the
administration of the Lpss-3. A bridged access point is provided at two
locations, (z) on the line side of the receiving switch and (27) on the
line side of the transmitting-line switch hybrid (Fig. 2). The switch-
initiator circuit monitors the 42.880-MHz pilot at these points for
LPss-3 operations.

Unlike L3 and L4, the L5 line switches are located within the line-
connecting equipment to reduce the cable length in the through signal
path. This was made possible through the use of parallel switch
cabling, as opposed to the series arrangement in L3 and L4 systems.‘

Two switches are provided in the receiving line-connecting circuits,
designated the RcvG sw and the TERM sw. The Rcv6G sw furnishes access
from the standby line when the associated regular line is out of service
for any reason. This switch also provides a termination to the out-of-
service line. The TERM sw is either automatically operated as a result
of a line overload condition, thereby preventing noise propagation to
other systems, or manually operated under certain abnormal line
conditions. The transmitting line-connecting circuit also includes
two switches, the TRMTG sw and the mco (message cutoff) sw. The
TRMTG sw provides access to the standby line when the associated
regular line is out of service. The mco sw, also controlled by the
LPss-3, provides a means for opening the transmitting end of a regular
line, primarily as an access for line equalization.

As indicated in Fig. 1, the line-switching signals occupy the band-
width between 68.76 and 68.78 MHz. These signals are blocked in the

line-interconnecting circuit at every switching station to prevent
interference with switching functions between switching sections.

Each transmit-receive bay at Tm and sPFrM stations has six measure-
ment points which can be accessed by the tTsc for automated remote
measurements of the Ld line status.° In addition to the measurement
points, an input access is also provided in the transmitting line-
connecting circuit so that out-of-service frequency characteristic
measurements may be remotely performed by the Tsc on a switching
section. This point, equalizing signal in (EQL sic IN), is also used for
line equalization described in an earlier section.

Another transmission surveillance system administrative function
performed by the line-connecting circuits is that of fault-locating
signal blocking. Fault-locating signals at 1.59, 1.60, 68.6, and 68.65
MHz must be blocked at each switching section to avoid interference
with fault-location routines in other switching sections. A high-pass
filter blocks the low-frequency fault-locating tones a minimum of 50
dB. An alternate high-pass filter design will also block the 2.976-MHz
equalizing pilot (Section 2.2) when required.

Since switching (Section 2.3) and fault-locating signals are always
blocked at each switching station, a single filter accomplishes both
functions. Because of the relative closeness of the 66.048 equalizing
pilot—which is not always blocked concurrently with the switching
and fault-locating signals—a low-pass filter was not a practicable
design. Instead, a band-elimination filter with a minimum of 50 dB of
suppression was developed for this purpose. In those cases where the
66.048-MHz pilot is also blocked (Section 2.2), a low-pass filter is
furnished to block all signals above the message band.

The smx requires a very stable frequency for the synchronization
of its carrier supply circuits. This frequency, 20.480 MHz, is generated
by the Bell System Reference Frequency Standard (BsrFs) located at

“Figure 4 represents a simplified block diagram of the line-connecting circuits for
a SPFM station. The circuitry is essentially similar to the Tm station, with the exception
of the line-interconnecting circuits, which are much less complicated.

a
RCVG LINE CONN | TRMTG LINE CONN
REGULAR sae
aptie ov | ae EQUALIZER
SW SW FLO & an
ng sta o Clear
SIGNAL
EQUALIZER Se
EQL
SIG IN TRMTG
LINE
TST
To FROM
LPSS.3 TSC
- To
PIL RESUP TSC
Less FROM SW
= | : re}
LINE CONN FROM GEN i a
DETR PIL GEN -
an TSC
OTHER
LINES Paha TRMTG
LINE
STANDBY LINES ah
1 10 o ft
FROM 42.880
RCVG BEF
EQUALIZER RCVG CVG TRMTG TRMTG
DIR SW sw ODI
SW <c | SW t TO
Sei TRMTG
| EQUALIZER
TO A RCVG TRMTG sae
LPSS-3 ! TST t TST PIL rH | a
DETR GEN 0
TO TO (TEMP PIL)

TSC TSC
Fig. 4-Simplified block diagram of spr station line-connecting circuits.

Hillsboro, Mo.® The ssrrs will supply the reference frequency signal
for all L5 systems, either directly, as in the case of the initial L5
Lillyville-Hillsboro route, or indirectly via other long-haul systems
in subsequent routes.

Since the reference frequency signal is only required on the first
regular working line and in only one direction, the line-connecting
circuit administration of that signal is limited to that line. As in the
case of the equalizing pilots, it is desirable to maintain the integrity
of the reference signal throughout the route without restriction;
however, if jumbogroup 1 or 2 is blocked in the through route, some
attenuation of the signal occurs that requires that a 20.480-MHz band
elimination filter be added to the receiving circuit of the through route
to block the signal sufficiently to prevent its interfering with the
reference signal reinserted in the transmitting line-connecting circuits
of that route. In such cases, the 20.480 signal is bypassed around the
through-line-connecting circuit, as described in the next paragraph.
The characteristics of the filter are similar to those of the 20.992-MHz
band-elimination filter described in Section 2.2.

In those offices requiring a reference signal for mx terminals (via
the srs circuitry), for reinsertion in a through route or for insertion
on an Ld branch route, a synchronizing receiving panel is provided.
This panel, which accepts the entire line signal from a port in the
receiving line-connecting splitting-hybrid circuit (Fig. 2), is furnished
with a 20.480-MHz bandpass filter, amplifiers, an adjustable attenu-
ator, and a hybrid tree that provides eight outputs for distribution as
required. The bandpass filter provides a minimum of 80-dB discrimi-
nation approximately +75 kHz from the reference frequency signal.

The line-connecting circuits provide access for message service
restoration both for L5 restoration and for the restoration of other
long-haul systems over standby Ld lines.

Access to restore a failed L5 facility is provided at each regular
line at the TERM sw (Fig. 2) in the receiving line-connecting circuit
and the TRMTG sw in the transmitting line-connecting circuit. These
access points connect—via the restoration patch bay—to the standby
L5 line of another facility or, through idle smx terminals, to spare
equipment of other long-haul facilities. Another access point is provided
at the TERM sw to allow monitoring of the failed lines at the restoration
patch bay. Similar points are provided with the standby line connecting

equipment to allow direct restoration access to other failed L5 facilities
or other types of long-haul systems through idle ymx equipment.

During the early development of the L5 line-connecting circuits, it
was recognized that considerable gain would be required at Tm stations
to offset the loss of the filters required for specific administrative
functions. Since as many as 18 mastergroups may be served on each
line-connecting circuit, redundancy was provided to assure reliability
consistent with the overall L5 system. Although without redundancy
the reliability of the amplifiers in isolation was considered adequate,
the associated power, fusing, and related cabling associated with the
active elements made redundancy necessary.

A beneficial byproduct of the redundancy in line-connecting circuits
is the ability to provide all optional equipment in a modular package,
thereby allowing in many cases in-service rearrangements of circuit
assignments with less danger of service outages.

The line-interconnecting circuit modules are depicted in Fig. 2 as
the A and B Turvu ckt or A and B BRANCH cKT. The A and B paths
terminate in a coaxial switch that provides the automatic protection
for the A or B unit, whichever is serving the line. Jumbogroup pilots
control the switch logic circuitry, which can be rather complex because
of the wide number of options available in the line-connecting circuits.
Three points of detection are required to satisfactorily implement the
switching logic: (2) beyond the output of the switch to indicate a
major alarm’ at loss of pilot; (¢7) at the output of the alternate leg of
the switch to monitor the condition of the idle line-connecting circuit ;
and (27) at the input of the modules to assure that the jumbogroup
pilots are present at the proper level. Since as many as three sets of
line-interconnecting modules may be required, i.e., one through circuit
and two branch circuits, up to nine detector circuits are required as a
maximum for monitoring the three jumbogroup pilots at the three
points of detection. The simplest case, of course, is when all jumbo-
groups are “through,” in which case only one jumbogroup pilot need
be detected (at the three points) for the full message load.

Line-interconnecting circuits for the standby line are considerably
simpler than the regular lines, since no jumbogroup blocking or syn-
chronization administration is required. All intermediate stations are
therefore ‘‘through routes” on the spare line for an entire frogging®

* All line-connecting alarms and status indications may be connected to remote
alarm systems.

section. Although active circuitry is furnished with the standby
line-connecting circuits, no automatic protection is required, inasmuch
as service does not normally traverse an office on the standby line. An
alarm is provided, however, to monitor continuity through the standby
line-interconnecting module, and a spare module furnished may be
manually patched in to replace it.

As shown in Fig. 2, the pilot generator circuit combines all the pilots
and maintenance signals so that only one input port is required for
their application to the transmitting line-connecting circuit. Several
options within the generator, as noted in Fig. 5, are consistent with the
various options required for line-connecting circuit administration.

The pilot-generator circuit provides the means for combining the
switch signals from the Lpss-3 system, the reference signal from either
the local standard or the receiving reference circuit (Section 2.5), and
the four locally generated equalizing pilots. The four pilots are derived
from free-running crystal oscillators mounted within the generator

panel, which are designed to maintain a frequency accuracy of approxi-
mately one part in 10° per year under normal office temperature
variations. Harmonic distortion is required to be at least —40 dB
relative to the fundamental.

In all sprm and T» stations,* the minimum options required are
the 42.880-MHz oscillator and the switch signal combining circuits.
The 42.880-MHz signal is required for regular lines to connect to the
pilot resupply switch (Fig. 2), which automatically reinserts the pilot
should (7) the pilot be lost in the through path, or (7) the mco switch
be operated for equalization or Tsc measurements, thereby sustaining
temperature regulation of the line during out-of-service conditions.
Since the 42.880-MHz pilot for standby lines is always blocked and
reinserted (Fig. 4), the oscillator is always required with standby
transmit-receive bay pilot generator panels.

Another feature provided for standby lines is a switch for reinserting
three of the equalizing pilots. This switch is provided at sprm and T™
stations where pilots are not blocked in the through path. Since there
may be times when the standby line is being used for protection for
sustained periods, the stations with through pilots would otherwise
be without dynamic equalization; thus, when a receiving line switch
occurs, three equalizing pilots are automatically reinserted for the
duration of the switch.

As discussed in the introduction, the basic jumbogroup trunk bay
design is unique in long-haul system development in that it provides
a direct standard interconnection between the ymx and certain multi-
plex terminals and systems of multi-mastergroup capacity. Any
combination of contiguous mastergroups in the basic jumbogroup
format may be interconnected from L4 lines, MMx-2 or LMD terminals,
radio systems using 3A wire line entrance links, or other mx terminals
as indicated in Fig. 6a.

Actually, a maximum of four inputs per jumbogroup is available
to the Biat bay, so that a more practical arrangement with contiguous
mastergroup assignments is shown in Fig. 6b. Through judicious
planning of the mastergroup assignments, line engineering personnel
can more economically engineer facilities; however, there will still be
cases where some interconnecting systems will carry identically

*prMm stations are not furnished with pilot generator panels; however, a special
portable, equalizing-pilot generator unit is available for installation and maintenance
tests.

Fig. 6—(a) Basic jumbogroup trunk function. (b) Example of basic jumbogroup
trunk arrangement with four-system input.

numbered mastergroups, thus requiring multiplex equipment to
reassign their frequency allocations.

The ssat bay is required whenever a Mx terminal is furnished, since
it provides not only the transmitting and receiving trunk circuits for
interconnecting the gmx to the various terminal arrangements, but
also the 5.888-MHz jumbogroup pilot. The Byet bay provides more
than 250 different list structures to accommodate all combinations of
contiguous mastergroups to associated systems.

The transmitting and receiving trunk circuits consist of signal-
processing panels and switch and logic panels. An example of the
transmitting trunk circuit (transmitting to smx), shown with a two-
input circuit arrangement, is given in Fig. 7a. The signal from mmx,
radio, tMD, L4, or another jmx terminal is split into two paths to
identical, redundant signal processing circuits consisting of filters,*
amplifiers, cable equalizers, and, in the case of L4, de-emphasis. Each
processing circuit connects to a coaxial switch controlled by the logic
circuitry in the switch and logic panel. The logic circuitry is activated
when the mastergroup pilot level associated with the highest-numbered
mastergroup in that particular circuit arrangement changes by a

*The filters are low-pass, high-pass combinations, similar to those used in L4
blocking and branching arrangements,! and are required for removing unwanted
signals in the various spectra of interconnecting systems.

ouTPuT 1 | TO
MMX,
FROM TO 3RD AND 4TH RADIO,
JMX RECEIVING OUTPUT LMD,
CIRCUITS AS REQUIRED OUTPUT 2 oe

predetermined amount. In addition, the logic circuitry provides local
and remote alarming.

The transmitting trunk circuit also contains the 5.888-MHz basic
jumbogroup pilot oscillator. The insertion of the jumbogroup pilot
at the Bsat bay allows the interbay trunk to the mx bay to be alarmed,
thereby offering further protection against personnel error or cabling
problems.

The receiving basic jumbogroup trunk circuits are very similar in
design to the transmitting circuits, as shown in Fig. 7b; however, in
addition, a 5.888-MHz band elimination filter is provided to prevent
interference with connecting systems, and pre-emphasis is furnished
when connecting to L4.

10. J. L. Garrison, A. Olsen, Jr., and T. H. Simmonds, Jr., “LS System: Trans-
eat Networks and Magnetic Components,” B.S.T.J., this issue, pp.
2203-2248.

Copyright © 1974 American Telephone and Telegraph Company
Tue Beut System TEecuHnicaL JOURNAL
Vol. 53, No. 10, December 1974
Printed in U.S.A.

The L& Coaxial-Carrier Transmission System equipment was designed
to provide long-term reliable service in diverse environments. The line re-
peaters, for instance, are housed in apparatus cases located in manholes
that are subjected to flooding, whereas the main-station repeaters and
terminal multiplexing equipment are located in the controlled environment
of underground or aboveground buildings, often only partially attended by
maintenance personnel. In addition to describing novel physical designs,
this paper covers other very important considerations such as thermal
design, manufacturability, incorporation of hybrid integrated-circurt
technology, efficient ‘“‘building-block”’ system growth capability, and long-
term reliability.

Proposed approaches to the physical design of L5 repeaters covered
a wide array of possibilities ranging from the buried repeater concept
to that of an ideal conversion. The former would capitalize on the high
reliability and trouble-free performance of solid-state repeaters to gain
cost advantages in outside plant for new routes, but would require
initially equipping all tubes of a coaxial cable with repeaters. The
latter approach would feature repeaters configured to closely resemble
existing L4 system repeaters.! It would encourage conversions from
the L4 to the L5 system and take advantage of as much existing out-
side plant as possible. The range of proposals also included such alter-
natives as limited-access enclosures, underground lockers, and single-
repeater apparatus cases, all of which would have required extensive
outside-plant apparatus development.
2147

Experience with the L4 system showed that accessibility to line
equipment is essential during initial installation of a system. This
precluded burying L5. Perhaps, after a shakedown period, buried
repeaters can be developed for new-route applications. One-mile
repeater spacing and continuing improvements in reliability of solid-
state devices may create a wish to bury L5 in the future. In this case,
the basic repeater, the least complicated and most numerous of the
L5 repeaters, is the most likely first candidate.

Clearly, the pursuit of a conversion design was the most logical
initial approach to the physical design of the L5 repeaters. There will
certainly be a continuing need to convert L4 routes to L5 when the
demand for circuits exceeds L4 route capacity. Also, the L5 trial in
mid-1970 was only realized by the use of existing plant designs where
possible. In addition, by pursuing a conversion design, the development
of the L5 system did not depend first on the acquisition of knowledge
of a wholly new repeater environment. The existing L4 environment
was usable for evaluation of the L5 repeaters during the development
period. Conversely, it was also understood that pursuing a conversion
design would not preclude the use of newly developed L5 outside plant
in applications at the 1-mile repeater points where new manholes are
required, and along new routes, if the need for new plant could be
justified. Thus, the established schedules for development, trial, and
commercial service for the L5 system, together with the economic ad-
vantages of reusing existing outside plant, all favored the conversion-
design approach to L5 physical design.

An early plan was directed toward ideal conversion—the situation
in which the L5 repeater shape would correspond closely to that of the
L4 repeater, allowing easy replacement in the existing L4 apparatus
case housing and chassis assembly. However, the proposed L5 system
fault-location scheme requires 14 copper-wire pairs in the coaxial
cable, whereas the L4 fault-location scheme requires only 5 wire pairs.
This became the controlling factor in causing a change in the L4 ap-
paratus case chassis. Once it was recognized that a chassis change was
needed, the degree of change was academic, and a modified conversion
plan was adopted. This plan was directed toward developing a repeater
configuration that would still fit the L4 apparatus case, but required
that the chassis of the apparatus case be changed to accommodate
administration of the L5 fault-location pairs, to promote good heat
transfer, and to simplify repeater installation. Since the Ld basic
repeater dissipates some 18 watts (compared to 13.5 watts for its L4

counterpart), it was necessary to design a more efficient thermal
circuit in the L5 repeater.

Figure 1 shows the L5 apparatus case chassis, the outside-plant item
required to convert an L4 apparatus case to L5. This is simply a base
plate holding guide pins that line up plug-in repeaters. The guide pins
serve also as retaining elements for the repeaters. Thus, the repeaters
are bolted on the plate by means of the retainer guide pins, eliminating
the need for the cantilevered guide and clamp arrangements used in

L4 (sometimes a source of problems because of the degree of precision
required to insure proper mating of the plug-in units). The repeater
is designed so that, when bolted in place, its leading surface is in
intimate contact with the apparatus case chassis. The repeater-ap-
paratus case chassis interface becomes an important link in the thermal
path from the repeater to the manhole.

Figure 2 illustrates the L5 basic repeater apparatus case arrange-
ment featuring four basic repeaters located on the periphery of the
chassis. The center space is occupied by the non-heat-producing fault-
location oscillator unit. Coaxial patch cords are used for insertion of
fault-location oscillator tones at both the input and output of the
repeaters via twin coaxial jacks located on the apparatus case chassis.
The twin jacks serve also as repeater by-pass points in the high-
voltage, series-powered L5 line to permit a power patch around a
repeater, without turning power down on a line, in the event repeater
replacement becomes necessary.

Apparatus cases are mounted in precast concrete manholes (6 by 12
by 64 feet high inside the basic repeater manhole) that are not intended
to be impervious to water. The apparatus cases are watertight and are
filled with dry nitrogen to 9-psi cable pressure to protect against water
seepage. Six apparatus cases are required in a basic repeater manhole
to fully equip a 22-tube coaxial cable. Adaptors, or ‘dummy’ ap-
paratus cases (one of which is shown at the top of Fig. 2), are installed
to permit an entire manhole to be racked and spliced initially. Ap-
paratus cases are installed only to cover the number of tubes being
equipped in order to defer expenditures until necessary.

The cross-connect apparatus case, an additional watertight ap-
paratus case that is somewhat different in design, is also located in the
manhole. It houses both the logic unit (used to turn on fault-location
oscillators) and the appearances of all the coaxial cable’s copper wire
pairs used for support systems such as fault location, order-wire, alarm,
and surveillance. A logic-unit support-shelf kit was designed to permit
conversion of this type of L4 apparatus case for use on Ld.

Figure 3 is an L5 basic repeater with covers removed. Elements of
the basic repeater are the fundamental building blocks of the L5
system and appear in every repeater in the L5 equipment.? These basic
elements, the preamplifier, the power amplifier, the low-frequency
networks, and the line-build-out (LBO) network, are assembled to an
inner chassis located inside a rugged die-cast aluminum outer housing.
Figure 4 is a simplified repeater cross section showing the relative
physical positioning of the apparatus on the inner chassis which is
assembled to a flange protruding from the inner surface of the outer
housing. The low-frequency networks protrude into the power separa-
tion filter (psr) cavity through a cut-out in the flange.

Because of the series-powering arrangement for the L5 repeatered
line, repeater circuit ground can be as much as +1150 volts de (1750
volts de under trouble conditions) from earth or sheath ground. In
Fig. 4, the cross section of the inner chassis assembly which is at circuit-
ground potential is shown clear to contrast with the heavy black cross
section of the outer housing, which is at earth-ground potential.
Insulation between these two items is provided by a conformal coating
of epoxy, 0.015-inch thick, applied to the inside cavities of the outer
housing. A thin layer of epoxy sheet adhesive is used to bond the inner
to the outer. The controlled-thickness epoxy interface provides the
structural strength required, preserves the integrity of the insulation
for the safety of personnel and the protection of precision, low-voltage
electronic components, and constitutes a very low impedance to the
thermal path.

The insulating and bonding techniques are the same as those de-
veloped for the L4 repeaters. As in L4, the bond joint was designed to
cover a relatively large area for strength and for enhancement of
thermal conduction, which is further enhanced by the massive cross
sections of the repeater’s structural elements. The epoxy is applied by
the fluidized bed process. The process is carefully controlled to eliminate
voids, thereby minimizing the likelihood of high-voltage corona

The individual apparatus units are contained in die-cast aluminum
cans fastened to the inner chassis by threaded inserts integral to the
inner chassis. The repeater covers are coated with epoxy on the inside
to complete the continuity of the insulation of the two cavities. They
are secured using one-way screws as a safety feature. In effect, ‘‘sealed’’
repeaters are shipped to the field. There are no field adjustments to
be made on a basic repeater, and repairs, if necessary, are made at the
factory. Therefore, there is never a reason to open a repeater in the
field. They are sealed to avoid the exposure to high voltage that would
occur if a coverless repeater were accidentally inserted into a powered
apparatus case. Thus, safety features are not compromised by the use
of the simplified apparatus case chassis design (Fig. 1). High voltage
appears on the center conductor of the coaxial jacks that are located
deep inside the apparatus case, and sealed repeaters keep the high
voltage from the reach of craftspeople.

The preamplifier and power amplifier are the heart of the L5
repeaters. They are thin-film hybrid integrated circuit amplifiers, and
in the basic repeater they are located side by side in the center part of

the repeater as shown in Fig. 3. Even though the amplifiers require
many discrete components such as transformers, inductors, capacitors,
and transistors, it was established very early in the L5 development
that conventional printed-wiring arrangements could not be used.
The precision required for L5 amplifier circuits was such that thin-film
circuits were essential to the amplifiers. However, L5 required large
substrates, holes in the substrate to mount discrete components, low-
value precision resistors, and high-conductivity conductor paths. Each
of these requirements suggested a departure from generally accepted
thin-film practice. Together, they constitute a unique set of require-
ments and resulted in uic’s that are unique in their physical and
electrical attributes.

Although most uic’s are small enough to fit many patterns on a
single 32 by 43-inch ceramic mirror, the L5 uic’s are large enough
(23 by 33 inches) to allow only one per mirror. Conventional, leaded
components are almost never mounted on a ceramic substrate, but
this was required here. This gave rise to the need for about 50 holes
per substrate. Holes complicate the manufacture of. substrates and are
avoided, if possible. To produce the low-value precision resistors, the
resistive metal layer was deposited to a thickness of 25 ohms per square,
as opposed to the conventional thickness of 50 ohms per square. This
allowed longer resistors and a larger proportion of anodized-to-un-
anodized area. The higher percent of the area anodized results in
greater long-term stability. Furthermore, thicker metal film is more
stable because resistance changes are caused by surface changes, and
the surface constitutes a smaller portion of a thicker metal film than
of a thinner metal film. An operating temperature of less than 90°C
is necessary to maintain end-of-life tolerance for the precision resistors.
Power resistors, on the other hand, would stay within their required
resistance tolerance at film temperatures as high as 110°C. The power
resistors and the discrete transistors are virtually the only heat pro-
ducers on the substrate. Therefore, the layouts were designed so that
the power resistors skirted the periphery of the substrate. The transis-
tors, however, had to be centrally located because of circuit require-
ments for short connecting conductor lengths. Massive heat sinks were
designed to divert the transistor heat from the precision resistors and
other temperature-sensitive components.

Figure 5 is the L5 power amplifier. The top view shows the two heat
sinks, each housing a matched pair of transistors, surrounded by other
discrete components. The bottom view, the thin-film side of the hybrid
integrated-circuit amplifier, illustrates the film topology and a number

of reflow-soldered barium titanate chip capacitors. A special conductor
metal system was developed for L5 to produce the needed high-con-
ductivity paths and to provide solderable pads for the chip capacitors
and the leaded conventional components. The high conductivity is
obtained by depositing ‘‘thick” gold. Electroplated rhodium is added
to prevent the gold from dissolving in the solder. A gold flash over the
rhodium protects the rhodium from oxidation, thereby retaining its
solderability. By selectively etching the gold to expose a rhodium strip
around the solder pads, solder ‘“‘dams’”’ are formed after the exposed
rhodium strip becomes oxidized and nonsolderable when exposed to
the elevated temperature of the film pre-aging process. Therefore,
during the solder reflow process, when the chip components are con-
nected into the circuit, the exposed rhodium strips confine the solder

to the area they surround. This avoids wetting and possibly degrading
the high conductivity paths. By precisely defining the solder area,
thereby limiting the amount of gold that can dissolve in the solder,
embrittlement of the solder joints is avoided. Complementary develop-
ments determined the correct amount of solder to be supplied as an
integral part of the chip component terminations and the precise
reflow soldering cycle, thereby assuring reproducible, reliable Hic
assemblies.

It was necessary to develop a structural support for the large, thin
ceramic substrates that would permit efficient thermal conduction
from the transistors, but that would keep the substrate free of me-
chanical stress. The substrates are not perfectly flat, and the amount
of bowing or warping is limited by die-screening subsequent to firing.
Ideally, the substrate would be fixed at three points (to define a plane)
and contact to the transistor would be by a flexible heat conductor.
However, the requirements are inconsistent for providing, simulta-
neously, a massive heat sink for adequate heat dissipation and a flexible
heat sink for stress-free mounting. Consequently, the structural support
developed was to hard-mount the substrate on the massive heat sink
and merely stabilize its outer periphery on three bosses of the die-cast
aluminum amplifier housing. The location of the three bosses and the
stabilizing screws is shown in Fig. 5, and a partial cross section of an
amplifier assembled in a repeater is shown in Fig. 6.

In Fig. 6 the large substrate, in cross section, appears virtually as a
beam with a fixed support in the center. The end supports (only one
shown) are designated ‘‘grommet suspension.” At these points, the
substrate is held between two soft, rubber grommets confined in a large
hole in the substrate by a shoulder screw anchored to the boss in the
amplifier housing. The role of the grommets is to stabilize the nonflat
substrate in whatever position it may happen to fall and have it
remain there, stress-free.

The substrate’s fixed support is the massive transistor heat sink,
also shown in cross section in Fig. 6. The discrete transistor, located
inside the heat sink, has a beryllium-oxide header hermetically sealed
to a Kovar* case. Heat sinking is to the beryllium oxide, which has the
higher coefficient of thermal conductivity, and thermal flow through
the transistor-heat-sink interface is enhanced by the force exerted by

the preloaded spring confined within the heat sink. While high, com-
pressive pressure is applied through the transistor to the heat sink,
none of it is transmitted to the delicate thin-film substrate to which
the transistor is soldered. The result is a stress-free structural support
for the large L5 substrates and an effective thermal circuit. As shown
by the arrows in Fig. 6, heat is conducted from the transistor through
the heat sink, through the amplifier cover into the inner repeater
chassis, and through the thin layer of epoxy to the outer repeater
housing.

Close spacing of the transistors on the substrate was required to
minimize parasitic inductance and capacitance in the electrical circuit
and resulted in a unique heat sink design. Spring-loading the transistor
within the confines of a massive heat sink was developed for use in the
L4 system amplifiers. However, the L4 scheme was not applicable to
L5 because of the combination of space limitations and the application
of the heat sinks to delicate, large, ceramic substrates in L5, in contrast
to the rugged epoxy-glass printed wiring boards used in L4. The L5
heat sink is a precision assembly that, for good heat transfer, presents
the largest possible metal cross-sectional area within the allowable
limits for the transistor lead lengths and transistor spacing. The screw
that preloads the spring against the transistor serves as the threaded
nut for mounting the heat sink against the flat amplifier cover. To
minimize thermal impedance, the heat sink’s functional faces have a

32 micro-inch finish, and an indium washer is inserted at the transistor-
heat-sink interface. This assures intimate contact at the thermal
circuit joints.

The heat sink’s cavity, large enough to accept the transistor with its
flanged case, is constricted at the threaded portion because of the
transistor spacing constraints. This constriction makes it necessary to
have a side opening in the heat sink to load the transistor and precludes
the use of conventional screw-machine methods of manufacture.
Double heat sink designs are used in the preamplifier and power ampli-
fier to maximize the amount of metal usable for heat sinking within the
transistor spacing constraints. This works out well, because the transis-
tors are used in matched pairs and can be preassembled into the heat
sinks for subsequent assembly onto the substrates.

The thin-film substrates are housed in die-cast aluminum cans.
These cans require a certain degree of precision to provide accurate
stress-free positioning of the substrates, as shown in Fig. 6. Substrate
supporting surfaces are held within tight tolerances with respect to
the cover supporting surface. The bottom amplifier cover is controlled
for flatness and is designed to be overflush to present maximum contact
area to the repeater inner chassis for heat transfer.

Having top and bottom amplifier covers removable provides ac-
cessibility to both sides of the substrate during shop processing and also
for trouble shooting, and the open die-cast frame of the amplifier
housing serves as a convenient carrier and substrate protector during
handling in the shop.

Connections to the substrate are made through 75-ohm miniature
coaxial connectors that clearly define electrical and mechanical
boundaries for the amplifiers. This substantially facilitates amplifier
testing, repeater assembling, and, if needed, repair and replacement.

To ensure reliable performance of silicon solid-state devices, the
physical design goal was to limit the maximum junction tempera-
ture to 125°C and, in the case of the elements of the L5 repeatered
line, can be expressed as

Tz = transistor junction temperature,
ATac = temperature rise in the repeater apparatus case,
ATr = temperature rise in the repeater,
AT, = temperature rise from case to junction of the transistor
because of its thermal impedance,
Ty = maximum manhole ambient temperature.

The maximum junction temperature for transistors in the L5 basic
repeater power amplifier has been determined by tests to be 128°C,
and comparative performance of L4 and Ld is shown in Fig. 7. Re-
calling that an L5 repeater dissipates some 18 watts in comparison to
133 watts for L4, it is evident that the thermal conductivity enhance-
ment was realized as anticipated because of the apparatus case chassis
design and the repeater packaging techniques developed for L5. At
128°C, maximum transistor junction temperature for L5 transistors,
thermal results of L4 and L5 are equivalent and, based on L4 per-
formance to date in the field, it appears that high-reliability per-
formance of L5 devices is ensured.

The ‘‘modified-conversion” design concept as applied to the L5
regulating repeater requires that two repeaters fit in one apparatus
case. The two repeaters are bolted to a newly designed apparatus case
chassis, similar to that for the basic repeater illustrated in Fig. 1, ina
layout that positions the fault-location oscillator unit between the two
repeaters. A maximum of 11 apparatus cases are required to accom-
modate an L5 22-tube coaxial cable, and these cases are arranged on
one wall of a precast concrete manhole 6 by 12 by 9 feet high on the
inside, 23 feet higher than a basic repeater manhole.

The regulating repeater package is one of the most difficult of the
L5 physical designs. A number of design constraints were imposed,
such as field access for adjustments, addition of constituent apparatus
and in-service testing, heat dissipation, high-voltage insulation, and
personnel safety.

This repeater contains the same circuitry as a basic repeater with
the addition of regulating circuitry, space for field insertion of two
line-build-out networks and a deviation equalizer, and a low-voltage
test point for in-service measurements. Like the basic repeater, a
rugged, epoxy-insulated outer aluminum die-cast housing holds the
captive retainer screws for bolting the repeater to the apparatus case
chassis. Unlike the basic repeater which has a flat-plate inner chassis,
the inner chassis of the regulating repeater is compartmented to accept
individually packaged circuit sections and units that do not re-
quire their own individual shield cans (see Fig. 8). Because of the
packaging density required to have two repeaters cantilevered from
the apparatus case chassis in a single apparatus case, the center flange
of the outer housing had to be kept to a minimum, allowing just
sufficient area to ensure a good bond to the inner chassis. Therefore,
compartmenting, in combination with individual apparatus housings
that act as stiffeners, produces an array of ribbing on both sides of the
large inner chassis and provides the rigidity required over the large
cross-sectional area of this repeater, which measures 103 inches wide
by 173 inches long.

The regulating repeater uses five thin-film hybrid-integrated circuit
(gic) amplifiers in addition to the preamplifier and power amplifier.
The five are flat-gain amplifiers, and their ceramic substrate size is
more nearly conventional, measuring 0.8 inch by 2.9 inches. Actually,
three designs, two of which appear twice as part of the two Vf net-

works,’ are used in the regulating repeater. In the Vf network, the
HIc’s are connected as slave-boards to epoxy-glass printed-wiring
network boards by means of 26-gauge leads on the uic’s. They are
formed to fit into holes in the printed-wiring board and are clinched to
touch land areas to which they will be subsequently soldered. The
substrates are structurally supported on transistor heat sinks that
enclose the discrete, heat-producing transistors and are fastened
directly to the can that houses the entire network. The heat sinks are
similar to those described for the preamplifier and power amplifier, ex-
cept that they are the single-cavity style. Here, too, conduction is the
primary mode of heat transfer. The heat from the transistors is
directed away from the substrate, through the heat sink into the
die-cast network housing and to the repeater framework, etc. Top
and bottom network covers are removable to provide accessibility

for trouble shooting but, more important, to allow assembly in a
prescribed sequence. The printed-wiring board is first mounted in its
frame, then the uic is fastened to the frame by the heat sink and finally
the nic leads are soldered to the board Jand areas. This assembly
sequence is essential to a stress-free assembly. Figure 9 shows the
collection of thin-film integrated-circuit substrates used throughout
L5. The 55-types and the 57-types are the ones described earlier for
use in the repeaters, while the remaining ones are used in flat-gain
amplifiers for terminal equipment described later in this paper.

To compensate in part for changes in cable loss because of changes
in cable temperature,? the regulating repeater requires access to
ground-temperature-sensing thermistors that are buried 15 feet from
the manhole at cable depth. The thermistors are contained in a cast-
epoxy cable stub and, at a distance of 15 feet, are unaffected by man-
hole temperature. A plug-and-jack connection through the apparatus
case chassis provides the thermistor connection to the repeater.

While the basic repeater is shipped to the field as a sealed unit, the
inside of the regulating repeater must be accessible for the addition of
line-build-out networks and for adjustments when the repeater is ac-
ceptance-tested in the field.? For this reason, the regulating repeater’s
two large covers are hinged as a safety feature to prevent exposure to
high voltage by the accidental insertion of a coverless repeater into a
live apparatus case. Manipulation of the bulky, hinged covers is
awkward, at best, during manufacture and field-acceptance testing,
but amounts to a relatively effective trade-off for a necessary safety
precaution.

High-voltage insulation lines the cavity on the inner chassis that
houses the coaxial jack used as the low-voltage access point for the
repeater. Even though the jack is electrically isolated from high voltage
by a transformer and is mechanically bonded to the outer housing
which is at earth or sheath ground, the insulation liner is provided as
a safety feature to protect personnel and equipment in the event a
fault should occur.

With equalizing repeater points located at 34-mile (maximum)
intervals, or roughly midway in the 75-mile (maximum) L5 power
spans, it is unlikely that, in conversion from L4 to Ld, they would
coincide with L4 equalizing repeater points spaced at 54-mile (maxi-
mum) intervals.* Therefore, the only constraint on the L5 equalizing
repeater, to comply with the modified-conversion design concept for L5,
is that similar apparatus cases be used for the sake of standardization.

However, L5 uses manually adjustable equalizers with less circuitry
than is required for the remotely adjustable equalizer used in L4. This
simplifies manhole arrangements in that the entire L5 equalizing
repeater was designed to fit in a single apparatus case, in contrast
with the two cases required to house the L4 equalizing repeater. For
a 22-tube coaxial cable, an equalizing repeater pre-cast manhole
measures 6 feet wide by 24 feet long by 8 feet high on the inside to
accommodate the layout for the 22 apparatus cases required to equip
the entire cable.

The equalizing repeater comprises three plug-in units—a regulating
repeater, a manhole E1 equalizer (mE1), and a fault-location oscillator
unit arranged in that order from left to right in the apparatus case.
As with the previous repeaters, these units are bolted to the apparatus

case chassis and are cantilevered inside the apparatus case. Equalizing
repeater points also provide access to buried-cable-temperature-
sensing thermistors.

The mel equalizer is a larger unit than the regulating repeater, and
was designed as a fabricated unit to keep its weight down. The unit
measures 12 inches wide by 17 inches long by 44 inches deep, and is
illustrated in Fig. 10. The array of individually packaged amplifiers
and Bode networks is visible. These use printed-wiring technology,
are interconnected by cables terminated in miniature coaxial connec-
tors, and are made accessible to the knobs of the mel faceplate by
insulated shafts that permit manual control of potentiometers for
equalizer adjustments. The knobs are ‘‘push-to-turn’”’ to prevent
inadvertent changes in adjustments, and universal couplings prevent
lateral forces from being transmitted to the delicate potentiometer
shafts. A power separation filter is contained in the lower cavity of the
MEI equalizer, and high-voltage insulation is provided by phenolic
strips. This unit, like the basic repeater, need not be opened in the

field and, for safety, it is sealed by having its covers secured with one-
way screws. The mounting base of the mE] is a one-piece aluminum
casting that permits the precise location of the mounting holes and
the coaxial plugs within the tolerances allowable for mating with the
apparatus case chassis.

Main stations in an L5 “backbone” route are single-story buildings
containing four general categories of main-station equipment—high-
frequency repeatered line, signal processing terminal, transmission
surveillance, and a fourth, broad or peripheral category comprising
such equipment as order wire, restoration, repeater acceptance test,
and equalizer adjustment. The main-station equipment, intended for
the central office environment, is bay-mounted in contrast to the
rugged, cast plug-in type equipment designed for the manhole en-
vironment. The main-station equipment is generally in shelves,
drawers, or panels in unequal-flange duct-type bays, on a 2-inch
mounting plate modular spacing, and forming a bay-front that is flush
with the bay uprights. The equipment space is 15 inches deep, the
front and rear guard rails are 2 inches and 3? inch, respectively, and the
duct formed by the uprights of two adjacent bays constitutes shielded
cabling space closed at the front and accessible from the wiring aisle
at the rear of the bays through the opening created by the smaller
flange (see Fig. 11). The space directly behind the cabling duct is
available for shelf interconnections, for overflow cabling if required,
and, in the case of a double bay, for additional equipment.

The deep equipment space is required because of the relatively
bulky, well-shielded apparatus assemblies used throughout the system.

This depth, in combination with the width of the standard bay, gives
rise to three-dimensional, high-density, equipment packaging which
saves coveted floor space in a main station but limits accessibility of
individual constituents of bay subassemblies. However, the limited
accessibility is tempered by the connectorized assemblies used through-
out the system to facilitate manufacture and maintenance.

Miniature coaxial connectors terminate individual pieces of ap-
paratus (amplifiers, filters, networks, etc.) to form precise, definable
boundaries for specific electrical requirements. These individual units
are arranged on a shelf and interconnected by miniature coaxial cables
terminated in the mating coaxial connectors. The shelf is screw-
fastened to the rear flange of the bay uprights, and is itself inter-
connected with small Bell System coaxial plugs and jacks to the bay
wiring harness for signal connections and with multicontact connectors
for power and alarm connections.

In some instances, plug-in modular printed wiring board (pws)
assemblies are arranged on a shelf and interconnected by gold-plated
fingers on the pwB and multicontact connectors that are part of the
back-plane wiring of the shelf.

High-density packaging tends to accentuate thermal considerations,
and limited accessibility coupled with high system capacity accentuate
reliability considerations, both prime considerations in the design of
L5 equipment.

The L5 bay height was standardized at 9 feet to accommodate
power-feed main and switching power-feed main-station buildings.
Terminal and terminal-main stations have higher ceilings and require
bays 10 feet, 6 inches and 11 feet, 6 inches high. For these, bay ex-
tenders are added to have the 9-foot standard bay fit in 10-foot, 6-inch
and 11-foot, 6-inch bay line-ups. Specific bays, intended for use only in
terminal and terminal-main stations and requiring the additional space
afforded by the higher ceilings, are 10-foot, 6-inch and 11-foot, 6-inch
designs. The new equipment building system (NEBS) requires that
future equipment be standardized on a 7-foot bay height, and a series
of 7-foot bays have been designed for L5.

For the L5 high-frequency repeatered line, the interface between
manhole repeaters and main-station equipment” is the power-separa-
tion filter (esr) cabinet (see Fig. 12). This cabinet confines all the

transmission area high-voltage equipment in one locked enclosure for
safety, and provides a single point to terminate the lead-covered,
pressurized, air-dielectric cable stubs coming from the multitube,
coaxial-cable splice in the station. This obviates the need to run the
lead cable and the high voltage in front of the line bays, as has been
done in the past. The run from the psF cabinet to the line bays is at
low voltage and uses single-unit air-dielectric and/or solid-dielectric
coaxial cable, depending on the length of the connection. Initially, it
was intended that the psr cabinet be located in close proximity to the
building cable entrance to minimize the run of lead cable in the
station. However, placing the psF cabinet at the recommended location
at the head-end of an L5 bay line-up optimizes the interbay cabling
for that line-up and, in most cases, precludes the need for the air-
dielectric coaxial-cable-run portion of the connection from psF cabinet
to the line bays.

The 9-foot-high esr cabinet can hold a maximum of 22 power
separation filter shelves required to equip an entire 22-tube cable. The
shelves are contained in the lower portion of the cabinet seen from the
rear in Fig. 12. The cable terminals are bolted to brackets in the top
portion. Connections are made by screw-type coaxial connectors for
the high-voltage runs and by plug-and-jack connections for the low-
voltage runs. The cabling is dressed in the ducts formed on the sides of
the cabinet by the standard bay uprights (used to structure the
cabinet) and the fabricated metal enclosure is assembled around the
bay. Bay extenders accommodate the 9-foot cabinet in higher than
9-foot equipment line-ups. The psr shelves are modular and can be
equipped on an as-needed basis.

The line-protection switching system (LPss-3)*° bay is the second
element in an Ld line-up, and is required at each end of an L5 switching
section (150 miles maximum), but not at the power-feed main stations
located at the 75-mile (maximum) points. This 9-foot bay contains a
preponderance of logic-type circuitry that is packaged in modular
fashion on plug-in PwB assemblies arranged on fixed equipment shelves
in the bay. The myriad interconnecting wires are formed into a sizable
cable harness, dressed in the cable duct, and connected to a terminal
strip at the top of the bay. Bay power is derived from plug-in dc-to-de
converters located and fused at the top of the bay. Status indications
and controls are concentrated in a large display panel located centrally
on the bay. Human-engineering design considerations predominantly

influenced both the physical arrangement of this panel and the selec-
tion of the lighted indicators and keys so that the system’s switching
state can be clearly communicated to the craftsperson. The presence
of plastic hinged protective covers on the more important controls
also provide the ‘‘keying’’ necessary to avoid inadvertent manipulation
of controls so as not to jeopardize service. Wire verification to ensure
the presence of all intrabay connections is required before shipment
of a bay containing the common equipment. Modular plug-in assemblies
can be shipped separately on an as-required basis as the system grows.

The line transmit-receive bays are the mainstay of the Ld line-up.
One line bay is required for each transmit-receive pair of coaxials
equipped in the system. This bay is essentially an equalizing repeater
with manually adjustable equalizer units, El and E2, in both the
transmit and receive sides and with the addition of a dynamic equalizer,
E38, in the receive side. This, together with the elements of line-con-
necting, powering, and alarm equipment, comprises the line transmit-
receive bay which appears at every main-station location. Four designs
accommodate the 9-foot requirements at power-feed and switching
power-feed main stations and the 10-foot 6-inch and 11-foot, 6-inch
requirements at terminal and terminal-main stations. The difference
among the four designs is the varying complexity of the elements of
line connecting, ranging from the very simplest in the power-feed
main-station design, where the transmit-receive bay is essentially a
one-way repeater, to the more complex one-way repeater of the
switching power-feed main-station design, which includes switches and
switch-initiator circuitry for Less, to the most complex terminal and
terminal-main station designs where elements of line connecting also
include blocking, adding, dropping, and branching circuitry for the
signal-processing features. This hierarchy of line bay designs is il-
lustrated in Fig. 18, with shading used to contrast the variable element
within the otherwise repetitive structures. Note that the only difference
between the two ‘“‘terminal or terminal-main”’ designs is that the 11-
foot, 6-inch bay has room for one additional line-branching unit.

The main-station transmitting and receiving repeaters together
equate to a regulating repeater used in manholes. The transmit and
receive elements have been split and repackaged in an equipment
shelf for bay-mounting (Fig. 14). Similarly, the main-station manually
adjustable equalizer (E1) contains the same elements as its plug-in
unit, manhole counterpart (Fig. 10), re-packaged in a fixed shelf for

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bay-mounting (see Fig. 15); and this unit, expanded to include addi-
tional networks and amplifiers required for additional bump shapes,
yields the design of E2 (see Fig. 16).

Two line bays are shown at the right in Fig. 17 as they appear at a
terminal or terminal-main station in an L5 five-bay line-up that con-
stitutes the equipment required to provide service on the first working

pair of tubes, with the first line bay being the protection bay and the
second the first regular bay.

The final step of multiplexing required to stack jumbogroup (sq)
signals for transmission over the L5 coaxial line is accomplished by
the jumbogroup multiplex® (sax). The smx is a completely solid-state
multiplex that utilizes thin-film hybrid-integrated circuit (Ic) ele-
ments contained in modular plug-in assemblies. The smx equipment
is mounted on a unitized bay framework shown in Fig. 18, 11 feet,
6 inches high by 52 inches wide by 15 inches deep. A complete bay
accommodates a maximum of three transmitting and receiving jumbo-
groups. Jumbogroup multiplex designs are also available in 10-foot
6-inch, 9-foot, and 7-foot bay heights to meet the needs of telephone
offices with lower ceilings.

The smx is a completely shop-assembled, shop-wired, and shop-
tested bay. It contains transmission, shaping, regulating, patching,
logic, alarm, carrier supply, and dc-to-de converter equipment and,
since 3600 voice circuits could be affected by a service interruption,
automatic protection switching is also provided for each jumbogroup.

Jumbogroup multiplex bays are generally located in the maintenance
aisle of a central office or main station where a minimum aisle spacing

of 4 feet, 6 inches is required to permit the use of rolling test consoles
without restricting the operating personnel. Since this area is a center
of office activity, a good appearance, without significantly increasing
the cost, was considered a design objective.

The bay cabling consisting mainly of office-type coaxial cable is
carefully segregated, primarily to minimize crosstalk and secondarily
to facilitate installation and shop wiring. Transmitting cables are con-
tained in the left-hand cable duct and receiving in the right-hand duct,

and, with few exceptions, all other wiring and cabling is confined to the
center duct. Office cable connections to the bay are made through
connectorized cables located at the top of the bay, making it unneces-
sary for the installer to enter the cable ducts.

The basic building block of the gmx bay is the jumbogroup shelf
assembly shown in Fig. 19, which contains the modular plug-in units.
For example, the receiving shelf contains the regulator, equalizer,
demodulator, de-emphasis, transmission protection switch, logic
switch control, and jack field. The framework and module construc-
tion for the transmitting side is the same.

Since all equipment modules performing similar functions are the
same physical size, a variety of bay configurations can easily be ac-
commodated. For instance, at a typical end office, the gmx bay with
jumbogroups 1, 2, and 3 could be provided, whereas at intermediate
stations the bay could contain all jumbogroups 1, 2, or 3 or any com-
bination of jumbogroups 1, 2, or 3 up to the bay capacity of three.

A lattice-type lightweight aluminum construction is used for the
jumbogroup shelf to optimize the strength-to-weight ratio and vir-
tually eliminate shelf deflection under load. To avoid scraping when
the aluminum modules are inserted in or removed from the shelf, a
low-friction tape is bonded to the load-bearing surface.

Each transmitting and receiving shelf assembly contains a built-in
jack field for jumbogroup patching or testing. To facilitate main-
tenance, a functional schematic diagram is provided directly above the
actual jack location. The jack field is recessed to minimize the pos-
sibility of a circuit patch plug accidentally being removed or damaged,
thereby causing a service interruption. To aid the craftsperson in
identifying jumbogroup equipment with its associated jack field, each
jumbogroup position is framed with a colored plastic strip. Each of the
three jumbogroup bay positions is identified with a different color.
Matching color route assignment cards covered with glare-resistant
plastic are also provided.

Directly above the writing shelves is a miscellaneous panel that
provides access to office trunks, alarm cutoff switches, and indicators
and the means for simultaneously testing all lamps in the bay. In the
present bay, incandescent lamps are used for alarm indications; how-
ever, in the very near future, these will be replaced by light-emitting
diodes (LED).

Modules are inserted from the front of the bay and held by a captive
screw in the rear of the shelf. These units are essentially plug-in, with-
out fixed mating connectors on the shelves. In this way, tolerance
problems associated with mating connectors are avoided. Most as-
semblies are contained in 10-inch wide by 4-inch high by 12-inch deep
aluminum modules, as shown in Fig. 20. Coaxial and pin connectors
are provided at the rear for transmission and power connections.

Filters, amplifiers, and other types of apparatus have been mounted
in separate housings for shielding, to facilitate manufacture and test-
ing, and for efficient field repair. Interconnections within the modules
are made by miniature coaxial cables.

During the development of the smx and other L5 main-station
equipment, it became apparent that there was a need for a family of
very small amplifiers. To meet this need, the 509-, 510-, 511-, and
512-series of HIc amplifier codes were developed. More than 230 of
these amplifiers are in a fully equipped ymx bay. The outside dimen-
sions of the amplifiers are 2.75 by 1.5 by 2 inches high. Input and out-
put connections are made through miniature 75-ohm coaxial connec-
tors. The circuitry of the amplifiers is a HIc with some components

appliqued, such as transistors, inductors, or chip-capacitors. Special
ground terminals are swaged into the chassis to ensure effective
grounding.

Considerable care was taken to limit the mechanical stresses be-
tween the ceramic substrate and the chassis. For example, all connec-
tions are made with soft gold-coated copper ribbon wire, looped to
allow slight mechanical movement of the substrate during thermal and
mechanical shock. As shown in Fig. 21, the substrate is mounted by
fitting the ends with silicone rubber gaskets and supporting them at
opposite ends of the chassis with adjustable aluminum blocks. To
efficiently conduct heat from the transistor, a spring-loaded heat sink
described earlier connects the transistor mounted on the substrate
directly to the cover of the amplifier.

As part of the overall smx development, a complete thermal analysis
of the bay was performed. Where possible, units dissipating the most
power were arranged in the bay to avoid hot spots. A 4-inch space is
provided for each jumbogroup shelf through the recessed jack field for
a front-to-rear air flow. Temperature measurements indicate a 55°F
maximum gradient between a transistor case in the smx and room
temperature. The complete thermal analysis shows that, even under
the worst condition, the equipment in the bay is well within the
temperature limits necessary for reliable performance.

The 24-volt power for the smx is supplied by the office power plant
over two separate feeders to the 24-volt to 25-volt regulated converters.

The feeder-converter assignments have been arranged so that the loss
of a feeder or a single converter will not result in a loss of service.

The interface between smx and the lesser units in the multiplex
hierarchy is the basic jumbogroup trunk bay’ (BseT). This bay offers
a very high degree of flexibility for the user in that it is designed to
allow a maximum of four inputs per jumbogroup and to include designs
for inputting radio signals, L-carrier mastergroup digital (LMD) signals,
L4 coaxial-carrier system signals, etc. The active circuits are re-
dundant and are packaged in pull-out drawers and fixed shelves.
The combinations of circuits available are so numerous that normal
bay coding or even normal panel coding is virtually impossible.
Only typical bay layouts are offered as a guide for ordering, and the
specific short- and long-range requirements of an office dictate the
composition of any BseT shipped from the factory. Intra-bay cabling
is laid out to accommodate multi-input options and suggested bay
layouts allow space for modular future growth.

The jumbogroup frequency supply (srs) furnishes three highly
reliable and precise signals to the ymx to generate the necessary car-
riers.’ These signals, at frequencies of 1.024, 2.560, and 20.48 MHz,
have an accuracy well within the one part in 108 requirement imposed
by the sax. This accuracy is realized by comparison with a reference
signal from a primary frequency source; however, even if the reference
signal is lost for several weeks, the inherent stability of the srs will
maintain the required accuracy.

The srs, shown in Fig. 22, is mounted in a unitized double 9-foot
high framework. A front aisle is required for normal maintenance and
a rear aisle for installation wiring. Since the number of srs bays re-
quired for coaxial systems is relatively low, the 9-foot bay arrange-
ment is provided for use in current design of main stations. A 7-foot
version of JFs is also available for future offices with the lower ceiling
height.

The srs is a fully shop-assembled, shop-wired, and shop-tested bay.
It also is completely solid-state, utilizing digital and thin-film hybrid-
integrated circuit (Hic) technology. The most important design ob-
jectives were precision and high reliability. To accomplish this, very
stable oven-controlled 39-type oscillators and one-for-one redundancy
automatically switched are employed.

The jumbogroup frequency generator (sre) is the basic building
block of the srs bay, and the 39-type oscillators are the heart of the
JrG. Each srs contains three sra’s (master, regular, and standby).
The master ura serves as a buffer between the reference signal and
both the regular and standby sre’s, thereby keeping the srs output
relatively immune from loss and/or disturbance of the reference signal.
As shown in Fig. 23, an integral part of the sre is the alarm display
panel, which indicates whether the sre is operating as the master,
in-service, or idle. It also indicates frequency offset, level failure, and
the frequency alarms. Test access points are provided for measuring
the 25-volt, 6-volt, and 5-volt power.

The sre patch panel provides patching facilities not only to remove
a failed sre from service but to rearrange the sra@’s by function (master,
regular, standby) to realize the best use of the remaining two. When a
JFG is removed from service, the protection switching logic must be
restructured to accommodate rearrangement of jre@ functions. This is
accomplished by operating the sre “‘out-of-service” switch on the alarm
and switch control panel when the patching has been completed. The
faceplate of the ura patch panel is a schematic block diagram with

appropriately positioned jacks interconnected with color-coded paths
to indicate the placement of patch plugs for the src, master, regular, -
or standby failure conditions. As in the mx, the panel is recessed to
prevent damage and possible circuit interruption.

The control center for the srs is the alarm and switch control panel.
It is a visual continuation of the patch panel schematic block diagram
and contains the level detectors and coaxial switch and the logic
required for the alarm and switch functions.

a ps
“CATION! {ED OEE ; CAUTION
So CRPRC AL oo PS dnt ep Bm penises wt : CRITICAL
SWITCHES 6° e HITCHES
eet ad Beg © a pes fe i

Most plug-in units in the srs follow the general main-station equip-
ment format; however, to meet very stringent circuit requirements, a
multilayer printed wiring board was used for the logic I unit shown in
Vig. 24 and the logic II and sre alarm units. The logic I printed wiring
board consists of four layers; the first two layers contain the signal
circuitry, the third a ground plane, and the fourth the power wiring.
This board contains over fifty integrated circuit packs and discrete
components. Connections are made through four multipin connectors
mated to the gold tabs provided on the board.

The complete power converter panels are provided, one per JFG.
Each converter receives power from a separate office feeder lead. This

ensures that, even if a feeder is accidently damaged or a main fuse
operates, the srs will continue in service.

The Ld surveillance’ physical designs cover the widest cross section
of types of equipment, ranging from the pure functional types located
in manholes to the sophisticated equipment located in the center of
activity in a control terminal or terminal-main station. In addition to
being influenced by functional design considerations, the latter type
of equipment was strongly influenced by human-engineering design
considerations and, since it is constantly in the forefront and under
scrutiny, was designed to have an aesthetically pleasing appearance.
While the repeaters may be the heart of the L5 system, the trans-
mission surveillance system (TSS) equipment is its nervous system,
feeding back information on the health and well-being of the system.

The fault-location oscillator (FLO) unit and the 31A gate are the
manhole-mounted units of the Ttss. A FLO unit is installed in every ap-
paratus case, a unit with eight outputs located centrally in a basic
repeater apparatus case (see Fig. 2, Section 1.1) and a unit with four
outputs for use with regulating and equalizing repeaters. The Frio is a
rugged unit made up of an inner chassis and an outer, fabricated,
shell-like housing. The inner chassis supports the face plate and con-
tains the four tone generators (two high frequency, two low frequency),
the combining circuitry, and the coaxial switches and associated logic
circuitry to turn on the unit. Connections within the unit are made
with miniature coaxial connectors and connections for the tone out-
puts are by way of large Bell System coaxial jacks located in the face
plate. The coaxial patch cords used to connect the FLO tone outputs to
the input and output of every repeater (through the twin coaxial-jack
assemblies located on the apparatus case chassis) are shipped with the
apparatus case and provide continuity through the apparatus case for
cable acceptance and corona testing. For this reason, the cords are
required to be high-voltage designs, even though they are primarily
intended to be used at low voltage. The protective, outer covering of
the FLo unit consists of a long, thin-walled, rectangular, aluminum
housing closed at one end, where it is welded to a rugged, cast, mounting
base. This base holds the multicontact connector for input power and
logic connections, and contains the large retainer guides that mate
with the retainer pins on the apparatus case chassis.

The 31A gate, the logic unit for the FLO’s, is used on a one-per-
manhole basis and is located in the cross-connect apparatus case
(Section 1.1). It consists of a large epoxy-glass PwB assembly housed in
a sheet-metal, fabricated can mounted on a hinged shelf that swings
out to allow access to wire terminals and protector blocks inside the
cross-connect apparatus case.

Rectifiers used to power the FLo units and control circuitry for the
FLO units are located in the main station in the line access bay that
serves as the interface between the line and the surveillance® equipment.
Connections to the surveyed functions in the line bays are made with
solid-dielectric, coaxial-cable runs to the line access bay where pro-
visions are made to store the excess cable created by the requirement
for equal-length, balanced connections. Switched access to the line
bays is provided by 1-by-12 dry-reed coaxial switches mounted on a
panel together with installer-wired arrays of coaxial jacks located on
jack strips designed to be removable from the panel for easy access.
“Hairpin-type” plugs are used for interconnections, and the entire
panel is recessed behind the bay uprights to protect the plugs from
inadvertent disconnects. The recessing is evident in Fig. 17, where the
line access bay appears as the third bay in the lineup. The lower part
of the bay contains the order-wire-access panel followed by several
fixed shelves containing cable equalizers. The last element in the bay
is a removable drawer-type storage shelf for the equalizer adjustment
unit (BAU) described below.

The transmission surveillance system is made up of the transmission
surveillance center (Tsc) and distribution bay combination for control
offices along an L5 backbone route and the transmission surveillance
auxiliary (TsaA) for other than control main stations.

The tsc is a large desk-type console that houses, in its sides, a
commercial computer, a transmission measuring test set, and associated
equipment. The central portion of the console contains the manual
control panel that displays an array of keys and status indicators within
convenient reach of an operator. The Tsa is virtually a Tsc without the
computer and is a standard 26-inch wide unequal flange duct-type bay.

Conduction and natural convection were inadequate to transfer heat
from the thermal sources within the tsc. To avoid local hot spots,

forced convection is required, and a fan is installed as part of the rsc
equipment.

Associated with a project of the magnitude of an L5 system is a
whole series of peripheral items of equipment too numerous to cover in
full detail here but of obvious importance to the system and its
workings.

Order-wire bays have been coded and, for the first time, make
convenient mounting arrangements available to the field where,
heretofore, only miscellaneously mountable equipment was provided.
In addition, connectorized versions of these bays have been designed
to avoid difficulties encountered during the Ld initial installation to
accomplish the myriad wiring connections for complete order-wire
fan-out in an office.

Multi-mastergroup restoration and zero-loss trunk bay designs in-
volve the use of inexpensive ‘‘pseudo-plug-in” construction where
simple modules are arranged on a shelf and connected at the rear of
the bay with jack-terminated cable arms that are part of the bay
wire harness.

Because of the relative inaccessibility of repeaters in manholes and
the expense involved in entering a manhole, repeaters that are fully
tested before leaving the factory are again tested at field maintenance
centers before being dispatched to the manholes. For this purpose, a
special repeater test stand and associated warm-up rack were designed
for repeater acceptance testing. The test stand simulates the apparatus
case chassis details, which make provisions for mounting the L5 man-
hole plug-in units for verification prior to installation. The verification
procedure is speeded up by pre-heating the repeaters on the warm-up
rack, which is comprised of five mounting panels, each of which can
hold two basic repeaters or one regulating repeater, yielding a total
capacity of ten basic or five regulating repeaters. The same retainer
guide pins that are part of the apparatus case chassis are used to secure
the repeaters on the warm-up rack panels. Powering is by a commercial
power supply, and the entire unit is mounted on casters for additional
flexibility in those offices designated as maintenance centers.

Required settings on manually adjustable E1 and E2 equalizers are
made using the equalizer adjustment unit (EAU) designed as a compact,
lightweight, portable test set (see Fig. 25). The unit uses pwB plug-in
modules, is designed for ease of assembly, offers accessibility for main-

tenance, and is stored in the pull-out drawer at the bottom of the line
access bay.

With the Bell System providing the major long-haul communica-
tions networks for the country, Bell Laboratories designs have always
been motivated toward reliability. Systems like the J- and K-carrier
systems designed in the 1930’s and installed in the 1940’s are still
providing reliable service. These systems, however, were limited to
transmitting 12 two-way telephone conversations over a cable pair,
whereas the new L5 carrier system transmits 10,800 conversations per
coaxial pair and 108,000 conversations on a fully loaded 22-tube
coaxial cable. To accomplish this, new more sophisticated technology
was required and to complement this effort a more definitive reliability
program was initiated.

We define reliability as ensuring that a given component, equipment,
or system will perform a required function, under stated conditions
for a needed period of time.

This comprehensive reliability program was instituted almost from
the start of the L5 carrier system development and has been proceeding
concurrently with it. This overall program covered the following
nine functional steps, some of which must still be completed and, there-
fore, will not be part of this report:

(t) Derive early estimates of equipment and system failure rates,
mean time to failure (mTTF), and availability.

(zi) Monitor the actual laboratory prototype experience.
(4i7) Analyze field-trial production experience.
(tv) Acquire and analyze actual field-trial data.
(v) Review and revise, if required, early estimates based on
laboratory and field-trial experience.
(vt) Monitor the Western Electric manufacturing and installation
initial route experience.
(vit) Conduct a full-scale reliability study of the initial service
route.
(vizi) On a continuing basis, obtain and analyze data on all Ld units
returned for repair.
(tx) Conduct a reliability study on a subsequent route, installed
about two years after the initial route.

The purpose and objectives of this program are to obtain accurate
data on the reliability performance of the L5 carrier system as early
as possible, to aid in formulating the early system design concepts, to
uncover potential problem areas so that corrective action can be
expedited before the first system is turned up for service, to compare
the actual system reliability performance with the early estimates, and
to review the reasons for any major differences, so that more accurate
estimates can be made on future systems.

Very early in the development program, a reliability study of the
L5 system design was made. As a first step, “‘black box’’ failure rate
estimates were made on all the repeaters and equipment units. While
it is impossible to predict for any individual electronics part either its
life or its rate of degradation under known stress, it is possible to treat
large populations of such parts on a probabilistic basis with acceptable
results. These results can then be related to the statistical behavior of
the system. The “‘black box’ failure rates were calculated by summing
the failure rates of the components and parts. The component rates
used were based on experience in well-defined systems, operating under
normal Bell System conditions and environment. For new devices or
technology with little or no previous experience, estimated failure rates
were derived after consultation with the device or component designers.
In these cases, the rates reflected a very conservative estimate.

Using the basic repeater which is the most numerous and least
sophisticated of the line repeaters as an example, it was estimated that
the failure rate would be about 900 rit’s, with a Fir defined as one
failure expected or experienced in 10° hours of service. This results in
an estimated mrtr of over 125 years for each basic repeater. The

regulating, equalizing, and main-station repeaters being substantially
more complex have shorter TTF estimates.

To obtain overall estimates, two basic system models were selected.
Both were 4000 miles long and employed automatic protection
switching for the coaxial tubes. In one case, the calculations were based
on switching section lengths at a maximum of 150 miles and, for the
second, a nominal 120-mile spacing was used, based on the L4 coaxial-
system route layout experience. There were some obvious trade-offs in
the two models selected. For instance, using the maximum length
section resulted in more line repeaters per switching section ; however,
in a 4000-mile route the number of main stations are reduced from
33 in a route with normal switching section lengths to 27 for the
maximum length sections.

An important consideration in the system availability estimates is
the average time to repair or restore service. Since it was difficult to
arrive at a time that was acceptable, calculations were made based on
optimistic and pessimistic restoral times. The optimistic times were
defined as an average of 15 minutes for main-station equipment
and three hours for manhole equipment, and the pessimistic times as
three hours for main-station equipment and six hours for manhole
equipment. Recent data on the service restorals for existing coaxial
systems show that the average repair times actually being experienced
are very close to the optimistic figures.

It is estimated that, for a pair of coaxial tubes in a fully equipped
22-tube cable, with a 1-for-10 automatically switched redundancy, the
availability of the electronics and switching contained in a 4000-mile
route using the optimistic restoral times would be 99.986 percent for
an average outage time of about 1 hour and 15 minutes per year. If
the pessimistic repair times are assumed for the same system, the
availability would be 99.948 percent per year for an average of about
4.5 hours per year outage.

One very early system question was whether, with double the
number of repeaters in a L5 switching section as compared with the
L4 system, the ratio of one spare coaxial automatically protecting ten
working coaxials was adequate to ensure reliable service. Availability
calculations were made on the system model with 120-mile switching
sections and optimistic repair times employing 1-for-10 and 2-for-9
protection ratios. With the 2-for-9 arrangement, the probability of
three simultaneous failures in a switching section was remote; how-
ever, the estimated outage attributable to the more sophisticated
switching system resulted in outage times for both strategies being

relatively close. Based on previous experience, cable failures constitute
a greater outage hazard in terms of system unavailability than do the
electronics. Therefore, using additional coaxials within the same cables
as protection for most cable faults would be counter-productive.
Taking into account these factors and the revenue lost by dedicating
a second coaxial for protection, the 1-for-10 protection ratio was con-
tinued for the L5 system.

The field trial of the L5 carrier system started on July 1, 1970 and
continued through June 1972 on a 125-mile cable route between
Cedarbrook and Netcong, N. J., shown in Fig. 26. The purpose of the
field trial was to test the L5 as a complete system operating in the
field environment before going into full-scale production and regular
service. The 240 basic repeaters and 48 regulating repeaters required
for the trial were constructed by Western Electric on a special basis
because the quantities exceeded Bell Laboratories model shop capa-
bilities. Since problems in the field can often be anticipated from
experience gained during manufacture, a data acquisition program was
initiated by Western Electric for the field trial equipment and, even-
tually, standard production.

A reliability study was made on the performance of the field trial
route. This was intended as prelude to a study of the initial L5
service route between Lillyville, Pa. and Hillsboro, Mo. Data were
collected on all failures of equipment and all system outages. A
thorough investigation was made on any units that failed. A case in
point was the failure of a basic repeater traced to the earth-ground
filter. After further analysis, including the X-ray photograph shown
in Fig. 27, it was found that high-voltage shorting was caused by a
capacitor unit assembled off-center without outer insulating material.
To prevent this in the future, high-voltage test screening was specified
and tighter quality controls were instituted by the supplier. Failure
rate calculations were made based on the field data; however, it was
recognized that the population was small, the equipment was not
from standard production, and, because of the level of field activity,
it was difficult to keep accurate data on every single occurrence.

Our experience with previous systems such as the L4 coaxial system
and the Ld field trial pointed out the need for even greater emphasis
on obtaining accurate data on the manufacturing testing results, the
initial defect rates in the field, the infant mortality rates (the infant
mortality period is defined as the first six months of service), and,
finally, the steady-state system reliability performance more commonly
known as the normal use period of the ‘‘bathtub” curve. To accomplish

@ MAIN STATIONS
mw POWER FEED STATIONS
@ EQUALIZING REPEATER
© REGULATING REPEATER
LIMITED ACCESS AND
DIVIDED HIGHWAYS
——— IMPROVED HIGHWAYS
— LOCAL AND CONNECTING
ROADS

this, an elaborate data collection program was initiated. Subunits such
as coded amplifiers and even Hic substrates were serial-numbered.
Special envelopes and data forms were attached to each unit at the
start of manufacture. As the unit was processed through the various
stages of manufacture, the test results were recorded on punched tapes
and stored in the envelopes. If components are removed at some stage,
they are stored in the envelopes and the data recorded on the forms.
When, for instance, a repeater is ready to ship, a complete pedigree
has been established for analysis and future reference. The components
removed are forwarded to the experts for analysis to determine the
failure mechanism. Through this program, we are able to detect
failure trends and initiate corrective action.

An important part of the reliability program is to establish the
initial field defect rate. In Fig. 28, a block diagram traces the pro-

cedures set up to process units returned for repair. Note that a “hot
line” was established by Western Electric to expedite the replacement
of any units returned for repair so that the service schedules would be
met. When units are returned for repair, the designers are able to
participate in diagnosing the problems. Data are recorded on all
repairs, and the components removed are kept for further analysis. For
the initial route, about 3200 basic repeaters were tested and installed
in manholes. Approximately 1.5 percent were returned for all reasons.
After further testing, it was determined that about 35 percent of these
met all test requirements, resulting in a 1-percent initial return rate.
On January 3, 1974, the L5 route between Lillyville and Morgan-
town, Pa., was put into regular service, and the Ld field reliability
program was started. On January 25, the section from Morgantown
to Hillsboro, Mo., was put into service, and this section was added to
the reliability study. Before the start of service, visits were made to
all offices on the route to discuss the objectives of the study and the
type of data required. During the early service period, each office is
contacted on a weekly basis to report all field problems, even those
not affecting reliability. The purpose is to get immediate feedback on
the system performance. A complete listing of all equipment in the
initial route by serial number was made so that the hours of service
necessary for availability calculation can be accurately computed.

J SWITCH TO PROTECTION COAX ‘one minute on mone!
(hutomatic (7 wanuat AUTO. LINE TERMINATE  [L] YES 0) no
LINE NO- REGULAR PROTECTION

REASON LINE FAILURE = [o> WAINTENANCE =] ResTOoRATION (]
(EXPLAIN IN IT) (EXPLAIN IN IIT)

IL.__CAUSE_OF LINE FAILURE IIL. MAINTENANCE IV. OUTAGE TIME CAUSED BY
(CINE EQUIPMENT FAILURE () woot ication CIN AODITION Te seer)
CI capie FArLuRE ( otter ([] pror.coax.IN USE
() TeRMINAL EQUI PuENT PROT.COAX.IN USE FOR
(JH V CONVERTER FAILURE RESTORATION
(JotHer causes (DO swITCHING FAILURE

DATE/TIME of FAILURE_DE> @ _I1G@2O0 * SEE INSTRUCTIONS FOR
are e320 EQUIPMENT TO BE REPORTED
DATE/TIME restoreo_Ub=2__ --
ATE
FAILURE SERVICE AFFECTING? Oves [Aft
NATURE OF FAILURE: CODE____ OTHER

LINE(S) FAILED LF a 4 DATE/TIME ———_ ——
LINE(S) RETURNED TO NORMAL oarte TIME
ee DATE/TIME ——_
Ce DATE/TIME ieee
_ Ly DATE/TIME ———

MAIL TO. MR. W)C. WESTPHAL, ROOM 3B13
BELL TELEPHONE LABORATORIES
4600 OSGOOD STREET
NO ANDOVER, MASS. 01845

A data form shown in Fig. 29 is filled out and returned to Bell
Laboratories for failures of any type, even if service was not affected,
and for the use of the spare line facilities for protection and for main-
tenance. These forms are returned to Bell Laboratories weekly for
compilation. Arrangements have been made so that complete data will
be recorded by Western Electric when the units are returned for
repair. At the time this paper was written, the system was in service

for a very short period of time and the field data were limited; there-
fore, no reportable results were available. This reliability study is
intended to continue for a minimum of one year.

As part of our continuing transmission system reliability program,
the general performance of the L5 system will be continually monitored
by our computer-aided reliability program (cARP), which utilizes data
received from the service centers on all units returned from any route
for repair. In addition, in about two years we intend to conduct a
reliability study similar to the one presently in operation on another
route.

Copyright © 1974 American Telephone and Telegraph Company
Tue Beuu SysteEM TECHNICAL JOURNAL
Vol. 53, No. 10, December 1974
Printed in U.S.A.

A family of ultralinear npn transistors has been developed for use in
the Ld coaxial-carrier system. These 3-GHz devices are characterized by
extremely low distortion and noise figure. The transistor comprises an
interdigitated base-emitter structure with a heavily doped base grid con-
nected to the peripheral base metal contact. The emitter contact ts overlaid
on the base-emitter region. Contact metallurgy consists of a platinum
silicide, titanium, platinum, and gold system. The transistor ts a highly
reliable device and meets all the performance requirements of the Ld
system.

To realize the circuit performance anticipated by raising the coaxial
carrier’s highest message frequency from the 17.5 MHz of L4 to the
60.5 MHz of L5, it was necessary to develop a new family of npn
transistors, the 76 and 77 types. The gain bandwidth product fr had
to be nominally 3,000 MHz; three-tone third-harmonic distortion
M 3x had to be less than —88 dB below 1 mW and the high-frequency
noise figure (NF) kept below 2.4 dB at 30 mA. Power dissipation was to
be 1.7 watts, which amounted to a considerable improvement in the
state of the art.

The 76-type transistors are used in the equalizing and regulating
repeaters and the 77-type in the basic repeater. The 77’s are used as
matched pairs to lower intermodulation distortion and the noise figure.

One basic design is used to meet all of the equalizer and repeater
requirements. Six codes were developed in all, each of which was
chosen for a specific purpose, such as low noise for the input stage of
the basic repeater, and low distortion for the output stage. Typical
characteristics for two codes are given in Table I.

Some early studies indicated that of the common geometries of
power transistors, an interdigitated structure would have the best
distortion performance; thus, the present devices are made with
interdigitated base and emitter diffusions, but with an overlaid emitter
contact. Base current is carried by the heavily doped base grid, under
the overlaid contact, and out to metallized regions at the sides. See
Fig. 1. This structure avoids metallizing problems caused by tight
metal-contact tolerances, and it minimizes the total area. Emitter
bonds are made over the active region, eliminating the emitter pad
and its parasitic capacitance. The emitters are 2.5 um wide and 105 um
long.

The epitaxial thickness is a compromise between being large enough
to provide sufficient series collector resistance to minimize second-
breakdown problems and yet not so large as to degrade device
performance.

Following the practice set by the L4! system, the transistors are
encapsulated in a small metal-ceramic package especially designed for
rf use. It comprises a kovar-beryllia structure that minimizes parasitic
capacitance and lead inductance and provides a relatively low thermal
impedance of 30°C/watt. The latter characteristic is important if
high reliability and low distortion (see Fig. 2) are to be obtained, since
the output transistors in the basic repeater will operate at 1.7 watts in a
peak ambient of 85°C.

The principal contributor to system noise from the output stage
of an amplifier is third harmonic distortion, M3z. In a transistor, this
noise arises from two main sources: from the essential nonlinearity of

the emitter current-voltage characteristic and from the falloff of fr with
current? owing to conductivity modulation of the collector-junction
space-charge region, or Kirk effect. The emitter nonlinearity is the
same exponential behavior of the forward-biased diode used, for
example, in mixers and for harmonic generation. The nonlinearity is
reduced by increasing the total emitter current. Poon? has shown that
if the emitter area and collector current (effectively the emitter
current) are both doubled, M32 decreases by 12 dB (see Fig. 3). If the
current density becomes too large, however, distortion arising in the
collector starts to degrade the device performance. Whenever the
current density exceeds qvN, where q is the electric charge, v the
carrier velocity, and N the density of fixed impurities at the collector,
the collector-base junction will move in towards the collector substrate
and cause fr to decrease. According to Poon,? in the Ld case there is a

It can be seen that any mechanism, such as the Kirk effect, which
increases the curvature of the fr vs J, characteristic, will contribute to
harmonic distortion. This means, of course, that as the emitter current
is increased to minimize the emitter-distortion term, the collector area
must increase proportionately. At some point it ceases to be economic

Fig. 3—Harmonic distortion as a function of collector current for several device sizes.

to increase the emitter current because both capacitance and dissipated
power become excessive. For the transistor size chosen, a current of
110 mA provides the lowest distortion.

The sustaining voltage for the L5 transistor turns out to be bounded
at both limits. At the lower limit, it is necessary to keep above the
supply voltage, so a value of 18 volts minimum is set. At the upper

limit the doping may become too low and a Kirk effect may occur if the
value of the sustaining voltage is made too high. This result can be
seen from the following relationships‘

where
Nezc is the collector epitaxial doping
N, is the surface impurity concentration
x; is the junction depth
n is a constant ~ 4 for the transistor concerned.

One of the basic differences between L4 and L5 transistors is brought
about by the increase of the highest message frequency from 17.5
MHz to 60.5 MHz. To accommodate the increased frequency, it is
necessary to design the transistor to have a nominal gain bandwidth
cutoff, fr, of 3 GHz. The solution to fr is given by

1
a fe th Fase Fe
where
Te = emitter time constant
T» = base time constant
tT; = collector depletion region time constant

where
K = Boltzmann’s constant
T = absolute temperature
q = electronic charge
J. = collector-current density
Cre = emitter capacity
W, = base width
W. = collector width
D, = diffusion constant for electrons
—_ Veo + ae
J = peW.
Ve» = applied voltage
V, = built-in junction potential
v = carrier velocity
Since r, dominates the frequency response, the necessity of keeping
the current density low can be readily seen.
Under this condition, rz, 72 and 7, reduce to
We , Ws
Oe MDs a Vs
a a 1 —J./Jo
= 20, | 1 — J./qu:Nec
| W. 1— fe |
Epc ’

where
v; = scattering limited velocity
Xmo = depletion layer width
e = dielectric constant.

Once the current density has been chosen to be suitably low, the only
available design factor for lowering fr is the actual base width. To
achieve an fr of 3 GHz, it is necessary to lower the base width to
0.27 um nominal.

Neilsen’s relation’ for the noise figure, NF, provides a simple means
of determining the choice of design parameters for minimizing the

R, = generator impedance of 50 ohms
re = emitter resistance = KT/qI.
fe = frequency at which a is 0.707 of its de value

It is apparent that once current, gain, and operating frequency have
been chosen, 7; is the only parameter at our disposal and, in fact, for
the L5 transistor, the r, term is the largest contributor to the noise
figure. The term 7; is made up of two parts. One is the lateral resistance
under the emitter stripe, which is minimized by using extremely
narrow emitters 2.5 um in width. The outer component consists of
the series resistance to the external base contact. This resistance is
reduced to as low a level as possible by the use of a very heavily doped
base grid.

During development of the 76- and 77-type transistors, several
hundred devices were subjected to accelerated aging of both shelf-
temperature and power at junction temperatures up to 300°C. Analysis
of the data predicts that the transistors will have a failure rate of
considerably less than 10 fits at an operating temperature of 150°C.
Under these conditions, it is no longer reliability that is the determinant
for system success, but rather, the temperature dependence of the
various device parameters.

A highly reliable ultralinear family of transistors has been developed
for coaxial-carrier-system use. Its high gain-bandwidth product, low
noise, and low third-harmonic distortion allow the L5 system to
successfully meet its operational requirements with an upper master-
group frequency of 60.5 MHz.

1. N. J. Chaplin, G. A. Dodson, and R. M. Jacobs, ‘‘L-4 System: Solid State De-
vices,”’ B.S.T.J. 48, No. 4 (April 1969), pp. 983-992.

2. H. C. Poon, “Implication of Transistor Frequency Dependence,” IEEE Trans.
Electron Devices, ED-21, No. 1 (January 1974), pp. 110-112.

3. C. J. Kirk, Jr., “A Theory of Transistor Cutoff Frequency (fr) Falloff at High
Current Densities,” IRE Trans. Electron Devices, ED-9, No. 2 (March 1962),
pp. 164-174.

4. A. B. Phillips, Transistor Engineering and Introduction to Integrated Semicon-
ductor Circuits, New York: McGraw-Hill, 1962.

5. E. G. Nielsen, “Behavior of Noise Figure in Junction Transistors,’’ Proc. IRE,
45, No. 7 (July 1957), pp. 957-963.

Copyright © 1974 American Telephone and Telegraph Company
Tue Bei System TECHNICAL JOURNAL
Vol. 53, No. 10, December 1974
Printed in U.S.A.

The development of transmission networks and magnetic components
for the LS system represents the largest network development project of
its type ever undertaken within the Bell System. Over 200 different coded
designs of networks, requiring in excess of 40 man-years of effort, were
required to meet the frequency-selective and signal-shaping requirements
of the system. Despite this effort, neither systems requirements nor systems
schedules could have been met without significant contributions from allied
technologies. This article identifies those technologies and describes design
techniques that have advanced the state-of-the-art capabilities in trans-
mission network and magnetic component design.

As in other analog systems, transmission networks of the L5 Coaxial-
Carrier Transmission System perform the indispensable functions of
frequency selection and signal shaping. Without the filtering functions
provided by certain of these networks, the basic multiplexing arrange-
ments on which the system depends could not be realized, nor could
the various fault-locating, equalizing, regulating, and switching pilots
be effectively separated from the message portion of the line signal.
Similarly, without use of the wide variety of equalizers provided, the
amplitude distortions introduced as the signal traverses the line could
not be as effectively corrected. Examples of such equalizers include
line-build-out (LBO) networks to compensate for cable spans shorter
than the nominal one-mile spacing, fived equalizers to compensate for
time-invariant and predictable effects, and adjustable equalizers to
reduce the unpredictable, statistical variations of the system.

To complete a listing of system functions performed by the trans-
mission networks of the L5 system would require a tabulation of all
major system functions. Over 200 transmission networks, requiring
in excess of 40 man-years of effort and representing the largest network
development of its type ever undertaken within the Bell System, were
developed for L5. Despite this large and sustained effort, the required
designs could not have been produced without significant advances in
computer-aided design, precision measurement, magnetic components,
and monolithic crystal filters. The purpose of the transmission net-
works section of this paper is to identify those technologies that have
influenced network developments for the L5 system and to describe
certain techniques that have advanced the state-of-the-art capabilities
in network design.

The majority of networks for the L5 system were developed using
classical network synthesis procedures. Insertion-loss synthesis tech-
niques were used exclusively for filter realization, Bode’s semi-infinite
slope approximation for LBo networks, cable equalizers and artificial
lines, and constant-resistance bridged-T networks, in various topolo-
gies, for fixed and variable amplitude equalizers. State-of-the-art
advances stemmed primarily from the stringent requirements imposed
by the system rather than from network theoretic developments.
Strong interaction between systems and network engineers, in frequency
allocation and the setting of reasonable guard bands, assisted greatly
in the problem of filter realizability. In practically all cases, develop-
ing techniques for reducing circulating currents, and developing prac-
tical design procedures for the compensation of parasitics were a
necessary adjunct to the physical realization of manufacturable
networks.

The development of modulation filters for the jumbogroup multiplex
terminal (smx)! is typical of that for the many filter designs produced
for the L5 system. Figure 1 is a simplified block diagram of the gmx
terminal as viewed by a networks designer. The modulation scheme
is typical of most analog multiplex systems except for the multiple-
modulation steps. Although the double and triple steps of modulation
increase the number of filter designs required, they reduce the com-
plexity of the individual filters. The final circuit and networks objec-
tives were the result of extensive discussions between systems and

Fig. 1—Simplified block diagram of mmx terminal emphasizing transmission networks.

networks engineers with the goal of optimizing the trade-off between
frequency-spectrum utilization and the setting of adjacent jumbogroup
guard bands.

As a result of such discussions, typical filter transition bands ap-
proximated a reasonable 10 percent, which allowed the use of small
slug-tuned, air-core inductors having a maximum quality factor (Q)
of approximately 150. All 14 of the filter designs—10 complex bandpass,
3 low-pass, and 1 low-pass-high-pass filter—are equal-ripple insertion-
loss designs. The degree of the filters ranged from a minimum of n = 7,
for a simple low-pass filter having a relatively low stopband objective
and a wide transition band, to a maximum of n = 20, for a more
difficult bandpass design having an 80-dB discrimination objective
across the unwanted sideband. Figures 2 and 3 show the schematic dia-
gram and measured characteristics, respectively, of one of the more
complex bandpass filters.

The initial design phases were concerned with the optimization of
the filter designs for practical element values, low passband delay
distortion, good out-of-band performance, and acceptable inband
return loss. Versatile insertion-loss synthesis, pole-placing, and analysis
programs were available.? Some of the features of the programs found
particularly useful during the design stage were: (¢) the ability to
choose arbitrary stopbands with Chebychev passband behavior, (72)
the ability to order loss peak removal to optimize element values, and
(tit) the automatic computation of network characteristics subsequent
to the solution of the synthesis problem.

Extensive use was also made of available programs to minimize
the delay distortion contributed by the modulation filters for any
future application of the mmx terminal to data systems. Theoretical
cutoffs were placed as far into the transition region as possible without
unduly increasing the degree of the filters or making the element
values impractical. This modification of the designs achieved up to a
50-percent reduction in delay distortion over the desired portion of
the passband with only a 10-percent increase in the complexity of the
filters.

As a result of the mx modulation arrangement, the carrier, in many
cases, fell close to the passband edges, making it impossible (with the
extended theoretical cutoffs) to adequately suppress the carrier leak.
Quartz crystals, as shown in the schematic of Fig. 2, were therefore
introduced in various shunt branches to suppress the unwanted
carrier. Highly stable, high-Q, AT-cut crystals, designed for maximum
suppression of unwanted responses, were used. Other components

Fig. 3—Characteristics of typical mx bandpass filter. (a) Component of loss
Sar pags by crystals at 42.496 MHz. (b) Stopband loss. (c) Return loss. (d) Pass-
and loss.

included adjustable air-core inductors and dipped-mica and tubular
capacitors. These components were mounted on printed wiring boards
(pwp’s) and packaged in sealed enclosures.

Laboratory development of the smx networks was a difficult and
lengthy process because of the wide bandwidths and high frequencies
involved. Precision scanning network analyzers were used widely in
the initial stages of development, but in all instances the computer-
operated transmission-measuring set? (corms) was used for final
characterization.

Monolithic crystal filters‘ have found wide application in the L5
system as narrow-band pick-off filters and as wide-passband band-

elimination structures. The monolithic crystal filter was first intro-
duced in the Bell System plant as simple two-resonator bandpass
filters for carrier-supply applications’ in the L4 coaxial system. For
the L5 system, over 40 designs, ranging in frequency from less than 3
MHz to nearly 100 MHz, were required for both bandpass and band-
elimination applications. To cover this frequency range, fundamental
and third-overtone crystal designs were necessary, and multiresonator
filtering arrangements were provided to meet the high out-of-band
suppression requirements.

The monolithic crystal filter, in its most elementary form, consists
of a quartz plate with an array of two-electrode pairs placed on its
major surfaces, with each pair establishing a resonator system. The
resonators are acoustically coupled via the elastic quartz plate. If
the quartz plate, electrode pairs, and acoustic coupling are properly
proportioned, a self-contained, two-port piezoelectric device, having
prescribed filtering characteristics, results. Higher-order filter func-
tions may be obtained by deposition of multielectrode pairs on an
extended quartz plate, with adjacent resonators acoustically coupled.
As an extension of this arrangement, multiple quartz plates may be
used, with adjacent resonators (on facing quartz plates) coupled
electrically (capacitively). A third alternative, the composite mono-
lithic crystal filter, uses Lc filter sections to couple individual two-
resonator monolithics. The latter approach was used exclusively in
the design of more than 30 monolithic crystal bandpass filters for L5
applications. Impedance-matching sections necessarily were included
in these designs to match the impedances of the coupled-resonator
designs to specified source and load impedances. This combination
of Lc coupling networks, Lc impedance-matching networks, and two-
resonator monolithics led to the concept of the monolithic crystal
filter (MCF) as a two-port, building-block device available for inclusion
in some larger frequency-selective network.®?

Before describing those methods of design peculiar to the composite
McF, a more detailed summary of methods of deriving higher-order
filter functions is required. Figure 4 shows the three different tech-
niques described above. In each instance, an impedance-matching
network is indicated at the input and output ports of the networks.
These may or may not be required. Figure 4a shows the multiresonator
MCF on a single piece of quartz with the interresonator coupling
supplied by the built-in acoustic coupling properties of the device
itself. To avoid acoustic coupling of certain unwanted modes, the
approach indicated in Fig. 4b is sometimes advisable. In this instance,
the single plate of Fig. 4a has been split into two parts and the facing

Fig. 4—Methods of deriving higher-order filter functions. (a) Multiorder mono-
lithic filter with acoustic interresonator coupling. (b) Multilithic filter with acoustic
and electrical interresonator coupling. (c) LC network coupling of two identical Mcr’s.

pairs of electrodes electrically coupled by the capacitor are shown in
the figure. The present design of the A6 channel-bank filters® uses this
technique. The third method of realization is represented in the
schematic of Fig. 4c. This schematic emphasizes the concept of the
McF as a two-port, building-block device employing supplementary
Lc filter sections (coupling networks) and impedance-matching net-
works for realization of a particular transfer function. In all three
examples shown in Fig. 4, the classical work of Dishal® provides closed-
form solutions in the design of narrow-band bandpass filters to exact
amplitude characteristics.

In view of the importance of the building-block concept to the
design of composite McF’s, devices and filter sections required for
filter realization are shown in Figs. 5 and 6. Emphasizing the role of
the mcr as a two-port device in composite-filter design, schematics
for the basic and reversed-phase connections of a symmetrical, two-
resonator McF and their equivalent circuits are shown in Fig. 5. The
associated Lc networks required in a composite filter design are shown

in Fig. 6. These include both impedance inverters and impedance-
matching networks. The impedance inverters are shown in their T- and
pi-configurations and consist of positive and negative capacitive
elements. In the final network design, the negative capacitances shown
in the figure are either absorbed in positive capacitances existing
within the network, or are approximated in a narrow-band sense by
inductances.

The impedance-matching networks of Fig. 6 are shown in their
L-configurations. It is important to note that the shunt elements of
these networks must also be designed to ‘‘absorb”’ the input or output
static capacitances of the associated McF’s.

Given the equivalent circuits of Figs. 5 and 6, design techniques
applicable to multiresonator bandpass filters become readily apparent.
A familiarity with the theory of coupled-resonator design, however, is
assumed. References 9, 10, and 11 provide excellent introductions to
the theory. Starting with a general n-resonator structure, the equiv-
alent, composite McF is easily derived by a series of network de-
compositions as shown in Fig. 7. A filter design having equal inductance
values throughout the structure is indicated. As shown in the figure,
impedance inverters, having element values corresponding to the
shunt capacitances of the structure, have been introduced into the
circuit at each of the shunt branch points. Certain sections of the

~ PLATE Co
7 aaa
{ \
stleaet. sees lvate M yasonton a Rte szsouaon
PAIR NO. 1 PAIR NO. 2 COUPLING
(a)
Cy -Cy -Cy Cy

Fig. 5—Schematics and equivalent circuits of monolithic crystal filters. (a) Basic
bandpass filter. (b) Reversed-phase bandpass filter.

resulting network can be identified as individual, two-resonator
mcr’s. Impedance inverters coupling the mcr’s are also easily identified.
If a composite filter design is sought, the impedance inverters indicated
in the figure must be changed to their equivalent pi-configurations and
the negative capacitance appearing in the series arms replaced by
equivalent inductances on a narrow-band approximation basis. The
final circuit arrangement is shown in the lower schematic of Fig. 7.
It should be noted that, for the composite filter structure comprised
of individual two-resonator mcr’s, the design must be constrained to
an even number of resonators. .

The insertion-loss characteristic of a composite monolithic crystal
bandpass filter, designed for use as a pick-off filter in the regulating
repeater of the L5 system, is shown in Fig. 8.

imMpEDANcE— | | | COUPLING | | | IMPEDANCE—
MATCHING R, NETWORK Ro MATCHING
NETWORK NETWORK

Fig. 7—Decomposition of an n-resonator bandpass filter to an equivalent, com-
posite, monolithic crystal filter structure. (a) n-resonator bandpass filter. (b)
Equivalent, composite, monolithic crystal filter. (c) Final composite filter.

Although the development of the monolithic crystal bandpass filter
was first reported in early 1965, it was not until approximately three
years later that the first disclosure of the monolithic band-elimination
filter (BEF) was made. Two types of monolithic BEF” may be derived
from monolithic crystal bandpass filters simply by the addition of
inductive or capacitive elements connected across the ungrounded
electrodes of conventional two-resonator monolithies. Different connec-
tions of the mcr plates, however, must be used to effect proper phase
relations in the two types of filter. These phase relations are obtained
by use of the basic or the reversed-phase connections previously

shown for the bandpass filter of Fig. 5. Network schematics and
equivalent circuits for the inductor- and capacitor-derived BEF’s are
shown in Fig. 9.

For the inductor-derived filter, the reject frequency occurs when the
magnitude of the reactance of the bridging inductor equals that of
the acoustic-coupling capacitors of the mcr. This reject frequency
occurs at the center frequency of the bandpass filter. For the capacitor-
derived filter, the reject frequency is obtained when the magnitude of
the bridging capacitance is made equal to that of the acoustic-coupling
capacitors.

The band-reject properties of either type of BEF are easily demon-
strated by application of Bartlett’s bisection theorem. The equivalent
lattice of either network is first derived from the open and short-
circuited impedances of the bisected network. Combining the re-
dundant elements in the series and lattice arms of the lattice and

plotting reactance as a function of frequency, the passbands and
stopbands can be readily identified. For the inductor-derived filter,
the insertion loss characteristic includes a passband on either side of a
reject band, with a second reject band extending to infinity beyond
the second passband. The inductor-derived monolithic BEF is, therefore,
a bandstop filter imbedded in a low-pass filter. The capacitor-derived
filter, however, is a bandstop filter imbedded in a high-pass structure.

For practical applications, the inductor-derived filter exhibits
superior passband performance compared to its capacitor-derived
counterpart. Consequently, the remaining discussion of the monolithic
BEF will be restricted to the inductor-derived circuit. The initial
design procedures, used successfully on both single-section and multi-
section monolithic BEF’s, were based on dividing the frequency
spectrum into two regions: (7) the stopband, over which the motional
reactances of the mcr change rapidly as the crystal passes through its
resonances, and (72) the passband, over which the motional reactances
of the mcF are sufficiently high to be considered as open circuits. The
extreme stiffness of the resonators makes it possible, therefore, to
attribute the frequency response, above and below the narrow stop-
band, solely to the low-pass filter section of the network. The narrow

9—Monolithic crystal, band-elimination filters. (a) Inductor-derived filter.
(b) Ganache filter.

stopband, however, is controlled by the interactions of the mcr and
the bridging inductor.

Single-section and multisection, inductor-derived, monolithic crystal
BEF’s, shown schematically in Fig. 10, may therefore be designed
using standard low-pass synthesis techniques to meet passband
requirements. The narrow reject band is formed by adding an esti-
mated number of mcr’s, with each monolithic bearing the proper
relationship to the corresponding bridging inductor. The stopband
performance can then be evaluated from an equivalent-circuit model
of the entire network with crystal units being added or removed until
the desired characteristic is obtained.

An alternative design technique to that outlined above results in a
closed-form synthesis procedure in which a set of filter requirements
such as passband ripple, cutoff frequency, and stopband bandwidth
are related directly to the set of equivalent circuits shown in Fig. 11.
This figure clearly delineates the low-pass and monolithic-crystal
sections of the composite filter.

Fig. 10—Inductor-derived, monolithic crystal Brr’s. (a) Single-section filters. (b)
Multisection filters.

Writing the cascade matrix for each of the component networks
of the composite structure, the complete matrix may be formed by
converting each of the component matrices to its equivalent admittance
form, adding the two resulting matrices and converting the sum back
to the cascade form. Such a procedure is essentially routine and
provides little insight into the synthesis procedure. If, however, these
matrices are expressed in terms of dimensionless parameters" and a
normalized frequency variable, the elements of a synthesis procedure
become apparent. In each instance, the several dimensionless param-

EQUIVALENT CIRCUIT OF INDUCTOR—DERIVED, MONOLITHIC
CRYSTAL, BAND-ELIMINATION FILTER

Q/K,2 WHERE Q = f/fy - fosg = 2(f-fol/fp *
AND K,9= 1/(wo? Cy YLiL2 )
* NARROWBAND APPROXIMATION
DIMENSIONLESS PARAMETERS AND DEFINING NETWORK SECTIONS
(b)

Fig. 11—(a) Equivalent circuit of inductor-derived, monolithic crystal, band-
elimination filter. (b) Dimensionless parameters and defining network sections.

A complete discussion of the cascade matrix approach to the design
of composite monolithic crystal BEF’s is beyond the scope of the
present paper; however, the usefulness of the method in describing
seemingly anomalous behavior will be illustrated. A particularly
interesting example is that of a two-resonator monolithic BEF having
an asymmetric stopband and a Butterworth passband with selected
parameters « = 0.5, D = 1 (see Fig. 11), and with f, < f.., where f.
is the reject frequency and f.. is the cutoff frequency of the low-pass
filter. The resulting characteristic of this design is shown in Fig. 12.
The solid-line plot is the insertion-loss or stopband model predicted
on the basis of the foregoing development. The dashed-line charac-
teristic is that of the Butterworth low-pass filter, and the circled
points indicate the characteristic as derived from computer-based
mesh analysis of the complete equivalent circuit. Two transition re-
gions are to be noted: the first, below the reject frequency, f., and the
second, above the reject frequency. In the region below the lower
transition region and above the upper transition region, the circled
points fall almost exactly on the computed characteristic of the low-
pass filter by itself. In the region indicated as the stopband model
region, the characteristic predicted by the cascade-matrix method of
analysis is duplicated. In the transition regions, however, the approxi-
mation is not nearly as good. To simplify the analysis, it was assumed
that the reactances of the low-pass filter remained constant in a
narrow-band approximation. This approximation breaks down, how-
ever, over the transition and passband regions of the per. The results
of the mesh-analysis computer runs, nevertheless, agree with the
predicted characteristic in the stopband to at least three significant
figures. The smoothest transition regions are attained in symmetric
stopband designs when the reject frequency is placed at an insertion
loss zero of a Chebychev low-pass filter. This is illustrated in Fig. 13
for the case of a 0.1579-dB ripple Chebychev passband.

Requirements for some of the BEF’s specified for the L5 system
included both high-loss stopbands (50 to 80 dB) and very wide pass-
bands (approximately 70 MHz), coupled with low-distortion and
high-return-loss requirements. Using Chebychev low-pass filters to

Fig. 12—Characteristic of band-elimination filter with asymmetric stopband and Butterworth passband. Transition regions are

Fig. 13—Characteristic of band-elimination filter with symmetric stopband and 0.1579-dB-ripple Chebychev passband. Transition
regions are indicated. Stopband center frequency f, is at insertion loss zero of the low-pass filter.

model the low-pass section of the composite filter, it was found possible
to meet all systems requirements. A specific example of design capa-
bility is given in Fig. 14 for a 42.880-MHz BEF, using three two-
resonator monolithics imbedded in a ninth-order Chebychev low-pass
filter. This low-pass filter was designed at an impedance level of 450
ohms to achieve optimal inductance values for the mcr’s. The filter
was then matched to the 75-ohm source and load impedances by use
of auto-transformers, as shown in the figure. Impedance-matching
sections, previously described for bandpass designs, could not be used
because of the narrow-band approximations involved. To meet the
passband insertion-loss deviation requirements of +£0.05 dB from flat
response over the entire frequency range of the L5 system, a simple
bridged-T amplitude equalizer was also required.

In the L5 coaxial system, a total of 28 adjustable bump shapes are
provided to correct for time-invariant amplitude distortions.!” Ten of
these shapes are allocated to the E1 equalizer for coarse correction of
distortions introduced by variations in the manufactured product.
The remaining 18 shapes are provided by the E2 equalizer to correct
whatever residual distortions remain after employment of all lower
levels of equalization, including the E1 equalizer.

The majority of bump shapes are realized by series-type, ae
bump, adjustable Bode equalizers. The remainder are supplied by
shaping networks introduced into the feedback paths of isolating
amplifiers. The amplifiers buffer the input and output ports of each
Bode network and supply gain to offset the losses introduced by the
equalizers. Each of the multibump equalizers, however, introduces no
more flat loss than would have been introduced by any one of the several
single-bump equalizers required to duplicate its performance. With
a reduction in the number of Bode networks, the number of isolating
amplifiers is also reduced. This, in turn, results in a reduction in inter-
modulation and thermal noise. Finally, the introduction of shaping net-
works into the feedback of the isolating amplifiers also reduces the
number of Bode networks required. The net effect of all these consider-
ations is to improve reliability, to reduce cost, and as previously in-
dicated, to reduce intermodulation and thermal noise.

All of the multibump Bode networks supplied for the E1 and E2
equalizers were derived from extensions of the simple series-type
adjustable network of Fig. 15a. The basic shaping network, as indicated
in the figure, is the bridged-T equalizer. Figure 15b shows the range
of insertion-loss shapes that may be obtained by variation of load
resistor, Ra.

As demonstrated by Lundry,!® the insertion loss of the generalized
Bode network of Fig. 16a can, to a good approximation, be expressed as

6 — 00 = K*pae*?, (1)
where
@ = the image transfer constant of the bridged-T network
6 = the insertion loss of the adjustable equalizer
6. = the insertion loss of the equalizer when Rz = R,
eo — ]
ef + 1

Fig. 15—Series-type, single-bump, adjustable, Bode equalizer. (a) Schematic.
(b) Characteristic.

where p(f) is the input reflection coefficient of the shaping network
relative to R, and is a function of frequency. Considering only the real

Fig. 16—(a) Series-type, single-bump, Bode equalizer. (b) Series-type, double-
bump, Bode equalizer. (c) Bump characteristic, H (f).

where a, is the insertion loss of the structure when R, = R,. Since
a, and x are constants, it is only necessary to consider p(f).

To realize a single-bump shape with the network of Fig. 16a,
Re |p(f)| is required to have the form

where H(f) is a bump-shaped function of frequency (Fig. 16c), and
p’(R.) is a function of a single resistor, independent of frequency.

Similarly, to realize a double-bump shape with the network of Fig.
16b, p(f) will have the form

where the subscripts a and b refer to two separate bump shapes. Both
H,(f) and H,(f) have the shape indicated in Fig. 16c. For the type of
designs discussed here, it is required that these shapes be in distinct
frequency bands. R, and R, provide independent control of the two
shapes.

One method of realizing eq. (5) is to use bridged-T sections as
shaping networks. To demonstrate this, a general expression for the
input reflection coefficient of a bridged-T network is required. Referring
to the schematic of Fig. 17b, which is an unconventional arrangement
of the circuit of Fig. 17a, the scattering matrix! of the network, with
Z1:Z2 = Rj, is given by

where a; and 5; refer to the incident and reflected voltage waves at the
ith port, respectively, and

The reflection coefficient p(f) = b:/a1, looking into port 1 of the
network, must now be determined. Ports 2 and 4 are terminated in
impedances whose reflection coefficients are pe(f) and p4(f), respec-
tively. Assume port 3 is terminated in R,, and as a result, p3(f) = 0.

Fig. 17—Bridged-T networks with ports 1 through 4 open-circuited. (a) Con-
ventional. (b) Unconventional.

dz = po(f)bz
as = pa(f)bs (7)
a3 = 0.
These port relationships, together with eq. (6), lead to
o(f) = Si(f)-p2(f) + Sta(f) -pa(f). (8)

where p.(R.) has been substituted for p.(f), since a resistive element
has replaced a frequency-dependent impedance. This has the form of
the reflection coefficient for a single-bump shape defined by eq. (4).
The single-bump network, based on the unconventional bridged-T
network of Fig. 17b, is shown in Fig. 18a.

Assume that the shaping network is such that Re [Si4(f) ] is a bump
shape centered at frequency f,. As a result, Sis 2 1 in the vicinity of
fa and s14 & 0 elsewhere. From eqs. (8) and (9)

The term Sis(f)-p4(Ra) represents a bump shape. As a result, half
of the desired equation, (5), has been realized. To realize the second
bump shape, p2(f) is made the input reflection coefficient of a similar
shaping network whose center frequency is f,, with fs << fa or fr > fa.
Networks having the assumed properties for Si, are shown for a two-
bump equalizer in Fig. 18b. For this network,

Equation (10) is now in the proper form to realize the defining
equation, (5), of the double-bump equalizer. The equivalence of the
two equations becomes apparent when the following substitutions are
made in eq. (10):

Fig. 18—Shaping network for adjustable bridged-T Bode equalizers. (a) Single
bump. (b) Double bump. (c) Triple bump.

To compensate for changes in cable loss relative to ground-tempera-
ture variations, regulating repeaters are placed in the Ld line approxi-
mately every 5 to 7 miles.” Relative to nominal cable loss at mean
ground temperature, both gain and loss compensation are required
according to whether the temperature is above or below mean tem-
perature. In either case, the shape, on a dB (log) basis, is proportional
to the square root of frequency. The networks that provide the compen-
sation, as in the L4 system,” are thermistor- and pilot-controlled,
wideband, square-root-of-frequency Bode networks. They differ from
the L4 designs, however, in that buffering amplifiers are included as
integral components of the networks, as shown in Fig. 19. Design

considerations, however, are basically the same and are not
reconsidered.

Three measurements are required to fully characterize the network:
(t) at high-loss setting, (77) at nominal setting, and (777) at low-loss
setting (see Fig. 20). The dynamic errors may be defined as

where An(f), Aw(f), and Azx(f) are loss measurements in the high-,
nominal-, and low-loss conditions, respectively, and Ky and Ky are
determined by requiring the error to be zero, by definition, at the pilot
frequency of 42.88 MHz. The static error may be defined as

Ar(f) is the expected loss in the nominal condition, which, because of
parasitics, gain shaping in the amplifiers, and interaction effects, is not
flat with frequency. Deviations from the ideal flat shape, however,
have the same effect as deviations from the nominal characteristic
of other components in the transmission path and may be corrected
by fixed and adjustable equalizers available in the system. Since
comparable dynamic equalizers are not available, more stringent
requirements are placed on the dynamic performance of the Bode
equalizer than on the flat or reset shape.

The necessity of requiring three loss measurements, corresponding
to three thermistor settings, to characterize network performance,
precludes the use of conventional methods for compensating parasitic
variations in the amplifiers and the network during manufacture. As
it is not possible to guarantee network performance from easily
measured amplifier parameters, the alignment procedure must correct
for both network and amplifier variations. Since the amplifier is not
adjustable, the network must provide tuning capabilities for both
shaping and amplifier compensation. Simple prescription of element

values or adjustment of resonances would not provide this capability.
Furthermore, an adjustment in a network component usually affects
all three shapes, making adjustment on a scanner too difficult. Thus, a
more sophisticated procedure is needed.

Since the amount of adjustment needed is small, but the allowed
deviations even smaller, linearization has proved to be valid. The
error after adjustment, ea(f), can therefore be expressed as:

where ey1r(f) is the initial error, Aysz(f) is the normalized deviation
caused by the jth component, and 2; is the amount of the deviation
used for adjustment. Similar relations hold for the nominal- and low-
loss errors. The composite error may be defined as the summed weighted
squares:

A weighting W less than unity allows a larger error in the nominal or
reset shape than in the dynamic errors ey or ez. Given an initially
determined set of errors for the three conditions, over an appropriate
frequency band, a set of parameter deviations 7; may be found that
_minimizes e?, using the so-called least-sum-of-squares procedure. This
procedure is well known and only requires the solution of a set of
simultaneous linear equations.

Unfortunately, in practice, the unconstrained solution often requires
an adjustment greater than the adjustable components can provide.
Thus, it was necessary to include a search for the best constrained
solution. The method chosen uses gradient information to determine
if a parameter must violate the constraints to lessen the summed-
squares error, e?. Other than being iterative, the method does not
require any information beyond that obtained from the least-sum-of-
squares procedure. While inclusion of the constraint algorithm con-
tributed to the success of the program in manufacture, a complete
description is omitted because of space limitations. It is necessary, of
course, to preset the adjustments to a prescribed condition, prior to
alignment, to correspond to bounds stored in the program.

Programming for the constrained least-sum-of-squares algorithm
was added to a streamlined measurement program for cotms. The
computer controls the measurement sequence and the alignment
procedure automatically. The only manual tasks required during
tuning are to switch the network into the different conditions and to
turn the slugs on the adjustable components. Measured, rather than

Fig. 21—Deviations in error with element adjustment in square-root-of-frequency
Bode network. (a) Li. (b) C4. (ce) L5. (d) L3.

computed, deviations from a typical network are used due to the
parasitic modeling problem. Because adjustments are effected on a loss
basis, it is neither necessary to count turns of the adjustable inductor
slugs nor necessary to measure inductance.

The deviations in the high-, nominal-, and low-loss shape errors,
caused by variations of four components, are shown in Fig. 21. The
curves for inductor L1, for example, show the deviation in the errors
corresponding to a 0.1-dB change at 70 MHz in the nominal- or flat-

loss-setting response. The result of the alignment algorithm is a
percentage of this deviation. For example, if the multiplier x; of eq.
(13) is determined by the least-sum-of-squares procedure to be 1.32,
multiplication of inductor L1 sensitivity (0.1 dB) by this number
results in 0.132 dB. Inductor L1 must, therefore, be adjusted for a
0.132-dB change in loss, at 70 MHz, to effect the optimum tuning of
that element. For inductor L3, the maximum deviations from flat
response occur at approximately 20 MHz over the 1- to 70-MHz
frequency range. This inductor is, therefore, tuned at that frequency.
Six adjustable components are included in the program. Only one
measurement, adjustment, and remeasurement iteration proved
necessary for the adjustment of the networks for optimum performance.
“Before” and ‘‘after’’ adjustment error curves are shown in Fig. 22.

Fig. 22—Network errors before and after adjustment. (a) Before adjustment. (b)
After adjustment. (c) Before adjustment. (d) After adjustment.

The computer-aided-adjustment program has proved its value
during the several years of manufacture of the regulating network.
Without corms, this approach would not be feasible. Without com-
puter-aided adjustment, the dynamic regulating objectives for the L5
system could not have been met.

In a system having as complex an equalization arrangement as L5,
the possibility of advantageous interactions among equalizers in the
complete hierarchy of equalization must be considered in any one
design. An example of such design considerations is the design of the
relatively simple, fixed-loss, deviation equalizer. This equalizer
compensates for the average error in the match of average-line-repeater
gain to nominal cable loss. Three deviation equalizers are installed
in each power-feed section.

Initial evaluation of the design led to the conclusion that require-
ments could be met with an equalizer consisting of two valley-shaped
and two bump-shaped bridged-T equalizer sections connected in
tandem, as shown in Fig. 23. It was necessary, however, that the
design be optimized so that the equalizer, in combination with the E1
and E2 adjustable networks, would provide optimum compensation.
This was achieved using the general-purpose optimization program
(@popr) tied together with an E1, E2 simulation program.”!

The E1, E2 program contains the measured response of all the bump
shapes included in the El and E2 equalizers. With approximately 15

Fig. 23—Deviation equalizer. (a) Valley-equalizer sections. (b) Bump-equalizer
sections.

percent of each El and E2 bump allotted to the deviation equalizer,
optimization resulted in an equalizer characteristic having deviations
from ideal placed at frequencies where the El and E2 adjustable
equalizers could best supply shaping to reduce the overall misalignment.

The E1, E2 simulator package simulated the iterative process that a
craftsperson would use in the field in making system equalization
adjustments. This yielded an effective design which has led to excellent
results on equalized lines.

Although essentially every transmission network for the L5 system
provided its own challenges in physical realization, the majority of
challenges related to compensation for parasitic inductance and
capacitance, to minimization of ground loops, and to reduction of
coupling between components. Environmental protection was provided
by drawn or fabricated metal enclosures with internally mounted
Pwp’s for supporting and interconnecting the individual circuit com-
ponents. External connections were, in most instances, made by use
of moisture-resistant plugs or jacks.

In several instances, however, substantially greater physical-design
effort was involved. Two examples of such design effort are: (z) the
design of the shaping networks of the line repeaters, and (77) the
physical design of the earth-ground filter.

The shaping networks of the basic, regulating, and equalizing
repeaters were specifically designed to reduce excess low-frequency
gain, to furnish surge protection, and to provide the power-separation
filtering requirements of the L5 repeaters. Figures 24a and b show the
schematics of the low-frequency A and B networks. The circuit
diagrams identify those portions of the networks concerned with the
different circuit functions. The network schematic of the bridged-T
equalizers used to obtain the desired insertion loss is shown in Fig. 24c.

The physical design of the shaping networks required considerable
interaction with system engineers. For overall efficiency and economy |
of space utilization, the surge-protection and power-separation
circuits were included in the shaping-network package. As indicated
in the photograph of Fig. 25, the two networks were placed side by
side in the lower section of the basic repeater housing. Requirements
on RF isolation, between input and output ports of the repeaters, led
to the use of cast-aluminum housings for packaging the shaping
networks. These housings, in turn, furnished the necessary mechanical

eS
FROM | SHAPING 70
EARTH | SURGE ! NETWORK PRE-
GROUND, (@ PROTEC- | AMPLIFIER
FILTER ,
(a)
POWER SEPARATION
Bio hte ee 5
|
|
J
BRIDGED 1
a OUTPUT
SHAPING
FROM
POWER NETWORK
AMPLIFIER
(b) (c)

Fig. 24—Low-frequency shaping networks. (a) Network A. (b) Network B. (c)
Bridged-T network schematic.

rigidity to meet the +0.025-dB tolerance on insertion-loss shape as
specified for manufacture.

The earth-ground filters of the line repeaters, shown in the photo-
graph of Fig. 25, with their coaxial jacks protruding from the repeater
housing, perform several important functions. Despite the seemingly
simple electrical circuit shown at the input and output ports of the
simplified block diagram of the basic repeater of Fig. 26, the earth-
ground filter must

(22) Maintain a low-impedance return path between the grounds
for all frequencies in the L5 band.

(izi) Provide a low-pass filter characteristic with a minimum of
120-dB isolation between repeater input and output (through
the ground return) for signals in the L5 band.

(tv) Carry the L5 line frequencies and the de powering current with
a minimum of amplitude distortion.

The physical design of the earth-ground filter” presented a design
challenge. As shown in Fig. 27, the filter contains two coaxial chokes in
the form of ferrite toroids and three high-voltage, high-reliability,
dual-dielectric capacitors packaged in a sealed enclosure. The filter,
minus the coaxial plug, is impregnated with polyisobutylene and
hermetically sealed to prevent the entrance of moisture or the leakage
of impregnant, which would contaminate the connectors.

To insure the high reliability of this design, considerable emphasis
was placed on the compatibility of materials, structural integrity,
sealing, and manufacturing control and testing.

The L5 system required the development of a wide variety of mag-
netic components numbering in excess of 100 different experimental
designs. This number includes original designs, subsequent improved
versions, and redesigns that reflected changing circuit concepts or
requirements. Ultimately, about 50 apparatus codes were issued to
specify these designs for manufacture. Many additions to existing
code series of magnetic components were also made available for L5
applications, but these are not discussed. The adjustable, air-core,
precision inductors specified for the manufacture of many of the trans-
mission networks developed for L5 are an example of magnetic com-
ponent designs based on existing code series.

Coded transformers and inductors appear in all segments of the L5
repeatered line. Frequently, these components are associated with
thin-film circuits and, as a result, miniaturized designs had to be made
available. Development of inductor structures for such applications,
however, predated the development of the L5 system, and, as a result,
emphasis is placed here on the development of various types of trans-
formers. Many transformer designs for the line repeaters required
pin-type terminals for interconnecting and mounting on PWB’s or
thin-film substrates. These transformers are discussed in Section 2.2,
Repeater Components.

/ FLANGE
/ SOLDER JOINT
Js
soe / FINGER STOCK
4 7s
ff /
ed FIRST ie ,SECOND CAPACITOR —_, THIRD CAPACITOR
/ ue // CAPACITOR / , i
/ 7
i // ,CORE TUBE/” 7 MINI-COAX Fa / KRAFT PAPER
7 7 fe / /

Magnetic components intended for terminal applications are fre-
quently used in complex arrays to develop hybrid trees arranged for
coaxial interconnection. The different methods used to establish an
interface between the magnetic components and the circuitry of the
system reflected not only on the problem of physical design, but on
the problem of electrical realization as well. Requirements on the
individual transformers constituting the hybrid trees had to be
tightened to meet overall system performance criteria. As a result,
these magnetic components had to be built out with resistors and
capacitors to form what are, in effect, special types of transmission
networks. Although this type of component is also used to a limited
extent in the regulating and equalizing repeaters, a representative
design is described in Section 2.3, Terminal Components. Finally, since
measurement limitations and techniques were important to all classes
of designs, these are discussed separately in Section 2.4.

The amplifiers in the basic and regulating line repeaters make
extensive use of bridge-type feedback,”? which offers the advantage
of making the uf loop independent of the line impedance Ordinarily,
the disadvantages of bridge-type feedback are the extreme impedance
levels and the power consumed in the bridge arms used to secure a
bridge balance. Both of these limitations, however, may be overcome
by use of hybrid transformers. Because of the very tight limits,
typically +0.02 dB, placed on reproducibility over the 1- to 70-MHz
passband, transmission-line design techniques** were extensively
adapted to the design of hybrid transformers. These designs specify
pairs, triplets, or quadruplets of insulated magnet wire twisted uni-
formly together and wound around a ferrite core. At low frequencies,
this arrangement behaves like a conventional transformer, but at high
frequencies, where parasitic elements predominate, the device behaves
like a transmission line. If the winding inductance and capacitance
have been properly proportioned, this approach results in greater
bandwidth than can be obtained from a conventional transformer
design.

In the case of a two-winding transmission-line transformer operating
between equal impedances, the characteristic impedance of the line,
Z. = VL/C, should be made equal to the terminating impedance.
Under these conditions, the transformer would theoretically have
infinite high-end bandwidth. However, the output hybrid of the basic-
repeater power amplifier was required to operate between an amplifier

impedance of 50 ohms and a line impedance of 75 ohms with the feed-
back-port impedance equal to 150 ohms. Since the transmission line
used to realize the hybrid could not simultaneously assume all these
different values, the best compromise was to make Z, equal to the geo-
metric mean of the output impedances, or 106 ohms. The ratios of the
terminating impedances also forced the use of a trifilar winding. The
impedance properties of such windings were studied for various wire
gauges, twist rates, insulation thicknesses, and materials. Ultimately, a
transmission line made from three strands of awe 40 polyurethane-
insulated wire, combined at a rate of 35 twists per inch, was selected.
Although other combinations would have produced the same Z,, the
high twist rate was used because it resulted in the best uniformity from
model to model. The schematic diagram and transmission charac-
teristics of the output hybrid are shown in Fig. 28. The divergence in
insertion-loss characteristics at high frequencies is the result of the
compromise value of impedance selection for Z,.

Although transmission-line transformers were preferred from a
component-design standpoint, they could not always be used because
their feedback-port impedance must be an integral multiple of both
the amplifier and line impedances. To overcome this constraint,
conventional layer-wound hybrid transformers were used at the input

Fig. 28—Schematic and insertion-loss characteristic of output hybrid for basic
repeater power amplifier.

Fig. 29—Transformer structure for conventional layer-wound hybrids using D-
cores.

and output of the basic-repeater preamplifier. The initial estimates of
the impedances both transformers would be required to match were
75 ohms at the line port, 60 ohms at the amplifier port, and a “low”
impedance at the feedback port. Various ratios were evaluated, and
the final ratio selected was 75:65 + 28 ohms, which results in a feed-
back-port impedance of 19.6 ohms.

To provide the best consistency in a layer-wound transformer, a
structure that includes a winding bobbin is preferred. Bandwidth
considerations, however, dictate that the structure would have to
approach the efficiency of a toroid having a winding wrapped directly
on the core and extending around its entire periphery. Core and
winding dimensions, therefore, are nearly coincident. The D-core
shown in Fig. 29 was chosen because the large area of the outer shell
produces a low magnetic reluctance. This means that the center-leg
length and area control the magnetic properties of the core. Since the
winding bobbin fits directly over the center leg, the winding dimensions
are only slightly larger than the effective core dimensions, and the
structural efficiency approaches that of the toroid.

Evaluation of amplifier models revealed that their high-frequency
gains were extremely sensitive to parasitics in the transformer wind-
ings. Requirements specified a +0.02-dB reproducibility from model to
model up to 70 MHz. Stringent winding procedures and in-process
checks were required to obtain this control in a conventional trans-
former. Wire gauges and pitches were selected to provide smooth,
even, single-layer windings. Carefully controlled paper insulation was
used between the most critical windings to insure that the parasitic
capacitance could be held to within +5 percent, which is three times
more stringent than normal manufacturing tolerances. Several different
types of winding machines were evaluated to obtain one capable of
providing the controlled-pitch and constant-tension features required
to maintain performance tolerances.

One of the more difficult designs required for terminal applications
was a 75:300-ohm, unbalanced-to-balanced transformer intended for
use in the Jmx modulators. The insertion loss of this transformer was
to be held flat to within +0.1 dB from 0.5 to 70 MHz, and the balance
of the center-tapped winding was to exceed 50 dB from 0.5 to 90 MHz.
Originally, separate designs were proposed for each jumbogroup
because no core material was available that would permit simultaneous
realization of these two requirements. In addition, an ability to main-
tain a balance of the required magnitude over this broad bandwidth
had not been demonstrated at these high frequencies. After carefully
studying the balance problem, however, it was felt that if an adequate
core material could be developed and the mounting structure rede-
signed, it would be possible to cover the entire range with a single
design by employing carefully positioned toroidal windings. The
balanced winding consists of seven turns of a bonded pair of wires
spaced evenly around the periphery of the core. These windings were
then connected in a series-aiding fashion to complete the center-
tapped 300-ohm winding. The 75-ohm winding was spaced between
adjacent bonded turns and all leads carefully dressed, resulting in a
sufficient degree of structural symmetry to maintain the balance.
Simultaneously, a new core having the necessary properties was
developed and the required mounting structure realized in a new
physical design.

Hybrid transformers, to provide combining and splitting functions
at a 75-ohm impedance level, were required for several different
applications in the L5 system such as Lpss-3, the E38 equalizer, line-
connecting arrangements, and smx circuits. Mounting and coaxial-

connector variations resulted in four distinct physical designs, as
shown in the photographs of Fig. 30. The basic electrical building
block consists of a transmission-line hybrid and an autotransformer
interconnected as shown in Fig. 30a. With reference to that schematic,
autotransformer Tl matches the 75-ohm input at jack J1 to the
37.5-ohm impedance at the center tap of hybrid T2. It is wound with a
twisted-pair wire whose characteristic impedance Z, = V75 X 37.5
= 53.0 ohms to optimize performance. Since the impedances to be
matched differ by 1:2, the tap must be placed at a position correspond-
ing to 1:v2. This means that the windings cannot consist entirely of
twisted-wire transmission line, but must include some free turns.
These are created by partially decomposing the twisted bundle, with
best results obtained by minimizing the number of free turns.

The hybrid T2 has a 2:1 turns ratio and is wound with a twisted
pair having a characteristic impedance of 75 ohms. The network
terminating resistance R has a value slightly greater than the ideal
150 ohms to account for transformer core losses, and the capacitor C
enhances the high-frequency response of the device. The electrical
performance achieved by these components over the 1- to 70-MHz
frequency range is as follows: (2) the transmission loss from input to
either output is 3.10 + 0.05 dB, (72) the trans-hybrid loss between
conjugate output ports is greater than 30 dB, and (777) the return loss
at any port is greater than 26 dB. In addition, the two output ports
of any individual hybrid track to within 0.02-dB loss and 0.5-degree
phase shift. An ideal hybrid would have 3.01-dB transmission loss,
infinite trans-hybrid and return losses, and perfect flatness and track-
ing, with both outputs exactly in phase with each other and the input.

In addition to their use as individual hybrids, particular pairs of
these transformers may be grouped to form large n-port arrays or
“‘trees.”” The transformers in Fig. 30b use standard Bell System
coaxial plugs and jacks so arranged that they may be interconnected
directly. The transformers in Fig. 30c use miniature coaxial connec-
tors and may be formed into trees by cabling them together. To reduce
the number of interconnections required, the hybrid at the right of
Fig. 30c, a dual hybrid, has an input, one —3-dB output, and two
—6-dB outputs. The hybrid at the left of Fig. 30c is a single hybrid
of the type shown in Fig. 30a.

The +0.02-dB reproducibility of insertion loss required for many
L5 magnetic-component designs could not be guaranteed initially by
direct measurements because neither the test set nor the apparatus

SINGLE HYBRID SINGLE HYBRID
(THREE-JACK DESIGN) (b) (TWO-JACK, ONE~PLUG DESIGN)

Fig. 30—Hybrid transformers for L5 terminals. (a) Building-block schematic.
(b) Hybrids with Bell System coaxial connectors. (c) Hybrids with miniature coaxial
connectors.

under test exhibited a return loss compatible with this extremely
tight limit. Instead, a system was used in which a set of transformers
known to have the proper characteristics in an amplifier was used to
calibrate a test fixture. The calibrated fixture was then used to measure
product. To guard against changes in the fixture or measurement
system, however, these measurements were correlated to those made
on reference transformers whose histories were well known. When
either fixtures or reference units needed replacement, the appropriate
steps in the calibration and correlation processes were repeated.
Meanwhile, a program to provide improved measurement capability
for these devices was begun.

To have a reproducible base line for loss measurements on coTms,?
a simple strap between the transformer input and output ports of the
test fixture was used. While this introduces reflections because these
ports often have different impedances, it is the simplest and most
consistent method of eliminating differences in test sets, connectors,
and cables. Furthermore, with sufficient padding close to the corms
ports of the fixture, satisfactory base-line reproducibility can be
obtained. Accordingly, alternate measurement schemes were discarded
in favor of upgrading the existing approach.

The test fixtures were originally constructed with ordinary Pwp’s.
The contacts to the pin-type transformer terminals were made with
miniature spring sockets, and the interconnection to the test set used
Bell System coaxial connectors. The resistive pads used to match the
transformer impedances to the coTms ports were made with various
types of discrete resistors for the different transformer codes. It was
recognized very early that the required reproducibility of measure-
ments would be difficult to maintain in a production environment with
this type of test fixture. Coaxial connectors can introduce impedance
discontinuities, and spring sockets can become contaminated or wear
out with frequent use. When replaced, resistive pads are often damaged
by the heat of soldering and change value. In addition, the associated
parasitic electrical element values are a function of exact mechanical
configuration and placement which are difficult to control.

Improved designs of the spring contacts and of the coaxial connectors
were introduced to improve the performance of the test fixtures.
Knife-edge contacts with force-free insertion were used to replace the
former, while the latter were superseded by precision 50-ohm versions.
As the next step, the PwB was replaced by an alumina substrate
having controlled dielectric properties. Tests quickly indicated new
problems. Because of the higher dielectric constant of the substrate,

the effective electrical length of the connections increased significantly,
causing difficulty with crosstalk and reflections. The chip resistors,
which were chosen for the matching pads, changed value on soldering
and caused reproducibility and rework difficulties. A final design
evolved in which the path lengths of the microstrip connections on
the substrate were reduced to a minimum and the resistors realized
as thin-film elements. This arrangement permits high accuracy in the
adjustment of resistance values, makes parasitic elements closely
reproducible, and eliminates the repair operations that had been
required with the original fixture design. External connection problems
were further reduced by coupling the coaxial connectors to the substrate
via precision spring-loaded microstrip launchers. These new fixtures,
shown in the exploded view of Fig. 31, have attained a reproducibility
of better than +0.01 dB.

Transmission networks and magnetic components are among the
least visible components of the L5 system. Despite this lack of visi-
bility, these components serve vital systems functions. Many of these
functions are reviewed in this paper. The principal objectives of the
paper, however, are to record those state-of-the-art advances that
have contributed to the success of the L5 system, and to identify those
allied technologies that have influenced network and magnetic com-
ponent development. The influence of the computer in analysis and
synthesis, precision measurement, and optimization is stressed, and the
added influence of development in the piezoelectric device and mag-
netic materials areas has been noted. In many instances, developments
in these allied technologies predated the L5 system and resulted from
a general philosophy of ‘‘tool building” initiated, specifically, to have
these technologies keep pace with future systems and component-
development needs. Without such a philosophy, L5 requirements
could not have been met on schedule.

Many individuals, too numerous to mention, have made significant
contributions to the development of transmission networks and
magnetic components for the L5 system. These contributions have
come from individuals in the authors’ own departments and from
co-workers at other Bell Laboratories locations. Special thanks,
however, must be given to R. L. Adams and G. J. Mandeville for
contributing to the sections on computer-aided tuning of networks
and the design of multibump, adjustable Bode equalizers, respectively.

2. G. Szentirmai, ‘‘A Filter Synthesis Program,” in System Analysis by Digital
Computer, ed. F. F. Kuo and J. F. Kaiser, New York: J. Wiley and Sons,
1966, pp. 130-174.

3. W. J. Geldart, G. D. Haynie, and R. G. Schleich, ‘‘A 50-Hz to 250-MHz Com-
puter Operated Transmission Measuring Set,” B.S.T.J., 48, No. 5 (May-June
1969), pp. 1839-1381.

. R. A. Sykes, W. L. Smith, and W. J. Spencer, “Monolithic Crystal Filters,”
IEEE International Convention Record, 1967, pp. 79-93.

. W. G. Alberts, J. B. Evans, T. J. Haley, T. B. Merrick, and T. H. Simmonds, Jr.,
“L4 System: Terminal Arrangements,” B.S.T.J., 48, No. 4 (April 1969), pp.
1024-1025.

. J. L. Garrison, A. N. Georgiades, and H. A. Simpson, ‘‘The Application of Mono-
lithic Crystal Filters to Frequency Selective Networks,” Digest of Technical
Papers, 1970 International Symposium on Circuit Theory, 1970, pp. 177-178.

7. H. A. Simpson, E. D. Finch, Jr., and R. K. Weeman, ‘‘Composite Filter Struc-
tures Incorporating MCFs and LC Networks,’ Proceedings of the 25th
Annual Symposium on Frequency Control, Fort Monmouth, N.J.: U. 8.
Army Electronics Laboratories, 1971, pp. 287-290.

. P. Lloyd, ‘‘Monolithic Crystal Filters for Frequency Division Multiplex,” ibid.,
pp. 280-286.

9. M. Dishal, ‘“Two New Equations for the Design of Filters,’’ Electrical Communi-
cation, 30, December 1952, pp. 324-337.

10. A. I. Zverev, Handbook of Filter Synthesis, New York: J. Wiley and Sons, 1967,
Chapter 6.

11. “Filters, Modern Network Theory Design,’’ Reference Data for Radio Engineers,
ed. H. P. Westman, Fifth Edition, ITT, 1968, Chapter 8.

12. J. L. Garrison and A. N. Georgiades, ““Band-Elimination Filters,’’ U. S. Patent
3,704,433, November 28, 1972.

13. A. A. Comparini, ‘‘Analysis and Synthesis of a Band-elimination Filter Using
Monolithic Crystal Filters,’’ unpublished memorandum.

14. L. Brier, “Der Entwurf von HF-Bandfiltern und Mechanischen Filtern mit
Dampfungspolen nach dem Betriebsparameterverfahren,’’ Guest Lecture at
oe International Science Colloquium, University of Ilnienan, September

16. F. C. Kelcourse, W. G. Scheerer, and R. J. Wirtz, “L4 System: Equalizing and
Main Station Repeaters,’ B.S.T.J., 48, No. 4 (April 1969), pp. 907-911.

17. E. H. Angell, Y.-S. Cho, K. P. Kretsch, and M. M. Luniewicz, “L5 System:
Repeatered Line,” B.S.T.J., this issue, pp. 1935-1985.

18. W. R. Lundry, “Attenuation and Delay Equalizers for Coaxial Lines,’”’ Transac-
tions of AIEE, 68, 1949, pp. 1174-1179.

19. H. J. Carlin, ‘‘The Scattering Matrix in Network Theory,” IRE Transactions on
Circuit Theory, CT-3, No. 2 (June 1956), pp. 88-97.

20. J. L. Garrison, L. P. Labbe, and C. C. Roch, “Basic and Regulating Repeaters,”
B.S.T.J., 48, No. 4 (April 1969), pp. 871-876.

21. R. M.-M. Chen, C. F. Hempstead, Y. L. Kuo, M. L. Liou, R. P. Snicer, and
KE. D. Walsh, “L5 System: Role of Computing and Precision Measure-
ments,” B.S.T.J., this issue, pp. 2249-2267.

22, R. A. Thatch and F. D. Waldhauer, ‘High Frequency Ground Isolation Filter
for ine Powered Repeater Circuits,’’ U.S. Patent No. 3,530,393, September
22, 1970.

23. H. W. Bode, Network Analysis and Feedback Amplifier Design, Princeton, N.J.:
D. Van Nostrand Company, 1945, Chapter III, pp. 37-38.

25. R. K. Bates and D. J. Zorn, “L5 System: Signal Administration and Intercon-
nection,” B.S.T.J., this issue, pp. 2129-2145.

The L& Coaxial-Carrier Transmission System 1s the first long-haul,
high-capacity transmission system for which the design was strongly
influenced by extensive application of computer-aided design (cap)
techniques. A tight time schedule required a parallel effort of (1) improving
and using the somewhat limited capabilities of existing caD programs
and (12) developing new programs having increased capability and effi-
ciency. Similar development of computer-controlled measurement tech-
niques provided necessary device and component characterization and
subsystem evaluation. The result is a powerful group of tools that are
independently important, but whose combined use helped make possible
the temely completion of the Ld system design. These tools, now ‘proved
tn,” will profoundly tnfluence the next system design philosophy.

At the beginning of the L5 project in 1968, the set of computer-
aided design (cAD) tools and the specialized computer-operated trans-
mission measurement set (corms) at the Merrimack Valley location of
Bell Laboratories were in an early stage of evolution. The tight system
development schedule prevented the thorough, leisurely development
of sophisticated programs that could provide the complete analysis and
characterization needed for high confidence in design. Rather, a
parallel effort was undertaken to augment existing design aids and to
develop improved ones. The result was immediate answers generated
inefficiently for early designs and a powerful group of programs that

have been ‘‘proved in” by use during the later stages of L5 develop-
ment. These were, however, not completed early enough in the design
cycle to allow their full impact to be exerted on the overall design
process in the manner that is now possible.

The following sections illustrate typical uses of computer programs
for seven phases of the design sequence. These include:

(¢) Small signal ac analysis of circuits having many nodes, in-
cluding sensitivity and tolerance analyses.
(iz) Optimization of element values in a circuit.
(177) Statistical measurements of early production models of re-
peaters for use in equalizer design.
(tv) Nonlinear distortion analysis.
(v) Component characterization.
(vt) Overall system analysis.
(vit) Equalizer characterization and alignment during manufac-
turing.

Limitations as well as successes are included to illustrate why and
how the evolution of programs took place.

Several general-purpose circuit analysis programs,! as well as
optimization? and Monte Carlo* programs, were in wide use in the
communications industry by 1968. However, none was able to handle
a circuit of the size (over 80 nodes) and complexity of the L5 repeater.
While the next generation programs were being developed, modifica-
tion of the existing programs and segmentation of the circuit were
necessary to solve the immediate circuit analysis problems.

Since the L5 system uses amplitude modulation, the repeater must
be a very linear device; this places strong emphasis on accurate,
small-signal ac analysis of all circuits in the signal path. The basic
repeater amplifier alone contains 80 nodes, 111 passive components,
8 transistors, and 4 wideband transformers. These numbers do not
include “‘parasitic’’ elements found to be essential for accurate modeling
so that, in practice, the actual node and component count must be
significantly increased.

To overcome the size problem, the circuit was at first partitioned
into blocks that could be analyzed with existing programs. One output
of each of these analyses was a multiport admittance matrix charac-
terizing the block at those nodes where it interconnects with other
circuit blocks. These block representations were stored in a library for
subsequent retrieval when the performance of a multiblock circuit was
to be calculated. This approach of using a frequency-dependent matrix
to represent subcircuits proved useful, not only for reducing node and
element count, but also for allowing the direct use of measured data
from cotms to characterize components not easily or accurately
modeled by lumped element values.

The circuit segmentation approach also aided the calculation of
sensitivities to element values. As each element in a given block was
varied to determine overall circuit sensitivity to that element, the
matrix representations of other blocks remained unchanged, saving
much calculation time for the overall problem. Sensitivity data proved
useful in many ways. Since a full-scale Monte Carlo analysis of the
circuit was impossible during early development, a linear sensitivity
model was used to set component tolerances based on their predicted
effects on frequency response. The sensitivity data on parasitic element
values also identified critical areas of the circuit where careful modeling
and control were essential. It further identified those elements to be
subjected to optimization and smaller-scale Monte Carlo analysis.

The segmentation approach also permitted use of the early optimiza-
tion and Monte Carlo programs.‘ These were particularly helpful in
the design of the feedback networks which, though small compared
to the overall repeater, were critical in determining the frequency
shaping of the repeater.

Although the “divide and conquer” method did allow some success
in the early stages of analysis, several difficulties became apparent
that were corrected in the next generation programs. The many steps
involved in the piecemeal approach led to analysis turnaround times
of the order of one day for a complete circuit. This severely limited the
number and type of ‘‘what if’? questions that could be raised and an-
swered. Not only was the overall turnaround time relatively long, but
all the segments had to be run in a batch mode, as no interactive
facilities were available for the designer to pose such ‘‘what if’’ ques-
tions quickly or effectively. These inefficiencies, coupled with the non-
static nature of the actual circuit as various engineers wanted to

implement changes suggested by earlier analysis steps, made the
turnaround time a serious problem. The multiple passes through
various analysis segments also led to interfacing problems and in-
creased probability of coding errors.

The program developed to overcome most previous limitations is
called circuit analysis program for efficient computer-optimized design
(cAPE-coD). It: achieves its power and efficiency through a sparse
matrix technique and special, machine-generated coding. It has
analyzed circuits with 120 nodes, provides a wide variety of engineering-
oriented outputs, allows simple problem formulation by engineers, and
provides interactive capability on a graphics terminal.

The formulation technique chosen for cAPE-cop uses the tableau®
approach. This method of arranging the system of equations that
defines the topology and behavior of a circuit not only yields an
efficient analysis scheme, but also gives flexibility and simplicity to the
program design while minimizing the restrictions placed on the circuits
that can be analyzed. For example, the following types of elements
that are seldom allowed in ac analysis programs are allowed in CAPE-
cop: controlled voltage and current sources; zero-valued resistors,
capacitors, and inductors; and devices that have no Y or Z matrix
representations, but have an S matrix representation (e.g., an ideal
transformer or a three-port ideal circulator). By inserting zero-valued
components into the original circuit where one suspects parasitics to
be important, the program can then systematically investigate their
effects without recoding, simply by varying the values from zero on
subsequent runs.

The elimination order for the tableau can produce, at one extreme,
essentially a nodal formulation; at the other extreme, a mesh formula-
tion; in general, a hybrid formulation results. This flexibility gives the
tableau method the capability of producing more accurate results than
other methods. Such accuracy improvements are important when an
element or subcircuit has a Y or Z matrix representation that is ill-
conditioned, e.g., the Y matrix of a very small resistance.

To minimize core memory requirements of the computer and to
allow analysis of large circuits having over 100 nodes, sparse matrix
reduction techniques are used throughout the program. For efficiency
in execution time, a loopless machine code program to solve the
tableau system of equations is generated first for each problem. This
efficiency is most important when the analysis package is used as a

subroutine in an iterative program such as optimization or tolerance
analysis, since the time spent generating the machine code is insignifi-
cant compared to the time for repeated analyses. In addition to this
machine code, dynamic memory storage allocation and memory
paging schemes are used to increase efficiency further.

Input data for cAPE-cop can be entered in completely free format,
which not only provides great convenience for the user, but also reduces
the number of coding errors in large circuits. The input “language”
was developed to be natural for an engineer, easily understood and
remembered. The caPE-cop currently accepts the following circuit
elements: resistors, capacitors, inductors, controlled or independent
voltage or current sources, two forms of transmission lines, two models
for transistors (one of which is used for distortion analysis, described
later), and N-port black-box devices. The latter can be characterized
by directly measured Y, Z, or S parameter data, or by a user-supplied
subroutine that calculates these parameters. In addition to having
fixed values, elements can be optionally defined as a tabular function
of frequency. This black-box capability was essential to the L5 repeater
analysis, since no sufficiently accurate lumped element model existed
to represent the wideband transformers used in the forward and feed-
back paths. Instead, two-, three-, and four-port matrix representations
were formulated from frequency-dependent measured data

Perhaps the most important improvement over previously used
programs which caPH-cop provides is the wide variety and flexibility
in its computed outputs. These include any branch voltage or current,
any node voltage, or any of the following transmission related quanti-
ties: insertion gain or loss; voltage, current, or power gain or loss;
return loss or reflection coefficient; load impedance or admittance;
driving point impedance or admittance; u8 gain and yw impedances;
and Y, Z, or S parameter matrix values.

The 8 gain, i.e., the loop gain of a feedback amplifier, is especially
useful for verifying stability in the amplifier. Since many of these
outputs are complex numbers, they can be listed in any of the following
forms: real and imaginary parts, magnitude and angle (in degrees or
radians), dB[ = 20 logio(magnitude) |] and envelope delay (the deriva-
tive of the angle with respect to angular frequency).

Sensitivity studies are facilitated by another feature of CAPE-cOD.
After an analysis with nominal circuit values has been completed, it
is possible to modify the values of any (one or several) element, sub-
ject to constraints if desired. A second analysis is then done, and the
output quantities can be the difference in any normal output that results

from the modification of the circuit element values. This provides a
direct measure of circuit sensitivity to any change, large or small, in
any element values.

For a system as complex as L5, no exact design procedure exists at
any level, either device, circuit, or system. Many iterations are required
from an initial to a ‘‘final’”’ design. Various tradeoffs are weighed, merit
criteria are assigned, and design parameters are adjusted to yield a
result that is optimum under given assumptions.

The crop is capable of minimizing a criterion function F(x) by
adjusting the set of n parameters 21, %2, --+, %n (or in vector notation
x) which may be bounded (above and/or below), unbounded, or fixed.
A simplified program flow chart is shown in Fig. 1. Built into the
program is a least-pth error criterion function that has the following
form:

yij(X) = y;(X, a:) is the jth response function value evaluated for
the ith independent variable a;. For example, y;, y2, and
yz; may be loss, delay, and input impedance of a given
circuit, while x represents the circuit parameters and a; is
the ith frequency point.
rij is the requirement of the jth response function for the zth
independent variable. If a range of requirements is re-
quested instead of single value, and if y,;(x) falls outside
the range, the r;; in the formula is the value closer to y:;(x).
p is any positive even integer.
W:; is the weighting constant of the jth response function at
the ith requirement point.

INCREMENT COUNTER
APPLY STRATEGY TO
FIND BETTER SET Xi
CALL USER’S SUBROUTINE
TO EVALUATE F(Xi)

If the least-pth error criterion is not adequate for an application,
the apor program also accepts a user-defined criterion function in the
form of a subroutine. A user-supplied subroutine must be given to
evaluate y:;(x) for the above error criterion. This subroutine is often
in the form of a complete ac circuit analysis done by another program,
such as CAPE-copD, but now treated as a subroutine.

A variety of strategies or algorithms exists for finding the optimum.
Since the efficiency of these strategies is problem-dependent, it is
desirable to have many available from which the user (or program
itself) can choose. The @roP is equipped with Fletcher-Powell,® steepest
descent,’ Nelder-Mead,° least-pth approximation® (least-squares when
p = 2), and many others. Stopping the search or switching to another
algorithm may be done either after a given number of iterations* or
after the percent improvement is less than a specified value for five
consecutive iterations.

Since the above algorithms work only for unconstrained optimization
problems, the optimization of a bounded parameter problem (which is

*In Gpop, an iteration corresponds to an improvement; i.e., in one iteration, the
algorithm searches until a better criterion value is found.

a simple form of constraint) is accomplished by transforming the
bounded parameters into free parameters. A variety of transforma-
tions have been built into the program. A user may also supply a
specific transformation for a particular application. The gradient of the
response function can be computed either from a numerical difference
approximation or from a user-supplied expression. Input and/or output
subroutines for pre- and/or postoptimization computation add another
useful capability. In some problems, it is necessary to apply parameter
constraints (such as interrelationships between parameters) inde-
pendently of any parameter bounds. The input or output routines in
GpoP allow the user to apply sequential unconstrained optimization
techniques to constrained optimization problems.”

The application of a@popr to a system design problem is given in a
later section; here, a circuit problem, the optimization of a fixed
deviation equalizer, is used for illustration. In practice, the fixed
equalizer applies an average correction for the difference between
repeater gain shape and cable loss shape."'? Residual errors are then
corrected in the field, using adjustable E1-E2 equalizers. The criterion
function does not (and should not) consider the error at the output
of the fixed deviation equalizer, but rather the final error after all

OVERALL EQUALIZATION COMPUTE ERROR
REQUIREMENT
APPLY E1
DEVIATION EQUALIZER ZERO-—FORCING DESIGN
INITIAL DESIGN
COMPUTE MEAN SQUARE
ERROR AND ADJUST E1
COMPUTE ERROR
CHANGE DEVIATION APPLY E2
EQUALIZER PARAMETERS ZERO-—FORCING DESIGN
COMPUTE TOTAL ERROR COMPUTE MEAN SQUARE
TRANSFER TO GPOP ERROR AND ADJUST E2

Fig. 3—Performance of fixed deviation equalizer before and after optimization.

three equalizations have been applied. Thus, the program must
simulate the field adjustment process before it can optimize the
parameters of the fixed equalizer. Figure 2 shows the flow chart for
this process, and Fig. 3 shows the result of optimization. Four stages
of bridged-T equalizers, having 32 elements, were needed in the final
design for the excellent match to requirements.

IV. STATISTICAL TRANSMISSION MEASUREMENTS OF REPEATERS
FOR EQUALIZER DESIGN

The equalization hierarchy of the L5 system defined two levels of
accuracy needed for transmission measurements on subsystems such
as the basic repeater, regulating repeater, and equalizing repeater.
These levels must be kept in mind to appreciate the measurement
problems that had to be solved. First, the basic repeater provides a gain
shape to compensate for cable loss (proportional to Vf), going from
4.97 to 32.04 dB over the range from 1.6 to 66 MHz. The manufacturing
goal was that basic repeater gains would not deviate from target shape
by more than +0.1 dB at the two-sigma points of the statistical dis-
tribution. Thus, a transmission measurement accuracy of about 0.01
dB at each frequency was needed. The wide dynamic range (30 dB)
and frequency range (nearly two decades) made this 0.01-dB target
difficult to achieve. Since measurements could be made on only a
limited number of basic repeaters, but the mean shape had to be
extrapolated to apply to many repeaters in the system, the accuracy
of individual measurements should, if possible, exceed the 0.01-dB

target. Only a test set such as the cotms was able to provide this
capability with the speed needed for a thorough job. Typical runs
included 100 discrete frequencies across the band and took one minute
per run. .

The second level of measurement accuracy was needed for the re-
maining steps in the hierarchy, the regulating and equalizing repeaters.
Since these provide mainly second- or third-order corrections, the
dynamic ranges to be covered were relatively small, but accuracies
needed approached 0.001 dB over the same wide frequency range.
Nothing can be above suspicion at the 0.001-dB level (i.e., one part
in 10). Mistermination or mismatch errors are the most frequent and
insidious problem. Over this frequency range, even the best available
pads needed to mask the test set impedances from those of the unknown
showed significant frequency shaping of both transmission and im-
pedance characteristics. Inexpensive mass-produced connectors, as
used in field installations, have poor connection repeatability and im-
pedance in terms of these accuracy requirements. But the temptation
is strong for the infrequent user to assume that an instrumental
resolution of 0.001 dB will guarantee him an absolute accuracy of
0.001 dB in any measurement. So regular accuracy verification of the
test equipment, use of selected highest quality test fixtures and masking
pads, and careful error analysis of each measurement was essential ;
when these checks were omitted, serious errors were often discovered
later.

A third complication arose from the need for detailed temperature
characterization of the repeaters. The maximum temperature coefficient
of 0.0016 dB per °F for the basic repeater is a value critical in the design
of the dynamic equalizer and the regulating repeater. A computer-
controlled environmental test chamber was used with cotms to ac-
cumulate sufficient data to determine this. Continuous control and
automatic data logging minimized the operator interaction, but round-
the-clock measurements required months to complete. Long intercon-
necting cables and a network of multiplexing relays introduced serious
line reflections, requiring special care in selecting and locating masking
pads. When the measurements were self-consistent, a standard sta-
tistical analysis program in another computer completed the tempera-
ture characterization.

Nonlinear distortion is a serious problem for a long-haul, frequency-
division-multiplexed, analog communications system using solid-state

devices. Most serious are the second- and third-order intermodulation
distortions, caused mainly by the transistors. Allowable distortions
must be far below fundamental signals in practice; for example, the
third-order distortion should not exceed a level 90 dB below funda-
mental. Calculation of such small distortions requires not only very
accurate transistor modeling, but also appropriate mathematical tools
to extract such a small fraction of the main signal.

Circuits having only a few transistors connected in cascade and a
linear feedback path have been analyzed by 8. Narayanan," using the
Volterra series approach. While answers are obtained in closed form,
the computer implementation is not general and cannot handle the
large complex circuits in an L5 repeater. So a new program was written
to treat the second- and third-order distortions of transistor circuits
under small signal conditions. It is called the nonlinear distortion
analysis program (NODAP) and is based on the perturbation method.

For this type of problem, successful analysis depends on an accurate
active device model. The integral charge control model (icm) of
Gummel and Poon, which contains many nonlinear effects not in-
cluded in conventional models, was chosen. Some more important
effects include conductivity modulation, base push out, Early voltage,
and avalanche effects. Expanding the 1cm equations around a given
bias point and expressing terminal currents in terms of junction
voltages leads to an appropriate small-signal model. This is readily
separated into linear and nonlinear parts, the linear part being repre-
sentable by a hybrid-7 equivalent circuit, or simply a two-port ad-
mittance matrix Y(jw) (Fig. 4). Two controlled current sources, ty.
and iy., represent the nonlinear part. These together represent the
intrinsic transistor, to which must be added appropriate parasitic
elements; the parasitics are treated linearly except for the inactive
collector capacitance.

The perturbation method works well (two iterations are sufficient),
since the nonlinearities are small for small-signal conditions. The
computational algorithm!® uses any linear circuit analysis program
(in this case, CAPE-coD). The result is an accurate, efficient, general-
purpose program. The only nonlinear elements in the circuit are iy.
and iy-, and they can be shown to be equivalent to two distortion cur-
rent generators having amplitudes and phases determined by the
second- and third-order nonlinear coefficients and the linear charac-
teristics of the transistor circuit. This approach for a single transistor

can be readily extended to multitransistor circuits, since the distortion
computation is carried out by a quasilinear approach and superposition
holds.

The sequence of calculations in NopAP is interesting, showing how
major programs can be combined. Given the circuit topology with the
IcM parameters, bias point of each transistor, and sets of three funda-
mental frequencies (fi, fe, and f;), NoDAP computes the linear and non-
linear coefficients for the transistor model. The linear characteristics
are passed to cAPE-cop to calculate output voltages, load admittance,
and junction voltages. These are returned to NopaP to calculate the
second-order intermodulation current sources at fi + fo, fi + fs, and
fe + fs. These sources go back to caPE-cop to calculate second-order
distortion voltages. This type of looping is similarly repeated for the
third-order distortion. The final step, by NopapP, prints out the cal-
culated second- and third-order distortion indices Merz and Mz.

A “linear” and “nonlinear” option in NopaP allows the “‘turning off”’
or “turning on” of the distortion sources in any stage in an amplifier,
to determine the contributions from that stage. The distortion effects
of various nonlinearities can thus be isolated, giving insight impossible
from direct measurements alone. Also, distortion vectors are computed,
allowing the design of distortion cancellation circuits.

The calculated results for several L5 circuits have been in very good
agreement with measured data, within 2 dB over a wide range of bias

conditions and frequencies. Figure 5 shows a comparison of predictions
and measurements over the L5 frequency band.

Accurate device modeling is essential to the success of NODAP in
predicting nonlinear distortion. A program called extraction of parame-
ters for the Icm of transistors (EPIC) has been developed to extract 30
IcM parameters from three sets of measured data. The raw data include
the de V-I characteristics, the junction capacitance characteristics,
and the small-signal ac characteristics measured over a wide range of
bias conditions. The previously described Gpor program is used to
obtain the best Icm parameters by optimizing the match to the data.
Those parameters associated with a single data set are matched first,

progressing to parameters that affect multiple data sets. This gives
accurate results (physically realistic) quickly.

The main L5 transistor parameters were carefully extracted, using
Epic. In addition to their usefulness in predicting distortion, the linear
model parameters were used in cAPE-cop for normal ac circuit analysis,
giving excellent results. Agreement between computed and measured
S parameters for the transistor was within 2 dB over the entire Ld
frequency range.

Almost all the electronic subassemblies are built with components
that are treated as classical “lumped elements” such as resistors,
capacitors, inductors, or diodes, or as multiport devices such as trans-
formers or transistors. Near the top L5 frequencies (66 MHz), the
lumped element approximation is often inadequate and transmission
line delays become important. Small capacitances and inductances,
normally considered ‘“‘parasitics,”” may have substantial effects in a
circuit, especially over the extreme bandwidths covered by feedback
amplifiers.

Simple passive components, such as two-terminal impedances, were
measured automatically over frequency ranges up to 1000 MHz. Not
only was the variation of element value with frequency obtained
(caused by inadequate models), but derived properties were also ob-
tained, such as the permeability and loss of magnetic materials, ‘‘Q”
variation with frequency, self-resonant frequencies of reactive com-
ponents, and linear two-element equivalent circuits of diodes. Such
measurements were made quickly and very accurately on the coTms,
using an impedance calculating program. Impedances from milliohms
to megohms can be measured over the range from audio to microwaves.
Accuracies of the order of 0.1 percent are realized for real and im-
aginary components between 1 and 1000 ohms. A recent addition to
the measurement tools is a computer-operated impedance bridge
(cozy) that provides even greater resolution (approaching 10 ppm)
and impedance range, though at frequencies limited to below 30 MHz.

The characterization of transformers and transistors as multiport
devices was at a more sophisticated level. Basic measurements of
voltage scattering (S) parameters were made automatically on coTMs,
using a special program and calibration technique which compensates
for the mismatch errors caused by imperfect test set impedances.”
These corrections are especially important since test set impedances
do not remain constant over wide frequency ranges. From the basic S

parameters, other matrix parameters such as H or Y were obtained
by computer data transformation. The H parameters are particularly
useful for transistors, while the Y parameters fit naturally into circuit
analysis programs based on nodal techniques. The multiport trans-
formers used in the L5 amplifiers were characterized by merging several
sets of two-port measurements into a 4-by-4 Y matrix representation.

Many of these characterization jobs were not carried out until
inexplicable problems appeared during the development of circuits.
However, with the presently available tools that are quick, accurate,
and easy to use, all critical component types can be characterized
automatically before use in model circuits. Such routine checks can
prevent many problems and save considerable time in the long run.
A component as simple as a resistor has many nanohenries of body and
lead inductance that can play havoc in high-frequency circuits, yet
it takes only a few minutes to characterize the component with modern
tools.

In repeatered analog transmission system design, the optimal al-
location of total system noise caused by various sources and subject
to certain constraints is an important but tedious task, if the con-
ventional approach is used. Manual trial-and-error modification of
transmission levels and calculation of the resulting noise at various
system points are time-consuming processes. All these are interde-
pendent functions of repeater gain, noise figure, modulation coefficients,
etc. A nonoptimum assignment may lead to failure to meet system
design objectives and to possible redesign of basic elements, starting
a new system design cycle. Since the process requires optimizing a
performance function subject to a given set of system constraints,
modern optimization techniques are applicable.

An intermodulation noise computation program called NEWwMoD was
written at Bell Laboratories in the 1960’s for the development of sub-
marine cable systems. This program employs a modified version of
Bennett’s product count method!® in a formulation that takes into
account the effect of the nonlinear phase characteristic of repeaters.
The NEWMOD program was used extensively in the development of the
L5 system, in which the introduction of phase shaping networks effec-
tively randomized the addition of third-order intermodulation prod-

ucts. This allowed the channel capacity of the system to be increased
by about 20 percent for the same noise performance.

The product count method requires too much computer storage and
processing time to allow the coupling of Newmop with erop for system
noise optimization. This difficulty can be avoided by modifying the
earlier formulation so that a high-order Chebyshev numerical-integra-
tion formula is employed to replace the product count approach. Based
on this new formulation, a program named noise optimization for
repeatered analog transmission systems (NORATS) has been imple-
mented and is briefly described.

The main objective of this optimization is to select the best repeater
output transmission level as a function of frequency for the given
conditions. For a noise-limited system such as L5, the total system
noise should be minimized. For an overload limited system, the
average power output of repeaters should be minimized so that the
total system noise still satisfies the requirements. In both problems, we
may consider the modulation coefficients as additional design parame-
ters subject to certain limitations. All these problems can be considered
special cases of minimizing the following criterion function

2,-++,m— 8,
Ym—2(X) = average value of the second-order modulation coefficient
(in dB),
Ym—1(X) = average value of the third-order modulation coefficient
(in dB),
Ym(X) = equivalent rms power of a single tone at repeater output
(in dBm),

r; and w; are the corresponding requirements and weighting constants,
p is a positive even integer, and the components of x are the coefficients
of the polynomial representations of the transmission level and modula-
tion coefficients as functions of frequency. The total system noise
function in the equation is the power addition of thermal, second-order,

Solutions to various design problems can be carried out using NORATS
by choosing appropriate weighting constants in the criterion function.
For a noise-limited system, the weighting constant w, corresponding
to the output power is set to zero, since this allows the optimum trans-
mission level to be determined by minimization of total system noise,
regardless of output power. A relatively large wn corresponds to an
overload-limited system. The weighting constants wm_1 and/or Wm_2
are set to zero if the corresponding modulation coefficients are given
instead of being treated as design parameters. Recent NoRATS runs
have confirmed that earlier designs approached an optimum.

The strategy of using distributed corrections for the separate tem-
perature variations of the cable and the repeaters (regulation), and for
deviations from nominal frequency shaping caused by manufacturing
tolerances (equalization), is a complex one. The strategy depends on
accurate characterization of the various equalizer and regulator cir-
cuits to be effective, and this was considerably simplified by the use of
corms. One special program for alignment during manufacture deserves
description. It illustrates that, in some problems, even the most
sophisticated measurement equipment alone is not enough to fill the
need. By incorporating within the computer-controlled test set an
algorithm for adjusting a set of nonorthogonal controls, considerable
time and cost savings are achieved.

The 4211A network generates a Vf shape for use in regulating the
L5 system against changes in cable loss with temperature.” Six ad-
justable elements, which are not independent, are needed to match the
resulting shape to requirements. While their individual effects are
readily measured, the interdependence makes final adjustment during
manufacture nearly impossible by trial-and-error methods. The problem
could be solved on a separate computer, but this would require vol-
uminous data transfer and lost time. The logical time and place to
carry out the adjustment were simultaneously with the measurement
process. The details of the procedure and its effectiveness are described
in another article in this issue." This particular combination of pre-
cision measurement, data reduction, computer prediction, and output

“instructions” to an operator, all done in a single machine, will be the
way of the future for such complex tasks.

The complex design problems to be solved in the development of the
L5 system led to the writing of a series of new computer programs
that are highly interrelated and yet very useful independently. carr-
cop provides ac circuit analysis with flexible input and output, speed
of computation, and the ability to handle large circuits with very few
limitations of types of elements. Continued development is under way
to extend its range of application into the microwave region and to
improve its interactive and statistical capabilities. The Grop program
implements modern optimization techniques through a variety of
algorithms and considerable flexibility in types of constraints allowed.
Both epor and caPsE-cop contain links to simplify their combined use,
making an extremely flexible combination. The NopAP introduces a
new method of handling nonlinear distortion problems, working to-
gether with carE-cop. Input data for NopAP are provided by EPIc, a
transistor model parameter extraction program that works together
with cpop. A new system design program, NoRATs, also works with
GPop to optimize the noise performance of repeatered analog trans-
mission systems.

The usefulness of computer-operated transmission measurement
equipment was enhanced by the development of advanced impedance
measurement and modeling programs. Automatic measurement of
many repeaters under computer-controlled environmental conditions
was facilitated, along with data links to transfer measurement and
characterization data directly to another computer where they could
be used in caPE-cop. A specialized alignment program was written for
use in COTMS.

While each of these new tools is important in its own right, their
combination and coordinated development provided an extremely
powerful set of tools. Clearly, the L5 project could not have been
completed on schedule without these computer aids to design and
measurement.

Many individuals have contributed to the development and ap-
plication of computer and measurement aids for the design of the L5
system. This paper covers only the key aspects of the total involve-

ment. The authors wish to thank all the other contributors for their
enthusiastic support.

1. M. A. Murray-Lasso, ‘‘Analysis of Linear Integrated Circuits by Digital Com-
puter Using Black-box Techniques,” in G. Herskowitz, ed., Computer-Aided
Integrated Circuit Design, New York: McGraw-Hill, 1968.

2. P. E. Fleischer, “Optimization Techniques in System Design,” in F. F. Kuo
and J. F. Kaiser, eds., System Analysis by Digital Computer, New York:
John Wiley, 1966.

. L. A. O’Neill, “Interactive Tolerance Analysis with Graphic Displays,’’ Proc.
Spring Joint Computer Conference, 1969, pp. 207-213.

. C. L. Semmelman, E. D. Walsh, and G. T. Daryanani, “Linear Circuits and
Statistical Design,” B.S.T.J., 60, No. 4 (April 1971), pp. 1149-1172.

. G. D. Hachtel, R. K. Brayton, and F. G. Gustavson, ‘The Sparse Tableau
Approach to Network Analysis and Design,’ IEEE Transactions on Circuit
Theory, CT-18, January 1971, pp. 101-113.

. R. Fletcher and M. J. D. Powell, “A Rapidly Convergent Descent Method for
Minimization,’’ Computer Journal, 6, June 1963, pp. 163-168.

. R. Fletcher, “A Review of Methods for Unconstrained Optimization,” R.
Fletcher, ed., Optimization, New York: Academic Press, 1969.

. J. A. Nelder and R. Mead, ‘‘A Simplex Method for Function Minimization,”
Computer Journal, 7, January 1965, pp. 308-313.

10. A. B. Fiacco and B. P. McCormick, ‘“‘Programming Under Nonlinear Constraints
by Unconstrained Minimization: A Primal-dual Method,’ Tech. Paper
RAC-TP-96, Research Analysis Corporation, Bethesda, Md., September 1963.

11. E. H. Angell, Y. S. Cho, K. P. Kretsch, and M. M. Luniewicz, ‘“L5 System:
Repeatered Line,” B.S.T.J., this issue, pp. 1935-1985.

12. J. L. Garrison, A. Olsen, Jr., and T. H. Simmonds, Jr., ‘‘L5 System: Transmission
Networks and Magnetic Components,” B.S.T.J., this issue, pp. 2203-2248.

13. W. J. Geldart, G. D. Haynie, and R. G. Schleich, ‘A 50-Hz to 250-MHz Com-
puter-operated Transmission Measuring Set,’ B.S.T.J., 48, No. 5 (May-
June 1969), pp. 1839-1381.

14. S. Narayanan, “Applications of Volterra Series to Intermodulation Distortion
Analysis of Transistor Feedback Amplifiers,’”” IEEE Transactions on Circuit
Theory, CT-17, November 1970, pp. 518-527.

15. H. C. Poon, ‘Modeling of Bipolar Transistor Using Integral Charge-control
Model with Application to Third-order Distortion Studies,” IEEE Trans-
actions on Electron Devices, HD-19, June 1972, pp. 719-731.

16. Y. L. Kuo, ‘Distortion Analysis of Bipolar Transistor Circuits,’ IEEE Trans-
actions on Circuit Theory, (7-20, November 1973, pp. 709-716.

17. J. G. Evans, ‘Measuring Frequency Characteristics of Linear Two-port Net-
works Automatically,” B.S.T.J., 48, No. 5 (May-June 1969), pp. 1813-1338.

18. W. R. Bennett, ‘Cross Modulation Requirements on Multichannel Amplifiers
Below Overload,” B.S.T.J., 19, No. 4 (October 1940), pp. 587-610.

W. G. Albert, Bell Laboratories, 1951—. Mr. Albert was initially
engaged in the physical design of terminal equipment for the L3
Coaxial System and microwave transmission systems. He also worked
on the physical design and development of the A4 and A5 channel
banks, L multiplex, mastergroup multiplex, and L4 and L5 coaxial
system line repeaters. He is now supervisor of the Reliability and
Digital Banks Physical Design Group. Member, IEEE.

Robert E. Anderson, B.S.E.E., 1940, University of Wisconsin; E. I.
duPont de Nemours and Co., 1940-48; Radio Research Laboratory,
Harvard University, 1948-45; Bell Laboratories, 1945—. At Bell
Laboratories, Mr. Anderson has worked on the development of video
transmission systems for television, the L3 carrier, automatic-restoral
alarm circuits for N2 and N38 carrier, the equalizer remote control
system for L4 carrier, and most recently, on the transmission surveil-
lance system for the Ld carrier system. Member, IEEE, Eta Kappa Nu.

E. H. Angell, B.S.E.E., 1964, Union College; M.S. (E.E.), 1965,
Harvard University; Bell Laboratories, 1964—. Mr. Angell has been
engaged in circuit design work for coaxial systems. He is currently
supervisor of the L5 repeatered line group. Member, JEEE.

John F. Barry, B.S., 1961, M.S., 1963 (Electrical Engineering),
Northeastern University; Bell Laboratories, 1961-1972; Western
Electric, 1972—. Mr. Barry initially worked on the development of the
Bellboy® personal signaling receiver, the 3A Fm terminals, and the
2A 1F restoration switch system. Later he worked on the development
of the jumbogroup frequency supply. Since 1972, he has been a member
of a special design group.

R. K. Bates, AT&T Long Lines, 1941-1953; Bell Laboratories,
1953—. At Bell Laboratories, Mr. Bates has worked on private-line
telegraph switching systems including the 82B1 system for the U.S.
Navy and the Delta Airlines reservation system. In long-haul carrier
work, Mr. Bates has worked on the L4 system and, most recently, on
the basic jumbogroup trunk bay for the L5 system.

Philip J. Baun, Jr., B.S.E.E., 1959, University of Wisconsin;
M.S.E.E., 1961, Northeastern University; Bell Laboratories, 1959—.
Mr. Baun, a member of the Coaxial Systems Department, is responsible
for the L5 transmission surveillance software. He has had previous
experience in the design and development of passive networks and in
the analysis and characterization of electronic circuits for toll trans-
mission systems via analog and digital computation. Member, IEEE,
Tau Beta Pi, Eta Kappa Nu.

Richard Ming-Ming Chen, Dipl. El. Eng., 1959, Chiao Tung Uni-
versity, China; M.S., 1965, Pratt Institute; Ph.D., 1968, Rutgers
University; Faculty Member, El. Eng. Dept., 1959-1962, Peking
Constructional Industrial College, Peking, China; Bell Laboratories,
1968—. Mr. Chen has been engaged in the areas of optimization,
simulation, tolerance assignment, and partitioning techniques for
large scale problems. Member, IEEE, Eta Kappa Nu, Sigma Xi.

Yo-Sung Cho, B.S.E.E., 1962, Seoul (Korea) National University ;
M.S., 1966, and Ph.D., 1968, Yale University; Honeywell E.D.P.
Division, 1964-1965 and 1967-1969; Bell Laboratories, 1969—.
Mr. Cho made equalization studies of the L5 coaxial transmission
system employing manual and automatic equalizers. He was also
engaged in the development of the equalizer adjustment system
which is used for the equalization of the L5 line. His subsequent
work includes design of exploratory repeater amplifiers for a future
coaxial transmission system. He is presently supervisor of the group
developing terminal multiplexing equipment for coaxial and radio
transmission systems. Member, IEEE.

Fred A. D’Altroy, B.A., 1949, M.A., 1951, University of British
Columbia; Ph.D., 1956, Purdue University ; Bell Laboratories, 1955—.
Mr. D’Altroy has been engaged in the development of semiconductor
devices since 1956. He is currently supervisor of a group having
responsibility for the Triac pnpn devices, pnp transistors, and npn
ultralinear transistor.

Albert F. Flint, A. E., 1962, Wentworth Institute; Bell Telephone
Laboratories, 1962—. Mr. Flint has worked on the design and develop-
ment of precision crystal oscillators. Since 1971, he has worked mainly
on the development of clock oscillators for several systems.

J. L. Garrison, B.E.E., 19384, and M.E.E., 1936, Polytechnic
Institute of Brooklyn; Bell Laboratories, 1936—. Mr. Garrison has
worked on the design of transmission transformers, on the final de-
velopment of transistors, and on technical publications. He now
supervises a group engaged in development of transmission networks.
Member, Sigma Xi, Tau Beta Pi, registered professional engineer in
New Jersey and New Hampshire.

John H. Green, B.S.E.E., 1966, and M.S.E.E. 1968, Northeastern
University ; Bell Laboratories, 1966—. Mr. Green has been involved
in various terminal system developments for L-carrier systems in-
cluding mastergroup multiplex and line protection switching systems.
He is currently involved in digital channel bank development. Member,
Tau Beta Pi.

B. H. Hamilton, B.S.E.E., 1949, University of Kansas; Bell Labora-
tories, 1950—. Mr. Hamilton has worked on development of power
conversion and control circuits, new energy conversion techniques,
and medium- to high-voltage dc-to-de converters. He is presently
supervisor of a circuit-development group. Senior member, IEEE,
member, Kappa Eta Kappa, Sigma Tau, Tau Beta Pi.

Charles F. Hempstead, B.S., 1949, Northwestern University;
Ph.D. (Physics), 1955, Cornell University; Bell Laboratories, 1954—.
Mr. Hempstead designed millimeter wavelength backward-wave oscil-
lators and studied solids for maser applications until 1961. He was then
concerned with Type II superconductors for high magnetic field
generation, followed by research on visual perception of motion.
Presently he supervises a group developing new applications of wide-
band computer-controlled test equipment for precise component
characterization. Member, IEEE, Sigma Xi, Phi Beta Kappa.

Fred J. Herr, B.S.E.E., 1942, Cooper Union; M.S., 1952, Stevens
Institute of Technology; Bell Laboratories, 1936—. At Bell Labora-
tories, Mr. Herr was first engaged in the development of measuring
equipment for coaxial transmission systems. He was later concerned
with system design analysis of long-haul coaxial and type-N short-haul
carrier systems. He participated in early feasibility experiments on
color television transmission on coaxial systems and worked on the
design specification of new video measuring equipment. He participated
in the laying of the Alaskan and second transatlantic submarine cable

systems and did the system design analysis and terminal maintenance
planning for the sp submarine cable system. Currently, he supervises
the test equipment planning and application group for coaxial trans-
mission systems. Member, Tau Beta Pi.

Richard M. Jacobs, B.S. (Chemistry), 1954, Brooklyn College;
B.S.E.E., 1959, University of Wisconsin; M.S.E.E., 1961, Lehigh
University; Bell Laboratories, 1959—. Mr. Jacobs has been engaged
in the development of transistors and integrated circuits since 1959.
He is currently head of a department responsible for the development
of discrete devices and integrated circuits.

Frank C. Kelcourse, B.S.E.E., 1959, M.S.E.E., 1962, Northeastern
University ; Bell Laboratories, 1959—. Mr. Kelcourse worked initially
on Fp terminals, including the design of transmission amplifiers,
modulators, and carrier supplies, and subsequently on the design of
wideband feedback amplifiers for the L4 line repeaters. He has super-
vised groups responsible for the final development of the L4 equalizing
and remote control systems and for the equalization, planning, and
design of the L5 system. He is currently supervisor of a systems
studies and applications group responsible for analyses and appli-
cations related to analog, digital, and hybrid transmission systems.
Member, Tau Beta Pi.

Kenneth P. Kretsch, B.S.E.E., 1959, Pennsylvania State University ;
M.E.E., 1961, New York University; Bell Laboratories, 1959—. Mr.
Kretsch began at Bell Laboratories in the switching research area,
participating in research into time-division switching systems. He
also participated in the development of a message switching system and
was responsible for system design of high-speed processors for use in
telephone switching systems. Currently, he is responsible for equaliza-
tion of the L5 coaxial cable system. Member, IEEE.

Y. L. Kuo, M.S. (E.E.), 1961, Oklahoma State University; Ph.D.
(E.E.), 1966, University of California; Assistant Professor, 1966-1970,
Purdue University; Bell Laboratories, 1970—. Mr. Kuo’s primary
interest is in the area of active device modeling and computer-aided
analysis of nonlinear networks.

M. L. Liou, B.S., 1956, National Taiwan University; M.S., 1961,
Drexel Institute of Technology; Ph.D., 1964, Stanford University ;

Bell Laboratories, 1963—. Mr. Liou is presently supervisor of the
Analysis and Interactive Computing Group providing analytical and
computational support to the transmission system development at
Merrimack Valley. His fields of interest have included system theory,
numerical analysis, optimization, and computer-aided design of circuits
and various components in radio and cable transmission systems.
Member, IEEE, Eta Kappa Nu, Sigma Xi.

Michael M. Luniewicz, B.S.E.E., 1958, University of Massachusetts ;
M.S. (engineering), 1961, Northeastern University ; Bell Laboratories,
1960—. Mr. Luniewicz has been engaged in circuit design and develop-
ment for multiplex terminals and coaxial lines.

Robert E. Maurer, B.S.E.E., 1962, M.8.E.E., 1964, Ph.D., 1968,
Northeastern University; Bell Laboratories, 1962—. In addition to
supervising a group responsible for the development of analog multi-
plex equipment, Mr. Maurer has worked on the design of the equalizing
repeater for the L4 system, the analysis and modeling of intermodu-
lation distortion, the equalization of random channels, and analysis
and exploratory development related to the transmission of high-speed
digital signals over sharply band-limited analog channels. He presently
supervises a system-planning group working on a new baseband digital
system for exchange area application. Associate Editor, JEEE Trans-
actions on Communications. Member, Tau Beta Pi, Eta Kappa Nu,
Phi Kappa Phi, Sigma Xi.

Samuel Mottel, B.S.M.E., 1950, City College of New York;
M.S.M.E., 1968, Newark College of Engineering; Bell Laboratories,
1952—. Mr. Mottel has been concerned with physical design of power
equipment. He has worked on power for carrier systems, microwave
systems, submarine cables, key telephones, ringing and tone plants.
He supervises a group responsible for physical design of a variety of
power equipment.

Joseph M. Nacci, B.S. (Physics), 1956, University of Rhode Island ;
Bell Laboratories, 1956—. Mr. Nacci has been active in the design
and development of a wide variety of silicon transistors and integrated
circuits. These include pnp transistors of all types, npn ultralinear
transistors, silicon based capacitors, and pnp integrated circuits.

Sundaram Narayanan, B. Tech., 1960, Indian Institute of Technol-
ogy, Kharagpur, India; M.S., 1963, and Ph.D. (Electrical Engineer-
ing), 1965, Carnegie-Mellon University ; Bell Laboratories, 1965—. Mr.
Narayanan has worked on nonlinear distortion in transistor amplifiers
and the use of high-speed digital signals over band-limited analog
channels. He was supervisor of a group developing precision signal
source and is presently supervisor of a group working on a new multi-
plex arrangement for L5 and a basic repeater design for an advanced
coaxial system. Member, IEEE, Sigma Xi.

James F. Oberst, B.E.E., 1964, Manhattan College; M.S., 1966,
and Ph.D. (Electrical Engineering), 1969, Polytechnic Institute of
Brooklyn; Assistant Professor of Electrical Engineering, Polytechnic
Institute of Brooklyn, 1968-1969; Bell Laboratories, 1969-—. Since
joining Bell Laboratories, Mr. Oberst has worked on various aspects
of pcm transmission over cable and synchronization for FpM terminal
equipment. He is presently working on pcm channel banks.

Arthur Olsen, Jr., B.S.E.E., 1959, Worcester Polytechnic Institute ;
M.S.E.E., 1961, Northeastern University; Bell Laboratories, 1959—.
Mr. Olsen has been responsible for the design and development of
transmission networks and magnetic components. He _ presently
supervises a magnetic components group.

Edward J. Panner, B.S.E.E., 1962, Lafayette College; Bell Labora-
tories, 1949—. Mr. Panner has been engaged in device development
principally for transmission systems involving technologies ranging
from klystrons and general-purpose tubes to transistors and integrated
circuits. Member, Tau Beta Pi, Eta Kappa Nu, Phi Beta Kappa.

Henry S. Pustarfi, A.E., 1955, Newark College of Engineering;
Bell Laboratories, 1951—. Mr. Pustarfi has worked on the development
of quartz crystal filters, temperature-control circuits, and thermoelec-
tric ovens. He is presently engaged in the development of crystal-
controlled oscillators, temperature-control devices, and precision fre-
quency standards.

Richard W. Sanders, B.S.E.E., 1959, University of Vermont;
M.S.E.E., 1961, Northeastern University; Bell Laboratories, 1959-
1972; Western Electric Company, 1972—. At Bell Laboratories, Mr.

Sanders was engaged in the development of coaxial transmission sys-
tems, specifically protection switching for the L4 and L5 Coaxial-
Carrier Transmission Systems. At Western Electric, he is presently
involved in test engineering for the voice-band interface frame and the
digroup terminal for the No. 4 Ess project. Member, IEEE, Tau
Beta Pi.

T. H. Simmonds, Jr., B.S.E.E., 1954, University of Virginia;
M.S.E.E., 1961, Northeastern University; Active Naval Reserve,
1954-1958; Bell Laboratories, 1954 and 1958—. Mr. Simmonds’
early work was on a variety of filters and networks for long- and
short-haul carrier transmission systems. As supervisor of a networks
group in the Transmission Systems Networks Department he is
responsible for work on transmission filters and networks for carrier
and radio transmission systems. Member, IEEE, Tau Beta Pi.

Robert P. Snicer, B.S.E.E., 1966, and M.S.E.E., 1967, Massachu-
setts Institute of Technology; Bell Laboratories, 1967—. Since joining
Bell Laboratories, Mr. Snicer has been engaged in transmission system
development and computer-aided design. He presently supervises a
network design group. Member, IEEE, Eta Kappa Nu, Tau Beta Pi,
Sigma Xi.

John L. Thomas, B.S.E.E., 1957, University of Maine; M.E.E.,
1960, New York University; Bell Laboratories, 1957—. Mr. Thomas
engaged initially in circuit design work associated with special applica-
tions of submarine cable systems. He worked on systems analysis and
supervised a group responsible for the circuit design of shore terminal
transmission facilities associated with the sF submarine cable system.
He later supervised a group responsible for repeater, equalizer, and
special test set circuit design for submarine cable systems. He is
presently responsible for the design of transmission surveillance and
fault location circuitry for the L5 coaxial system. Member, Phi Kappa
Phi, Tau Beta Pi.

Edward D. Walsh, B.S.E.E., 1965, Gannon College; M. Eng.,
1966, and Ph.D., 1968, Rensselaer Polytechnic Institute; Bell Labora-
tories, 1968—. Mr. Walsh engaged initially in the frequency domain
characterization of high-frequency active and passive devices. He has

developed a general-purpose frequency domain simulation program
for transmission circuits. He is presently working on Monte Carlo
simulation programs for transmission circuits. Member, IEEE.

R. J. Wirtz, B.S. (M.E.), 1950, Brown University ; Bell Laboratories,
1956—. Mr. Wirtz was at first involved in resistor development and the
physical design of the N38 Carrier System. He later supervised the
physical design of the L4 Coaxial System. He is currently supervisor of
the Coaxial System Physical Design Group responsible for the design of
long-haul carrier systems.

Donald J. Zorn, A.A. (electronic engineering), Wentworth Institute,
1959; B.S. (industrial technology), Northeastern University, 1965;
Bell Laboratories, 1959—. Mr. Zorn has worked in the development
of A5 channel banks, L-multiplex, mmx-multiplex, and the L4 coaxial
system, and is currently working on the L5 coaxial system. In addition,
he has worked on several special development projects, including a
carrier supply for channel banks used in the initial Telstar operations
and a special pilot supply for the Norap headquarters carrier system.
Member, Sigma Epsilon Rho.

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