The Detection of Two Modulated Waves Which Differ
Slightly in Carrier Frequency—Charies B. Aiken

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Bancroft Gherardi Jewett
H. P, Charlesworth W. H. Harrison B. Colpitts
L. Morehouse H, D. Arnold 0. B, Blackwell
Philesader Norton, Editor J. ©. Perrine, Associate Editor

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The Detection of Two Modulated Waves Which Differ
Slightly in Carrier Frequency *

The present paper contains an analysis of the detection of two waves
modulated with the same, or with different, audio frequencies and differing
in carrier frequency by several cycles or more. Both parabolic and straight
line detectors are treated and there are derived the expressions for all of the
important audio frequencies present in the output of these detectors when
such waves are impressed. There are discussed the types of interference
which result when one station is considerably weaker than the other and
simple attenuation formule are employed in estimating the character and
extent of the interference areas around the two transmitters. Beyond the
use of such formule no attention is given to phenomena which may occur in
the space medium such as fading, diurnal variations in field intensity, ete.

length assignment wanders from its proper frequency, waves
are likely to be received which differ in carrier frequency by several
cycles or more. Under such conditions the two signals may be thought
of as made up of entirely distinct frequencies and phase relations
between analogous components of the two waves need not be con-
sidered. In the important case in which the carriers are of identical
frequency this is no longer true and phase and its dependence on
position and transmission phenomena must be taken into account.
This case will be reserved for future study, the present work being
limited to a consideration of the phenomena connected with the
detection of distinct frequencies. -

The most important undesired frequency which is present in
the output of the detector is the beat note between the two carriers.
It is sometimes carelessly assumed that if the frequency of this beat
note is reduced below the audible range the only remaining interference
will be due to the speech from the undesired station. Such is not the
case and it will be shown later on that when the beat frequency is
reduced below the audible range, but not to zero, there remains a group
of spurious frequencies which will introduce an interfering background.
When the undesired carrier is of relatively small intensity this back-
ground is a great deal stronger than the interfering speech. It is
therefore desirable to obtain quantitative data on the interfering spec-

trum which occurs in the receiver output, in terms of the intensities
and degrees of modulation of the input signals.

It is to be expected that the results obtained will depend, to some
extent at least, on the type of detector which is used. The square law
characteristic is a fair approximation to that of any detecting device
which is worked over only a small range and hence an analysis of this
characteristic may be expected to serve as an excellent guide to general
detector performance. When large signals are impressed on the
detector the functioning of the device may approximate more closely
to that of the ideal straight line detector. It has been felt that a study
of these two types would furnish data from which the performance of
any intermediate type of detector could be inferred without great error.
As the problem of the square law detector is very much the simpler it
will be considered first.

There will be assumed two broadcasting stations transmitting on
frequencies which differ by a relatively small amount, the beat fre-
quency being restricted to the audible range or less. Each of the
carriers will be assumed to be modulated by a single audio frequency,
the modulating frequencies at the two stations being, in general,
different. The total signal impressed on the receiving detector will
then be of the form

v is the total alternating voltage impressed on the detector.
E is the amplitude of the desired carrier.

m is the degree of modulation of the undesired signal.
w,/2m is the frequency of the desired carrier.

w2/2m is the frequency of the undesired carrier.
p/2x is the frequency of the desired modulation.
q/27m is the frequency of the undesired modulation.

We shall first suppose this signal to be impressed on a detector which
will be assumed to have a characteristic in the neighborhood of the
operating point, of the form

An expression of this type will accurately represent a small portion of
any continuous characteristic. The present analysis requires that the
impressed e.m.f. shall be of small amplitude in order that the limits of
the portion of the characteristic thus represented may not be exceeded.
This restriction is necessary in treating square law detectors.

The audio frequency output of the detector will be due entirely to
the second order term in (2). Hence it will be sufficient, for our
purposes, to square the expression for v. We are interested primarily
in the ratios of the amplitudes of the various undesired audio fre-
quencies produced to the amplitude of the desired signal of frequency
p/2x. Such a ratio will be designated as a relative amplitude. Neg-
lecting circuit constants, etc., which will apply equally in all the
expressions for the various frequencies, the amplitude of the desired
component of the audio frequency output is readily shown to be /2.1/.
The expression for v* is reduced to first power sinusoids and the ampli-
tude of each frequency converted to a relative amplitude by dividing
by #°M/. The case in hand yields twelve undesired audio frequencies,
the relative amplitudes of which are listed in table I. Before com-
menting on these results we shall consider the straight line detector.

TABLE I
Angular Ratio to | Angular | Ratio to
Velocity | | Velocity
M e
em
q EM | salmaaa 2EM
p + | em
e
EM

In making analyses of rectification by a straight line detector it is
customary to reduce the sum of the various impressed radio frequencies
to a single radio frequency, the amplitude and phase angle of which are
slow functions of time. The most common example of this type of
treatment is a combination of the carrier and two side bands of single
frequency modulation into the familiar expression for a modulated

wave in which the amplitude of the radio frequency is an audio
frequency function. In this case the radio frequency phase angle is
constant. In the case of a single frequency modulation with one side-
band eliminated there are impressed on the detector input only two
frequencies. These may be combined in a well known manner.!
Thus, if the impressed voltages are of the form acosx and bcosy, then
the amplitude is given by

The expression for the phase angle will not be given here as it can be
shown that if a and b are unequal and the difference between the
frequencies x/27 and y/27m is small compared with either frequency,
then the variation of the phase angle with time may be neglected in
computing the audio frequency components. In the present case we
have two radio frequency waves the amplitudes of which are not
constants but are slow functions of time and these may be substituted
for a and b in (3). Thus the effective amplitude of the total input
signal may be taken to be

u = @1— Wo.
The problem then resolves itself into an analysis of the detection, by a
straight line detector, of a single radio frequency component. The
results of such an analysis are well known and it can be readily shown
that the audio frequency output may be obtained, except for a factor
of proportionality, by resolving the amplitude into its audio frequency
components. In the present case the amplitude to be resolved is
given by (4) which may be written

The interfering signal B will be taken to be always less than the de-
sired signal A, and hence A? + B? > 2AB, from which it follows that
(A + B)? > 2AB(1 — cos ut.) Hence the radical may be expanded
by the binominal theorem, giving

It is to be observed that each of the terms of this series, except the
first, contains time in the denominator and hence further expansions
are necessary. The denominators of the various terms can be ex-
panded by the binominal theorem in such a way as to put all the
expressions containing time in the numerators, the expansions being in
powers of
(ME cos pt + me cos gt)/(E + e).

By the proper trigonometric transformations it is possible to reduce the
final expression for S to frequencies in p, g, « and the sums and differ-
ences of the various multiples of these quantities. An additional
discussion of this analysis is given in an appendix. In order that the
various series involved may converge with a manageable degree of
rapidity it is necessary to limit the relative amplitudes of the interfering
carriers and the degrees of modulation as well. Consequently the
solutions are restricted to intensities of the interfering carrier of 0.1, or
less, of the desired carrier and to degrees of modulation of either signal
ranging from 0.1 to .5. These limits are suitable also because we are
interested chiefly in interference by a relatively weak signal, the inter-
ference caused by a signal, the carrier amplitude of which is greater
than 0.1 of that of the desired carrier amplitude being near the tolerable
limit in the majority of cases. The upper value for the modulation of
0.5 is approximately equal to the average degree of modulation of a
station employing as deep modulation as is practical, only the peaks
running up to nearly unity. The value of 0.1 for the lower limit is of
course transgressed by soft passages in speech or music. However, the
range here specified is sufficiently large to give an excellent idea of
what may be expected from various degrees of modulation of desired
and interfering signals and the results of more extreme cases may be
inferred from the data here developed. Under these limits it is found
that the only audio frequencies of any importance which appear in
the output are:

It is now possible to make a comparison between the performance
of the straight line and the square law detectors. In Figs. 1 to 4 are
shown the relative amplitudes of the interfering frequencies in the two
cases for various degrees of modulation. The data for the square law
case are indicated by dashed lines and for the straight line case by
solid lines, and where the two coincide this is noted on the figures. It
is to be noted that the expression for the amplitude of the desired
frequency p/27 is a complicated function. However, computation
shows that over the range in which we are interested, the value of this
expression does not differ from .\JE by more than 1 per cent and,
therefore, this value has been assumed in computing the relative
amplitudes of the other frequencies.

Probably the most striking feature to be noted-in comparing the two
cases is the similarity of the results. This is particularly evidenced by
the carrier beat note of frequency u/27 the amplitude of which differs
in the two cases by an inappreciable amount. The spurious fre-
quencies (¢ + «)/2m also are practically identical for both detectors.
There are, however, several important differences as follows:

The group of spurious frequencies of angular velocity p + q + u,
which is of appreciable importance in the square law case, is entirely
absent from the range of magnitude considered when a straight line
detector is employed. The frequencies (p + u)/27 are greater in the
square law case over the range which we have considered, but the
curve which represents them has a smaller slope than in the straight
line case and for larger values of the interfering signal the intensities of
these frequencies would be relatively less with the square law detector.
The intensity of the undesired speech q is definitely less in the straight
line case than in the square law case but the slope of the g curves is

about the same for both except for 1J = m = 0.5. It is of interest to
observe that the interfering speech received on the straight line detector
is very much less in intensity than would be the case if the strong
desired signal were absent, and that the variation of the amplitude of
this frequency with intensity of the undesired carrier is greater when
the desired frequency is present. We have here an analytical descrip-
tion of the familiar masking effect which occurs when a strong unmodu-
lated carrier is received simultaneously with a weak modulated signal.
For example, when e/E = 0.1 it can be seen from Fig. 1 that the
relative amplitude of the component of frequency g/27 is 0.0063 for
the case of the straight line detector. If this component were un-
affected by the presence of the strong signal it would have an amplitude
proportional to em and a relative amplitude of em/E.\W which for the
values here considered is 0.1. Hence the ‘ masking" effect is here
responsible for a reduction of 24 db.

Lastly, it may be mentioned that there are in the case of the straight
line detector certain frequencies of small amplitude which are entirely
absent from the square law case. However, no frequency is shown
the relative amplitude of which is less than 0.01 for all four pairs of
values of .\/ and m, as such frequencies are unimportant. An exception
is made with regard to p + u. This is always less than 0.01 over the
range considered but is included for the sake of comparison with the
square law results.

The second harmonic of the desired signal is of importance only in
the square law case. It is of the nature of a distortion which is inde-
pendent of the interference and may be omitted from the consideration
of the undesired audio frequencies which are a result of the interference.
From Figs. 1 to 4 it is evident that the most important interfering
frequencies are those of angular velocity, +u, p+uand p+tg
+ u, the last being of importance only in the case of the square law
detector. It is with these frequencies, together with that of the
interfering speech g/27, that we shall be chiefly concerned.

When the relative magnitudes of the interfering frequencies, which
are tabulated on page 3, are multiplied by *.V/, the resulting quantities
are proportional to the absolute magnitudes of these frequencies. It
is to be noted that the frequencies of greatest interest have absolute
magnitudes which are linear functions of \/ or m except (p + q + u)/27
which is proportional to mM, and u/2x which is independent of both 1/
and m and will, therefore, be unaffected by the type of modulation
employed at either station. In case there are several frequencies

present in the modulation of each station the radio frequency waves
will be of the form E(1 + VM, cos pit + Me cos pot + +++) cos wit and
e(1 + m, cos git + me Cos got + +++) cos wet. For every frequency of
the former case which contained ./ as a factor of its amplitude we shall
now have several frequencies respectively proportional to 1,, M2 etc.
while an analogous new group will correspond to the former frequencies

DESIRED SIGNAL=E (i+MCOSPT) COS w,T
>, UNDESIRED SIGNAL = e(i+MCOSQT) COSwaT
3 |_ DATA FOR SQUARE LAW DETECTOR ————
Ww DATA FOR STRAIGHT LINE DETECTOR
&
fa)
ied
2,
2
Ww
/
/
a
no!
3 o
Q
< es
VA)
¥
/ |
.001 CY, |
0001 001 01

Fig. 1—Relative amplitudes of undesired frequencies as a function of the ratio of
the amplitudes of the desired and the interfering carriers. Modulation of both
stations small and equal.

containing m. Hence we shall have two frequency spectra derived
from the desired speech spectrum containing the p’s, but one of the
spectra will be shifted upward in frequency by an amount u/27 and
the other downward by the same amount. Two additional spectra
will be derived in a similar manner from the undesired speech spectrum
containing theg’s. The frequencies of the type (p + q + u)/27 will be

numerous as there will be a product of the .\/’s with each of the m’s.
However, these are of even moderate importance only when the
modulations of both stations are high, and a square law detector is
employed at the receiver.

Hence we may picture the interference as made up chiefly of dis-
placed frequency spectra of the type mentioned above, of a carrier

Fig. 2—Relative amplitudes of undesired frequencies as a function of the ratio
of the amplitudes of the desired and the interfering carriers. Modulation of desired
station small and of interfering station large.

beat and of the interfering speech, which is weak but important because
of its intelligibility. The results in the case of a straight line detector
would not be very greatly different. The frequencies of the type
(p +q +u)/2r would be negligible, the two spectra derived from
p + u would be much less important and certain new, but rather small
cross product frequencies would appear.

In estimating the interference the carrier beat can be considered by
itself and from the data at hand there can be derived the areas around
each of two stations having approximately the same carrier frequency,
inside of which the amplitude of the beat note will be down a given
number of db from that of the desired speech. The same is true of the
interfering speech when it is different from the desired speech. The

Fig. 3—Relative amplitudes of undesired frequencies as a function of the ratio
of the amplitudes of the desired and the interfering carriers. Modulation of desired
station large and of interfering station small.

frequencies (p + u)/27,(q + u)/27,(p & q + u)/27z, etc., will combine
to form a disturbing background which we shall designate as “displaced
side band interference." This may be taken to include all of the
interfering frequencies except those of the undesired speech and its
entirely unimportant harmonics. (The frequency 2/27 is not here
classed as an interfering frequency.)

From Figs. 1 and 4 it is to be noted that when m = M the frequencies
(g + u)/2e are the largest components of the displaced side band
interference if a straight line detector is used and have the same
amplitude as the (p + u)/27 components if a square law detector is
used. When m > VM the g + u group is much more important than
the p + u group as is evident from Fig. 2. When \J > m the g + u

f
4 “|
44
Ol
4
207 fy
A |
7
/
fe?
001 af
000! e OO! Ol
/e= RATIO OF CARRIER AMPLITUDES
Fig. 4—Relative amplitudes of undesired frequencies as a function of the ratio of

the amplitudes of the desired and the interfering carriers. Modulation of both
stations large and equal.

group is less important but this case is of no great interest for if the
stations are transmitting identical programs, with similar degrees of
modulation, it cannot occur and if the programs are different then the
interference is determined primarily by what happens when m > JM.
Consequently we may consider that the g + u group constitutes the
most important part of the displaced side band interference except

when a square law detector is used and the programs are identical. In
such a case we shall assume that both stations employ the same degree
of modulation and that therefore the g + u and p + wu groups are of
the same importance.

We have distinguished between three types of interference, namely,
carrier beat, unwanted speech and displaced side band. We shall now
compute, for several values of attenuation, percentage modulation
etc., the areas around a transmitting station inside of which each of
these types of interference, due to a second station, will have a relative
importance which is not greater than a certain specified amount.

In estimating these areas we must deal with two possible cases which
may arise in practice: (1) The two stations transmit different programs.
(2) The programs are the same. The carriers are assumed to differ in
frequency in both cases.

The importance of the various types of interference which are
present, will be determined by their ratios to the intensity of the
desired speech. In the present case in which the two stations transmit
different programs, the amount of interference which may be tolerable
will be determimed by what occurs when the modulation of the desired
station is low, while that of the interfering station is high. Hence, in
studying this case we shall make use of Fig. 2, which gives data com-
puted on the basis of a modulation of 0.1 for the desired station and
0.5 for the interfering station.

Taking up first the consideration of the carrier beat note, we shall
determine the curve along which the intensity of the beat is down a
given number of db from the desired speech. The position of this
curve will depend on the degree of modulation of the desired signal,
since the lower the modulation the more noticeable will be a beat note
of a given intensity. When we have specified the db difference which
must exist between these two components of the receiver output the
carrier ratio can be picked off from the u line of Fig. 2.

In order to determine the curve along which this carrier ratio exists
we shall proceed as follows:

The desired station will be considered to be at the origin of a system
of rectangular coordinates and the undesired station will be at the
point (D, O). We shall assume that the powers of the desired and
undesired stations are P; and P2, respectively, and that their distances
from a point in the coordinate plane are d; and d2; then if we denote the

This equation is based upon a convenient form of the Austin-Cohen ?
formula for the intensity of the field radiated from a radio transmitter.

in which J is the wave-length in meters, d is the distance from the
transmitter in miles and @ is an attenuation constant which may range
from zero up to 0.01 or even more. In writing down equation (8) we

From (8) there have been computed curves for the case in which
P, = P, and for various values of K and @. X has been taken as 300
meters and D, the distance between the stations, as 1,000 miles.

In Fig. 5 are shown several curves for a = 0.001. For small values
of K, the curves are practically circular and are of small area. As K
increases, the curves become oval shaped and it can be readily shown
that for values of K greater than a certain critical amount, the curves
will not close but will be of a shape which is roughly hyperbolic.

In Fig. 6 are shown curves corresponding to a value for @ of 0.002.
It is to be noted that an increase in a enormously increases the area
inside of which the ratio of the carriers is less than a certain value.
The effect of a will of course be dependent upon the magnitude of the
distance between the stations and will be more pronounced the larger
this distance. For the present case in which D = 1,000 miles, there
is not much point in considering values of a larger than 0.002, since the
attenuation would be so great as to make the effect of one station on the
service area of the other of very little consequence.

If we specify that the carrier beat must be at least 40 db down from
the speech output due to a 10 per cent modulated signal, then curve 1 of
Figs. 5 and 6 will represent the areas inside of which this requirement
will be met, while if we call for an interval of 20 db between these two
components, curve 5 of Figs. 5 and 6 will represent the areas in which the
condition is satisfied. It is evident that if a rigid restriction is placed
on the permissible beat note interference which may be allowed, and if
the attenuation is of a small value then the area in which the beat

note may be neglected is extremely small. On the other hand this
area increases very rapidly as the attenuation increases.

We may use the same sets of curves in considering the displaced
side band interference. From Fig. 2 it is evident that by far the most
important components of this interference are those represented by the
(g + u) group. In order to estimate this interference we must follow
some rule for combining the g + u component with the g — u com-
ponent. In order to do this in a strictly correct manner we should
have to take into account the frequencies and sensation levels of the
components. However, it has been shown * that over a considerable
portion of the audio frequency range, and for sensation levels of
approximately the magnitude in which we are interested, the inter-
fering effect of these frequencies may be taken to be approximately
equal to that due to a single frequency of twice the amplitude of
either component. We shall therefore take our data from the dash-dot
curve of Fig. 2. From this curve it appears that if the displaced side
band interference is to be 40 db down from the desired speech, we must
have a carrier ratio of 0.002, while if it is to be 20 db down from the
desired speech the corresponding carrier ratio is 0.02. The curves
corresponding to these values are shown by 2 and 6, respectively, on
Figs. 5 and 6.

From this it appears that the area in which the side band noise is not
objectionable may be a great deal larger than that in which the carrier
beat is of a tolerable intensity. If the frequency of the carrier beat is
reduced below the useful audible range then the former area may be
considered to be entirely free from interference of any kind. Conse-
quently, it is highly desirable to limit the maximum possible differences
in the carrier frequencies to a value which is definitely below the audio
frequency pass band of commercial radio receivers and loud speakers.

Turning now to the undesired speech, we note that it is of very little
importance compared with the displaced side band interference. Thus,
if this speech is to be 40 db down from the desired speech, the value of
the carrier ratio is 0.044 for the case of a square law detector, while for a
difference in level of 20 db, the carrier ratio is 0.14. A curve for the
case of a 40 db difference is indicated by 7 of Fig. 5.

The comparison between curves 7 and 6 emphasizes the fact that we
may have considerable areas of intolerable displaced side band inter-
ference in which the intelligible speech from the undesired station is
not noticeable. Of course, this interference is often classed as distorted
speech but the distinction is convenient in the present discussion.

3 J.C. Steinberg, ‘‘The Relation Between the Loudness of a Sound and its Physical
Stimulus,”’ Phys. Rev., Sec. Ser., Vol. 26, pp. 507-523.

In this case the programs are identical and consequently the speech
from the two stations will undergo simultaneous fluctuations of
intensity. We shall here assume that the two stations have the same
degree of modulation at any instant. We may then take our data
from the curves for which 1f = m. However, this does not apply to
the carrier beat note, since its intensity is independent of the degree of

Fig. 5—Curves along which the ratio of the carrier amplitudes received from two
stations has a constant value K, as indicated. Attenuation small.

modulation of either station and its interfering effect will be determined
by conditions which exist when the desired station has a low degree of
modulation. Hence the discussion of this component of the inter-
ference will be exactly the same as in the preceding case.

Referring to Figs. 1 and 4, it is evident that by far the greatest
portion of the displaced side band interference is due to the gq + u
components, in the case of the straight line detector, and the g + u
and p + « components in the case of the square law detector. The

identity of the curves for these components in the two figures shows
that the degree of modulation has practically no effect on the relative
importance of the interference which occurs when the same programs
are transmitted.

If we again assume that the total interference may be represented by
a fictitious component of twice the amplitude of the g + u component,

Fig. 6—Relative amplitudes of undesired frequencies as a function of the ratio of
the amplitudes of the desired and the interfering carriers. Attenuation constant
a twice that of Fig. 5.

we may take our data from the dash-dot line of Fig. 4. This should
represent the case fairly well for the straight line detector but when a
square law detector is used, greater interference should result due to the
importance of the p + u terms. However, we shall consider only the
q + u group and the phenomena associated with the square law case
may be readily inferred. In order that the displaced side band
interference may be 40 db down from the desired speech the carrier
ratio must have a value of 0.01, while if it is to be 20 db down, this
value must be 0.1. The first value corresponds to curves 5 of Figs.

DETECTION OF TWO MODULATED WAVES 17
5 and 6, while the second value corresponds to curves 8. We observe
that there is a tremendous difference between the areas which may be
considered to be free from displaced side band interference and those
which will be free from carrier beat interference, in case the beat
frequency is allowed to wander into the audible range. The comparison
between the two areas is given by curves 1 and 5 for the 40 db interval
and by curves 5 and 8 for the 20 db interval.

The speech from the interfering station will now be the same as the
desired speech and can have effect only in so far as it adds to or subtracts
from the desired speech. It will be noted from Figs. 1 and 5 that for
carrier ratios of less than 0.1 this component is always down more than
40 db and may be safely neglected.

The foregoing discussion serves to illustrate the types of interference
which may be expected when two stations are operated on approxi-
mately the same frequency. The data discussed have involved low
values of attenuation. This is of particular interest when the distance
between stations is large since with high values of attenuation either
station will have very little effect on the service area of the other. Of
course at night time we may have signal strengths which will be of the
order of magnitude of that given by the simple inverse distance law
involving zero attenuation. This possibility probably presents a
serious limitation on night time common frequency broadcasting but
should be of little consequence during the daylight hours. Conditions
will be somewhat different for stations that are placed nearer together
and specific results can be readily computed for any given spacing.
The equations which have been discussed can be applied to any such
case and the areas corresponding to those in Figs. 5 and 6 determined.

One point which is emphasized by the results which have been
obtained is, that with a carrier frequency difference of several cycles
satisfactory reception cannot be expected in the regions which lie
midway between two transmitters. The field strength of one station
must be at ali times predominately higher than that of the other and
consequently the use of pseudocommon frequency broadcasting should
be restricted to stations of wide geographic separation. It should then
be possible to furnish high grade service to relatively small densely
populated areas in the immediate vicinity of either transmitter,
reception at a considerable distance from both stations being ad-
mittedly unsatisfactory. However, if the carriers are strictly isoch-
ronous much larger service areas should be feasible.

APPENDIX
Equation (5) is
I II
AB(1 — cos ut)
Ill IV
_ A*B(1 — cos ut)? A*B(1 — cos ut)? (5)
2(A + 2(A + B)®
To expand these terms we write
1 1
(A+B)"” (E+e+ME cos pt+me cos qt)"
o 1 (1 _ n(ME cos pt+me cos qt)
~ (E+e)" E+e
4 n(n+1)(.ME cos pt+me cos qt)?
+e)?
rE +e)"

It is evident there are present in S an infinite number of frequencies
and it is necessary to select those which are of appreciable magnitude
relative to that of the desired frequency of amplitude E./.  Fortu-
nately these are not very numerous.

In deciding whether or not a given term should be retained there
are two points to be considered: (1) whether all the terms of a given
frequency total to a value sufficiently large to call-for the presence of
this term in the final result; (2) what per cent accuracy should be
required in the frequencies which are retained. Thus if it is desired to
retain all frequencies the relative amplitude of which is greater than
_ 0.01 we cannot arbitrarily retain all individual terms which make a
contribution of 0.01 or greater and neglect those of relative importance
of less than 0.01. Thus if a term of a given frequency has a relative
amplitude of 0.01 and another term of the same frequency a relative
amplitude of 0.009 the second term should be retained. Otherwise we
should have a large percentage error in the value of the amplitude of
this frequency. On the other hand it is not desirable to maintain
the same degree of accuracy for the case of retained frequencies of
slight relative importance as for those of large importance. As a
compromise all individual terms have been retained which, after
division by E.M, are of a magnitude greater than 0.005 for any values
of \/, m and e/E which are here dealt with. An exception is made in

the case of a term in cos pt derived from term III of (5). This term is
slightly larger than the above limit when J = 0.5 and e/E = 0.1 but
as it decreases rapidly with a decrease in e/ it has been omitted for the
sake of simplicity.

Having chosen this limit of 0.005 for the relative magnitude of
individual terms it can be shown to be permissible to neglect term IV
and all subsequent terms of (5). Furthermore, only a few of the large
number of terms yielded by III need be retained.

After applying these rules there appear several frequencies that are
never as large as 0.01 in relative magnitude and these have been
omitted from consideration. As has been stated in the body of the
paper, an exception is made in the case of the frequencies (p +g +1)/27.
If a given frequency exceeds 0.01 for any one of the four pairs of
values of ./ and m, it has been shown on the figures for all of the
pairs.

After the formula (5a) has been applied to S and the expressions for
A and B inserted there remains the necessity of reducing products and
powers of various sinusoidal terms to sums of simple first order
sinusoids. This is a tedious procedure but is a matter of simple
trigonometry and will not be set forth in detail.

From (5a) it can be seen that if .W or m is near unity the series will
converge very slowly. Furthermore, since to obtain relative magni-
tudes we divide by .V/, it is impossible to obtain satisfactory convergence
due to small values of ./ in the denominator. Hence it is necessary to
limit ./@ and m to 0.5 or less and in addition .\/ must be no smaller than
0.1. It would be permissible to allow m to become less than 0.1 but as
little would be gained by this m has been restricted to the same range
as M.

Por accurate determinations of hysteresis loops and initial magneti-
zation curves of magnetic specimens, a laborious routine involving
the use of a ballistic galvanometer is usually necessary. This article
describes an apparatus by means of which these curves may be obtained
photographically with quantitative accuracy. Attempts to devise
such a scheme have previously been made. Ewing! describes one
which was used with short, thick specimens in a magnetic yolk.
Fleming? invented a device, the Campograph, which made use of a
magnetometer and had the advantage of making possible the use of
long, thin, specimens, thus reducing eddy current and demagnetization
effects. J. B. Johnson * describes the most recently published design,
embodying a vacuum tube amplifier and a Braun tube oscillograph.
This hysteresigraph is used with frequencies of the order of five cycles
per second, or higher, and consequently introduces an eddy current
loss, a disadvantage in a great many measurements.

The greatest difficulty has always been to devise an instrument
which would accurately record the total change in magnetic flux in
the specimen. The ideal instrument would be a fluxmeter with no
- restoring force and no friction. Fluxmeters are on the market in
which the restoring force is negligible only over short periods of time
or in which there is no restoring force but where the friction is appreci-
able; but if it is required that the magnetic cycle have a period of more
than a few seconds, such fluxmeters are out of the question. In
addition they require that the search coil be of such low resistance
that it must have too few turns for use with long thin specimens, in
which the flux is small. These difficulties have been overcome in the
apparatus described below, in which the principal feature is the use of a

fluxmeter in which the suspended coil has its restoring torque counter-
balanced for all deflections within a range sufficient for accurate
delineation of magnetic curves.
DESCRIPTION OF THE APPARATUS

The operation of the apparatus is as follows: a long, sensitive, photo-
electric cell is fitted with a V-shaped slit, as shown in Fig. 1; a beam of

lig. 1—The photoelectric cell circuit.
light is reflected from the mirror of the fluxmeter and focused on the
slit of the photo-electric cell, which is connected, in series with a
source of e.m.f., across the terminals of the fluxmeter; the e.m.f. is
adjusted once for all to such a value that, if the beam is at rest when
at the narrow end of the slit, at any other position the current con-
trolled by the cell will develop a torque in the Muxmeter coil which just
balances the restoring torque of the suspension. The fluxmeter
deflection will then be proportional to the change of flux which has
occurred within the search coil S. It may be found necessary to shape
the slit empirically to correspond to the unequal sensitivities of the
photo-electric cell at different spots. The fluxmeter used is a Leeds
and Northrup type 2290 HS galvanometer. It has a critical damping
resistance of about 100,000 ohms, and when used with about one
hundred ohms in the external circuit it is much over damped.

for changing the magnetic field in the specimen, is shown in Fig. 2. A
drum D, carrying photographic paper, is placed in a light-tight box
provided with a long, narrow slit parallel to the axis of rotation of the
drum. A beam of light froma second lamp is reflected by the fluxmeter
mirror and focused on the slit. This beam is reflected by the same
mirror which reflects the beam onto the photo-electric cell, the two

beams being incident at different angles. Attached to the shaft of
the drum is an arm A, which slides along the rheostat R. A battery B
is connected across R, and a center tap soldered to it. Between the
arm A and the center tap a varying e.m.f. is produeed which is applied
to the field coil #. This e.m.f. reverses its sign every time the arm A
slides past the center of the rheostat, and the latter is curved in a
manner calculated so that the field current will be proportional to the
angle of rotation of the drum from the position for zero current. The
search coil S of Fig. 1 is placed within F, and consequently when D is
rotated it moves the photographic paper past the slit so that the
distance moved is propwrtional to the change in field current, while at
the same time the fluxmeter deflects the beam of light along the slit so
that the deflection is proportional to the time integral of the changes
of flux within S. As the drum is turned from one position to another,
a curve with rectangular axes is thus registered, the scales of which
may be calibrated in terms of Band H. Figs. 4 to 7 are some examples
of curves taken with the apparatus.

In Fig. 3 the electrical circuits are shown in detail. R; is the rheostat
controlling the field current, and A is the arm which rotates with the

out thermo-electric potentials and current from the photo-electric cell
PHOTO

due to stray light. R,and R; are adjusted according to the amount of
flux in the specimen, in order to keep the maximum deflection within
the desired limits. The mutual inductance J/ is used to balance out
the potentials produced in S when no specimen is within it, so that the

fluxmeter deflection is proportional to the change in B -- 77. The
drum is conveniently rotated by an electric motor, connected by gears
so that the drum makes about one revolution in two minutes, and it is
desirable to have this rate variable. The motor may be reversed, so
that complete hysteresis loops may be recorded.

In setting up the apparatus the photo-electric cell may be con-
veniently placed above or below the drum, and one lamp above the
other. The lamp used to illuminate the photo-electric cell should
furnish a brilliant beam, and it was found that a 250 watt Mazda
projection lamp was quite satisfactory.

The circuit is calibrated by passing a known current through the
primary of a known mutual inductance, the secondary of which is
connected in series with the search coil S. By measurement of the

deflection _produced the relation between the quantity of electricity
passing through the fluxmeter and its deflection can be determined.
From this relation and other known constants the change in induction
of a magnetic specimen producing a given deflection may be calculated.

magnetic specimen be removed, Ry and R, be set on infinite resistance,
the magnetizing coil F be shorted, and a change in the field current

Fig. 8—Line taken for calibrating the apparatus.
= instantaneous primary current,
instantaneous secondary current,
resistance of secondary circuit,
mutual inductance of ,
self inductance of secondary circuit.

where Qs; is the quantity of electricity that has passed through the
fluxmeter in time fo. Now let Qa = — Kéy, where 6 is the deflection
produced when Qy flows. Then:

and the quantity of electricity which has passed through the luxmeter
for any other deflection is
1
(1)

Now suppose a magnetic curve recorded with R, adjusted until the
deflection is due solely to the magnetization of the specimen. Let the
resistance of the fluxmeter plus that of the secondary of M be denoted
by R,, and that of S plus R; be denoted by R,. Then if the field

current ig is varied slowly enough, the time lag in the secondary
circuit will be negligible and we shall have for the instantaneous
current in the fluxmeter:

where A is the cross sectional area of the specimen and _N is the number
of turns in the search coil. Then

r. being the total secondary resistance when K was determined. This
equation, then, gives B — H for any given deflection 4, in terms of
known constants. For any fluxmeter, K is determined once for all by
passing the current 7 through a mutual inductance and measuring the
deflection 54 on a photographic record. The other constants are
changed in a calculable way when the number of turns in the search
coil, the resistance settings, and the cross-sectional area of the sample
are changed.

Since it is the voltage applied to the magnetizing coil F which is
proportional to the angle through which the drum has rotated, there

is a lag in the field current behind the field registered on the drum, due
to the self inductance of the coils. Added to this there is a lag in the
secondary due to its self inductance, and another lag due to the time
required for the fluxmeter to act. The effect of these is to widen the
loop. In Fig. 9 is shown a curve traced with no magnetic sample in
the field coil, and with d#//dt so great that the lag is appreciable.

Fig. 10 shows two loops, the outer one representing a loop as taken on
the apparatus, and the inner one the true loop corresponding thereto.
Let B be some induction near zero, on the traced loop. B will be
incorrect for the indicated value of JJ by an amount Bo -— B, such
that if the field were held constant at that point while the drum con-
tinued to rotate the curve would approach By as an asymptote, as
indicated by the dotted curve. If d///dt is not zero, B may be regarded
as momentarily approaching By as an asymptote. The equation for
B at any instant is:

where \, is the time constant of the circuit and Boy is not a constant”
but a function of HJ and ¢. If we assume that dB/d/H/ is constant for a
small region in the neighborhood of B = 0, we have, putting AJ/,
equal to the error in coercive force IJ,

Data taken with no magnetic specimen inserted show that this linear
relation actually exists. Added to this there is an increase in /7, due to

eddy current lag. Johnson * has derived an equation for this, and with
a slight modification to make it applicable to cylindrical specimens, it
is:

where p is the resistivity of the specimen, and r its radius. This gives
us for the total error,

with varying dH/di. The specimen used was a cylinder of 81 per cent
Ni permalloy, 60 cm. long and 0.1 cm. in diameter, and was placed in a
magnetic yolk. Its hysteresis loop, as shown in Fig. 11, has an

unusual slope, 225,000 at B = 0. This gives \2: = .055 sec. From
this series of curves the straight line showh in Fig. 12 was obtained, for
which A; + Az = 0.314 sec. By another set of loops in which the

deflection is produced by an air core mutual inductance, A, is found to
be 0.134 sec. This determines dz as 0.180 sec., in disagreement with
the value 0.055 sec. calculated from Eq. 5. Johnson assumes in his
derivation that dB/d// is constant and hence that the shape of the
curve before H, is reached has no effect on A/7,. It is probable that
if the equation were changed to allow for dB/dH being a function H,

that the difference could be accounted for. At any rate, this error is
negligible for all but specimens with exceptionally high dB/dH or
great thickness, and the true coercive force can always be found by
taking two loops with different values of d/Z/dt and extrapolating to
diI/dt = 0.

Another possible source of error is the passage of a large fraction of
the photo-electric cell current through the search coil, the field being
thereby altered. The maximum photo-electric cell current used is on
the order of 5(10)-7 amperes. Since the search coil is unlikely to have
more than about 400 turns per centimeter, this would make the
maximum error in /7 about 2.5(10)~ gauss, which is negligible for most
measurements.

As a test of the accuracy of the instrument, a comparison was made
with curves made by ballistic galvanometer measurements. Fig. 13
shows a loop taken of the specimen which Bozorth used in some
previous measurements.‘ Both the coercive force and the maximum
induction taken by the two methods agreed to within less than one

per cent. Fig. 14 shows an initial magnetization curve which gives a
value of the initial permeability agreeing accurately with the value
determined ballistically.

A fluxmeter with no restoring torque is also useful in certain types of
current measurements. If the average value of a current which
fluctuates too much to be read on a slowly moving meter is desired, it

Fig. 14—An initial magnetization curve of the specimen of 78.1 per cent
nickel permalloy.

can be integrated on the fluxmeter, and the average value obtained by
dividing the total quantity of electricity which has passed through by
the time during which the measurement was made. Also if a current is
too small to be read directly on a galvanometer it may be possible to
maintain it for a sufficient length of time to give a readable deflection
on the fluxmeter, and again the current will be obtained by dividing
by the time.

In conclusion I wish to thank Dr. R. M. Bozorth for suggestions
given during the development of the apparatus, and Mr. A. W. Metz
for his assistance in taking the curves.

A bar to the attainment of television images having a large amount of
detail is set by the practical difficulty of generating and transmitting wide
frequency bands. An alternative to a single wide frequency band is to
divide it among several narrow bands, separately transmitted. A three-
channel apparatus has been constructed in which prisms placed over the
holes in a scanning disc direct the incident light into three photoelectric cells.
The three sets of signals are transmitted over three channels to a triple elec-
trode neon lamp placed behind a viewing disc also provided with prisms
over its apertures so that each electrode is visible only through every third
aperture. An image of 13,000 elements is thus produced. For the suc-
cessful operation of the multi-channel system, it is imperative to have very
accurate matching of the characteristics in the several channels.

as they should be, of such a size as to be just indistinguishable, or
unresolved, at a given observing distance, the number of image ele-
ments increases directly with the area of the image. The number of
such indistinguishable elements in everyday scenes, in the news
photograph, or in the frame of an ordinary motion picture is aston-
ishingly large. An electrically transmitted photograph 5 inches by 7
inches in size, having 100 scanning strips per inch, has a field of view
and a degree of definition of detail, which, experience shows, are
adequate (although with little margin) for the majority of news event
pictures. It is undoubtedly a picture of this sort that the television
enthusiast has in the back of his mind when he predicts carrying the
stage and the motion picture screen into the home over electrical
communication channels. In this picture, the number of image
elements is 350,000. At a repetition speed of 20 per second (24 per
second has now become standard with sound films) this means the
transmission of television signals at the rate of 7,000,000 per second,—
a frequency band of 31% million cycles on a single sideband basis.
This may be compared to the 5,000 cycles in each sideband of the
sound radio program, or it may be evaluated economically as the
equivalent of a thousand telephone channels.

When we examine what has been achieved thus far in television, we
find that the type of image successfully transmitted falls very far
short of the finely detailed picture just considered. Probably the
most satisfactory example of television thus far demonstrated is the

72-line picture used in the two-way television-telephone installation of
the American Telephone and Telegraph Company in New York.!
Here the object to be transmitted is definitely restricted to the human
face, which fills the whole field of view, and is adequately rendered by
the 4,500 image elements used.

The gap between the 4,000 elements of this image and the 350,000
considered above is enormous, not only in figures, but in terms of
technical possibility of bridging. Even if we are forced to content
ourselves with relatively simple types of scenes for television trans-
mission, still the fact must be squarely faced that a very much larger
number of image elements must be transmitted than has thus far been
found possible; and a far wider frequency band utilized than has ever
been used in any communication problem. Now the situation is,
simply stated, that all parts of the television system are already having
serious difficulty in handling the 4,500-element image. Consequently,
a major problem in television progress is to develop means to extend the
practical frequency range.

It will be worth while to survey briefly the points in a television
system where difficulty is now encountered when the attempt is made
to increase the amount of image detail and the accompanying band of
transmitted frequencies. Consider in turn the scanning discs at
sending and receiving ends, the photoelectric cells, the amplifying
systems, the transmission channels, the receiving lamps.

In the scanning disc at the sending end, which we shall assume
arranged for direct scanning, increased detail means either loss of
light or increase in the size of the disc. In either case, the factor of
change involved is large. For instance, if the number of scanning
holes is doubled in a disc of given size, providing four times the number
of image elements, the holes must be spaced at half the angular distance
apart, and twice the number of holes, imagined placed end to end,
must be included in this half diameter scanning field. The holes will
therefore be of one-quarter the diameter or 1/16 the area. The light
falling on the photoelectric cell at any instant is the light transmitted
by one hole; in this case, 1/16 the amount with the disc of half the
number of holes. In general, the light transmitted by the disc to the
cell decreases as the square of the number of image elements. If the
disc is enlarged so as to hold the transmitted light unchanged, its
diameter increases directly as the number of image elements. It is
obvious that any considerable increase in the number of image ele-
ments—such as ten times—demands either enormously increased
sensitiveness in our photo-responsive devices, or quite fabulous sizes of

discs. Perhaps the most pertinent conclusion from this survey is that
the disc, while quite the simplest means for scanning images of few
elements, is entirely impractical when really large numbers of image
elements are in question. As yet, however, no practical substitute
for the disc of essentially different character has appeared.

Turning now to the photoelectric cell. The question of adequate
sensitiveness to handle a large number of image elements is intimately
connected with the method of scanning, as has just been brought out,
so that no simple answer is possible. It is, however, probable that a
very considerable increase in sensitiveness over anything now available
must be anticipated, whatever scanning device is adopted. In the
matter of frequency range there is definite information.? In cells
depending on gas amplification (such as argon or neon) a characteristic
behavior is a falling off of output with frequency, greater the higher
the voltage used, which, becoming noticeable at about 20,000 cycles,
may at 100,000 cycles be so considerable as to constitute a practical
block to transmission. Vacuum cells are free from this failing, but
are much less sensitive. Systematic experiment and development of
photoelectric cells with particular reference to extending their range of
frequency response is indicated as a necessary step in the attainment
of a many-element image.

Taking up next the circuits associated with the photoelectric cell, we
find, in general, that the higher frequencies progressively suffer from
the electrical capacity of cells and associated wiring and amplifier
tubes. This in turn calls for auxiliary equalizing circuits, with
attendant problems of phase adjustment, and for increased amplifica-
tion. Amplifiers capable of handling frequency bands extending from
low frequencies up to 100,000 cycles or over offer serious problems.

Communication channels, either wire or radio, are characterized by
increasing difficulty of transmission as the frequency band is widened.
In radio, fading, different at different frequencies, and various forms of
interference stand in the way of securing a wide frequency channel of
uniform efficiency. In wire, progressive attenuation at higher fre-
quencies, shift of phase, and cross-induction between circuits offer
serious obstacles. Transformers and intermediate amplifiers or re-
peaters capable of handling the wide frequency bands here in question
also present serious problems.

At the receiving end of the television system, conditions are similar
to the sending end. The neon glow lamp, commonly used for re-
ception, is already failing to follow the television signals well below
40,000 cycles, and, in the case of the 4,500-element image above
* Loc. cit., p. 456.

referred to, the neon must be assisted by a frequently renewed ad-
mixture of hydrogen, which again cannot be expected to increase the
frequency range indefinitely. In the scanning disc, as at the sending
end, increasing the number of image elements rapidly reduces the
amount of light in the image. With a plate glow lamp of given
brightness, the apparent brightness of the image is inversely as the
number of image elements.

From this rapid survey, it is clear that at practically every stage in
the television system, we encounter serious difficulties when a large
increase in image elements is contemplated. It is not claimed that
these difficulties are insuperable. One of the chief uses of a tabulation
of difficulties is to aid in marshalling the attack upon them. But the
existing situation is that if a many-element television image is called
for today, it is not available, and one of the chief obstacles is the difficulty
of generating, transmitting, and recovering signals extending over wide
frequency bands.

One alternative, which prompted the experimental work to be
described below, is the use of multiple scanning, and multiple-channel
transmission. The general idea, which is obvious from the name given
to the method, is to divide the image into groups of elements, the
various groups to be simultaneously scanned, and to transmit the
signals from the several groups through separate transmission channels.
In place of apparatus to generate and transmit a frequency band of n
cycles, we arrange m scanning processes each to provide frequency
bands of n/m cycles width; n/m being chosen as within the limits set by
the available practical elements of a television system. It will appear
that the method which has been developed does provide an image of
manyfold more image elements than heretofore, and may make easier
the problem of transmission over practical transmission lines.

The multi-scanning apparatus which has been constructed and
given experimental test uses scanning discs over whose holes are
placed prisms of several different angles. At the sending end, the
beams of light from successive holes are thereby diverted to different
photoelectric cells. At the receiving end, the prisms similarly take
beams of light from several lamps and divert them to a common
direction. The mode of action of the prisms is illustrated in Fig. 1a,
where a three-channel arrangement is shown, which is that actually
used in the experimental apparatus. In the figure, the disc holes are
shown disposed in a spiral, at such angular distances apart that
three holes are always included in the frame f. Over the first hole of a

set of three is placed a prism P, which diverts the normally incident
light upward; the second hole is left clear; the third is covered by a
prism P, turned to divert the light downward. If we imagine the
prisms removed and a single channel used instead of the three that are
proposed, it is clear that the holes would have to be spaced three times
as far apart so that no more than one would be included in the frame f
at one time. The diameters of the holes, and the radial separation of
the first and last in the spiral would be unchanged. Quite apart,
therefore, from the smaller frequency bands which are sufficient to
carry each of the three sets of signals, which is the principal objective
sought, there is realized in this arrangement a reduced size of apparatus
for the same size of disc holes.

Studying more closely the division of the light into three sets of
beams, it is important to note that the signals transmitted by any one
of the three sets of holes are continuous—as one hole of a given prism
series passes out of the frame the next of the same series comes in.
The signals generated in each photoelectric cell are accordingly exactly
like those of a single-channel system.

Before describing the details of the apparatus, the general relation-
ship between the number of image elements, band width, number of
channels, and shape of picture may be developed. For this purpose,
let the following symbols be used.

B = frequency band available in one channel, in cycles per second.
F = repetition frequency of images, per second.

a/b = ratio of tangential to radial dimensions of frame.
a = angular opening of frame.

We shall assume that the picture elements into which the frame is
imagined divided are symmetrical in shape, i.e. either circles or squares.
We then have that

the number of picture elements in the radial direction = number of
holes = n;
the number of picture elements in the tangential direction = (a/b)-n.

Now the number of signal cycles corresponding to this number of
elements is (1/2): (a/b)n.

and the number of cycles per second for each channel = (1/c)
-(1/2)(a/b) Fn? = B.
The angular opening of the frame a = 360  C.
The number of picture elements = m?*- (a/b).
These formule may be utilized upon assuming values for any of the
variables, to fix the values of the other. In the present case, it was
decided for reasons of simplicity to restrict the number of channels to 3.

Fig. 1—Schematic of three-channel television apparatus. (a) Receiving end
disc with spiral of holes provided with prisms. (6) Sending end disc with circle of
holes provided with prisms. (c) General arrangement of apparatus.

The band width was chosen as that found feasible in the two-way
television system, namely 40,000 cycles. The picture shape chosen
was that of the sound motion picture, for which a/b = 7/6. The
repetition frequency assumed was 18 per second, again following
closely that of existing experimental synchronizing apparatus. Sub-
stituting these values in the formula rearranged to give , we get for
the number of holes,

scanning, in which the object is imaged on the disc, was chosen, since
beam scanning would introduce the problem of separating the light
reflected from the object from the several spots simultaneously pro-
jected from the disc. Since the light going through the disc must be
separated into several beams to be directed into separate photoelectric
cells, the full aperture of the image forming lens must be divided by C,
the number of channels, with a consequent proportional loss of light to
each cell. (This loss counterbalances the decreased size of disc above
noted.) It therefore becomes necessary to insure a very high illumi-
nation of the object. In the present case, it was decided to use motion
picture film to provide the sending end image, since this can have a
large amount of light concentrated through it by an appropriate lens
system.

The use of motion picture film permitted a simplification of the
transmitting disc, which is illustrated in Fig. 1b. This consists in
arranging the scanning holes in a circle instead of a spiral, and pro-
ducing the longitudinal scanning of the film by giving it a continuous
uniform motion at right angles to the motion of the scanning holes.
The continuous motion of the film also avoids the loss of transmission
time that an intermittent motion demands for the shutter interval.

At the receiving end, a spiral of holes is used as shown in Fig. la.
There again, because of the division of the light into three beams, the
angle which can be subtended by the light source (neon lamp) is much
restricted. In consequence, the neon lamp cathodes are of small area,
and a lens system has been used to focus their images on the pupil of
the observer’s eye. Other methods of receiving, which promise to be
less restricted as to position of observation, are possible, however, as
discussed below.

With this survey of certain of the more important features of the
system, we may proceed to a more detailed account of the apparatus as
constructed. The general arrangement of parts is shown in Fig. 1c
and in the photographs, Figs. 2, 3, 4 and 5 in all of which the symbols
are uniform. Both sending and receiving discs were, for simplicity of
operation, mounted on the same axis, driven by the motor \/. This
means that no question of synchronization entered. Synchronization
is in fact a separate problem, having nothing to do with multi-channel
operation and has been very completely solved in connection with other
television projects.' If it should be decided to transmit the multi-
channel image to a distant point, the apparatus could be cut in two
and each end, after separation to the desired distance, operated by
synchronous motors controlled in approved fashion. Similarly, no
long transmission lines were included.

Starting at the extreme right end of the schematic drawing Fig. 1c,
we have an arc lamp 4, a cylindrical lens L;, a condensing lens Ls, the
two lenses together concentrating a line of light on the film F. Be-
tween the film and the disc is a lens L; which images the film on the
disc. Directly behind the disc D,, with its circle of prism covered
holes, is a second cylindrical lens L4 which concentrates the transmitted

Fig. 2—Sending end of three-channel television apparatus, showing film driving
arrangements.

light laterally, upon the three photoelectric cells S;, Sz, S;. By virtue
of this lens arrangement, the light falls upon the cells in three small
practically stationary spots. Additional apparatus not shown in the
diagram but visible in the photographs are gears by means of which the
film is driven from the disc axle through a differential, which permits
the film to be framed up and down. The light beam is directed through
the film at right angles to the axis of the discs by means of two prisms,
whereby certain conveniences in driving and handling the film are
attained.

At the receiving end, the three sets of signals were supplied to the
three electrodes of a special neon lamp .V, shown in Fig. 5, which is
provided with a hydrogen valve to enable it to respond to the higher
frequencies. Condensing lenses L; and Ls image the three electrodes

Fig. 3—Sending end of three-channel television apparatus, showing sending prism
disc and photoelectric cells.

at the eye, where another lens L; is placed at the eye to focus the face
of the disc D.. By this system, nine electrode images are formed, of
which three are superposed at the eye, an successive scanning holes are
seen illuminated by each electrode in turn. This viewing arrangement,
by which the image is visible to only a single eye, is adequate for an
experimental investigation of the multi-channel method, but some other
scheme would of course be needed if the method were developed into a
practical form. Of several schemes, mention will be made here only

of the possible use of a triple grid of neon tubes, using a triple distrib-
utor of the type used in displaying images to a large audience in our
initial work in 1927.4

The three-channel apparatus, when all parts are properly function-
ing, yields results strictly in agreement with the theory underlying
its construction. The 13,500-element image, in resolving power and

amount of detail handled, is a marked advance over the single-channel
4,500-element image. Even so, the experience of running through a
collection of motion picture films of all types is disappointing, in that
the number of subjects rendered adequately by even this number of
image elements is small. ‘‘Close-ups’’ and scenes showing a great
deal of action, are reproduced with considerable satisfaction, but
scenes containing a number of full length figures, where the nature of
the story is such that facial expressions should be watched, are very

Fig. 5—Three-electrode neon lamp used for three-channel television reception.

far from satisfactory. On the whole, the general opinion expressed in
an earlier paragraph is borne out, that an enormously greater number
of elements is required for a television image for general news or
entertainment purposes. This, however, was anticipated, and the
real question is whether the results of this experiment indicate that
the finer grain image is best attained by resort to multi-channel means.

This leads to a discussion of what has proved to be a serious practical
difficulty with the multi-channel apparatus. This is the problem of
keeping the several channels properly related to each other in signal
strength. In the experimental apparatus, the direct current com-
ponents (introduced at the receiving end) and the alternating current
signals, are separately controlled, manually, by potentiometers.
These have fine enough steps so that with care, with a non-changing
image, a uniform picture may be obtained. If, however, for any
reason the signals on one of the channels becomes too strong or too
weak, the picture exhibits at once a strongly lined appearance. The
eye is quite sensitive to irregularity of this sort, and the transition
from a smooth grainless image to one showing a periodicity of 1/3 the
number of constituent lines largely offsets the higher resolving power
afforded by the actual number of scanning lines used. A characteristic
practical defect of the system as set up is that any marked change in the
general character of the signal, such as is produced by a shift from
close-up to a wide angle view may throw out the existing signal
balance sufficiently to show objectionable grain in the picture.

Differences of this sort in the three signals are of course caused in
general by differences in the characteristics of the three circuits. Such
differences can arise from overloading of amplifier tubes, whereby one
or more may be working on a non-linear portion; by rectifying action
of different amounts in the tubes immediately associated with the neon
lamps, or in the neon lamp electrodes themselves. A remedy is the
careful design and test of all parts of the system to insure the greatest
possible uniformity of performance. When this is carefully done, the
behavior of the three signals is reasonably satisfactory.

We are, as a consequence of this work, in a position to make a
general comparison of the two chief theoretical means for achieving a
television image of extreme fineness of grain, which are (1) extension
of the frequency band, and (2) the use of several relatively narrow
frequency bands. Both, because of the diminished amount of light
which finer image structure entails, demand enhanced sensitiveness of
the photo-sensitive elements at the sending end, and increased efficiency

fo the light sources at the receiving end. The multi-channel scheme
described has some advantage in compactness over the equivalent
single-channel apparatus, but since it is restricted to narrow angles of
illumination of the discs the overall efficiency of light utilization is not
essentially different. Comparing now the demands made upon the
electrical systems the differences between the two methods are clear
cut. Method (1) demands an extension of the frequency range of all
parts of the apparatus, the attainment of which depends upon physical
properties and technical devices whose mastery lies in the indefinite
future. Method (2) demands a multiplication of apparatus parts, and
careful design and construction of these parts so as to insure accurately
similar operation of a considerable number of electrical circuits and
terminal elements. The attainment of the necessary uniformity of
performance of the several electrical circuits and terminal elements,
while involving no fundamental problems, must present increasing
difficulty with the number of channels used.

Of the numerous microphones which have been developed since Bell's
original work on the telephone, only two are used extensively in sound
recording for motion pictures, namely, the condenser microphone and the
carbon microphone.

The condenser microphone was first proposed in 1881 but owing to its
low sensitivity was limited in its field of usefulness until the development
of suitable amplifiers. In 1917, E. C. Wente published an account of the
work which he had done on a condenser microphone having a stretched
diaphragm and a back plate so designed as to introduce an appreciable
amount of air damping. ‘The major portion of the condenser microphones
used today in sound recording embody the essential features of the Wente
microphone. Marked progress has, however, been made in the design and
construction of these instruments with the result that they are not only more
sensitive but also more stable. ‘The factors which contribute to this im-
provement are described in detail in this paper. Recently a number of
articles have appeared in the technical press calling attention to certain
discrepancies between the conditions under which the thermophone calibra-
tion of the condenser microphone is made and those which exist in the studio.
‘The nature of these discrepancies and their bearing on the use of the micro-
phone are discussed.

Microphones in which the sound pressure on the diaphragm produces
changes in the electrical resistance of a mass of carbon granules interposed
between two electrode surfaces have been used commercially since the
early days of the telephone. In recent years the faithfulness of the repro-
duction obtained with the carbon microphone has been materially improved
by the introduction of an air damped, stretched diaphragm and a push-pull
arrangement of two carbon elements. This instrument is finding extensive
use in sound recording and reproduction fields where carbon noise is not an
important factor. The outstanding design features of the push-pull carbon
microphone are described in this paper and suggestions made as to the
precautions to be taken in its use if the best quality, maximum life, etc.
are to be obtained.

Bell’s original work on the telephone, only two are used exten-
sively in sound recording for motion pictures, namely, the condenser
microphone and the carbon microphone. It has therefore been
suggested that it would be fitting to review at this time the con-
struction of these instruments and consider some of their trans-
mission characteristics and the precautions which should be exercised
in their use.

In 1881, A. E. Dolbear! proposed a telephone instrument which
could be used either as an electrostatic microphone or receiver. This

* Presented at Soc. of Motion Picture Engineers’ Convention, Oct. 20, 1930;
Journal, Soc. of Motion Picture Engineers, Jan., 1931.

instrument consisted of two plates insulated from one another and
clamped together at the periphery. The back plate was held in a
fixed position whereas the front was free to vibrate and served as a
diaphragm. It is obvious that, if the diaphragm were set in vibration
by sound pressure, the electrical capacitance between the two plates
would be changed in response to the sound waves, and if a source of
electrical potential were connected in series with the instrument a
charging current would flow which would be a fairly faithful copy of
the pressure due to the sound wave. Apparently Dolbear realized
that the current developed in this way would be minute, for in the
telephone system which he proposed as a substitute for the one using
Bell’s magnetic instruments he employed the electrostatic instrument
only as a receiver and adopted the loose contact type of microphone.
At approximately the same time an article appeared in the French
press ? calling attention to the use of a condenser as a microphone and
commenting on the fact that this type of microphone had been found
to be less sensitive than the loose contact type.

Owing to the low sensitivity of the condenser microphone, the field
of usefulness of this instrument was extremely limited for a number
of years and it did not assume a position of importance among the
instruments used in acoustic measurements and sound reproduction
until suitable amplifiers had been developed. The development of
the vacuum tube amplifier, however, filled this need. In 1917 E. C.
Wente ? published an account of the work which he had done on an
improved condenser microphone having a stretched diaphragm and a
back plate so located relative to the diaphragm that in addition to
serving as one plate of the condenser it added sufficient air damping
to reduce the effect of diaphragm resonance to a minimum.’ The
response of this instrument was sufficiently uniform over a wide range
of frequencies to make it not only useful in high quality sound repro-
duction but a valuable tool in acoustic measurements in general.

The major portion of the condenser microphones used today in
sound recording embody the essential features of the Wente micro-
phone. Marked progress has, however, been made in the design and
construction of these instruments since the initial disclosure and it
will no doubt be of interest to many to consider briefly the nature of
this advance.

3 **A Condenser Transmitter as a Uniformly Sensitive Instrument for the Absolute
Measurement of Sound Intensity,” E. C. Wente, Physical Review, July 1917, pp.
ee “Electrostatic Transmitter,” E. C. Wente, Physical Review, May 1922, pp.

_ *A discussion of the theory of air damping is given in “Theory of Vibrating
Systems and Sound,” I. B. Crandall, pp. 28-39.

In the early microphones employing air damping the diaphragm was
composed of a thin sheet of steel which was stretched to give it a
relatively high stiffness. When assembled in the microphone the
stiffness was further increased by that of the air film between diaphragm
and the damping plate with the result that the resonant frequency
was well above the frequencies which it was desired to transmit and
the diaphragm vibrated in its normal mode over a wide frequency
range. In such a structure the mechanical impedance for frequencies
below resonance is due almost entirely to stiffness reactance. Hencea
constant sound pressure produces substantially the same displacement
of the diaphragm at all frequencies within this range and uniform
response results except at the very low frequencies where an appreciable
reduction in the stiffness of the air film occurs. The effective mass of
a steel diaphragm is, however, relatively large and necessitates a
comparatively high stiffness to secure the desired resonant frequency.
From the standpoint of securing maximum sensitivity of the micro-
phone, i.e. displacement of the diaphragm per unit force, it is of course
important to make the stiffness as low as possible and employ as small
a value of mechanical resistance as is consistent with the degree of
damping required. An improvement in both respects can be effected
by decreasing the mass of the diaphragm for with a reduced mass a
given resonant frequency can be obtained with lower values of stiffness
and the desired damping constant secured with less mechanical
resistance.

The aluminum alloys have therefore replaced steel in the diaphragms
of most of the condenser microphones in use today. <A typical example
of such a microphone is the Western Electric Company’s instrument
(394-type) shown in the photograph, Fig. 1, and the cross-sectional
view, Fig. 2. The diaphragm of this instrument is made from alu-
minum alloy sheet .0011 inch in thickness. The edges are clamped
securely between threaded rings, gaskets of softer aluminum being
provided to prevent damage at the clamping surfaces. The requisite
stiffness is obtained by advancing the stretching ring until a resonant
frequency of 5,000 cycles is obtained. The method of determining
the resonant frequency of the diaphragm is as follows. The diaphragm
assembly to be tested is coupled to a condenser microphone which is
provided with a suitable circuit for measuring its output. A special
telephone receiver is placed in contact with the diaphragm on the
side opposite to the coupler. Current from a vacuum tube oscillator
is then passed through the winding of the receiver, setting up eddy
currents in the diaphragm under test. The forces which are developed
as a result of the reaction of the magnetic field produced by the eddy

currents and that of the permanent magnet of the receiver set the
test diaphragm in motion. The resonant frequency is determined by
noting the frequency at which the output from the condenser micro-
phone is a maximum.

In the early Wente microphone the damping plate was a continuous
surface. Subsequent work by I. B. Crandall °® showed that the re-
quired amount of damping at the resonant frequency could be obtained
without adding unduly to the impedance at other frequencies by cut-
ting grooves in the plate. ‘This reduced the stiffness introduced by the
air film and decreased the irregularity in response at low frequencies
previously mentioned. The grooves in the damping plate of the

Western Electric Company's 394-type microphone are cut at right
angles. Holes, tapered at the outer end to reduce resonant effects,
are bored through the plate at the intersection of the grooves to form
connecting passages between the air film at the front and the cavity
at the back. In order to prevent the resonance which would result
if the grooves extended into the portion of the chamber surrounding
the damping plate, the outer ends are closed by an annular ring which
is pressed over a shoulder on the plate. The surface of the damping
plate is plane within 8 X 10-° inch. The departure from a plane in
any individual case is determined commercially by the interference
pattern developed when an optically flat plate is placed over the
damping plate under test.

_ §“The Air Damped Vibratory System: Theoretical Calibration of the Condenser
lransmitter,” 1. B. Crandall, Physical Review, June 1918, pp. 449-460.

A duralumin spacing ring .001 inch in thickness separates the
damping plate from the diaphragm. It is essential that all dust and
dirt be excluded from this space. To prevent foreign material from
entering through the holes in the plate a piece of silk is fastened over
the outer surface. The assembly of the diaphragm and the damping
plate is made in a dust-proof glass cabinet.

If the back wall of the condenser microphone were rigid, changesdn
the separation between the damping plate and the diaphragm of
sufficient magnitude to affect not only the sensitivity of the instrument
but also its frequency response characteristic would result from vari-
ations in barometric pressure. Complete compensation for these

changes in pressure can only be obtained by permitting free inter-
change of air between both sides of the microphone diaphragm.
This is, however, objectionable owing to the fact that sufficient
moisture is likely to be introduced to start corrosion and affect the
insulation between the damping plate and the diaphragm. A com-
pensating diaphragm of organic material has therefore been introduced
which prevents this undesirable effect of humidity but is sufficiently
low in stiffness to equalize the changes in pressure encountered in the
normal use of the microphone.

In order to prevent transmission losses at voice frequencies due to
the presence of the compensating diaphragm, an acoustic valve is

inserted between the damping plate and this diaphragm. This valve
consists of a disc of silk clamped between two aluminum plates of
unequal diameters. Gas in passing from the damping plate to the
compensating diaphragm moves laterally from the edge of the smaller
plate through the silk to a hole in the center of the larger plate. The
impedance of this path is high at voice frequencies but low enough for
steadily applied pressure differences to permit compensation for changes
in barometric pressure.

After the damping plate and diaphragm are assembled the space
between the clamping rings is filled with beeswax to make the joints
gas-tight and exclude moisture. A hole is, however, provided for
filling the microphone with nitrogen. The purpose of the nitrogen is
to prevent corrosion of the damping plate and diaphragm surfaces
and eliminate any reduction in pressure due to oxidation of the sealing
compound.

It has been customary for some time to determine the response
characteristics of a condenser microphone by the thermophone
method. In making this measurement the diaphragm of the micro-
phone is coupled acoustically to the thermophone in the manner
shown in Fig. 3. The thermophone consists of two strips of gold foil

Fig. 3—Cross-sectional view of the thermophone and the condenser microphone.

which are mounted on a plate and fit into the recess in the front of
the microphone. Capillary tubes are provided for filling the space
enclosed between the plate and the microphone diaphragm with

°“The Thermophone as a Precision Source of Sound,” H. D. Arnold and I. B.
Crandall, Physical Review, July 1917, pp. 22-38. “The Thermophone,” E. C.
Wente, Physical Review, April 1922, pp. 333-345. ‘Speech and Hearing,” H.
Fletcher, 1929, Appendix A.

hydrogen. This is done in order to make the wave-length» . che sound
developed in the recess as large as possible compared with dimensions
of the chamber. If this were not the case the sound pressure at dif-
ferent positions in the chamber would not be in phase and the condi-
tions on which the computations of the magnitude of the sound
pressure are based would not be met. A direct current of known
value is passed through the foil. Superimposed upon the direct current
is an alternating current of the desired frequency which causes fluctu-
ations in the temperature of the foil and in the gas immediately
surrounding it. These fluctuations in temperature in turn cause
changes in the pressure on the microphone diaphragm. The magni-
tude of the pressure developed on the diaphragm can be computed
from the constants of the thermophone and the coupling cavity, and
the voltage developed by the microphone for a given pressure deter-
mined with suitable measuring circuits.? Obviously, such a calibration
affords a measure of the response of the microphone in terms of the
actual pressure developed on the diaphragm and is independent of the
external dimensions of the instrument. Hence, it does not take into
account any effect which the microphone may have on the sound field
when used as a pick-up instrument for recording or broadcasting pur-
poses. The thermophone calibration is often referred to as a ‘‘pres-
sure”’ calibration and the response obtained by placing the instrument
in a sound field of constant pressure, a “field” calibration. A thermo-
phone calibration of a representative Western Electric 394-type con-
denser microphone is shown on Fig. 4.

For many of the uses to which the condenser microphone is put, for
example the calibration of head type telephone receivers, the condi-
tions under which it operates agree with those under which the thermo-
phone calibration is made. There are, however, cases where this
agreement does not exist, for when a microphone is inserted in a
sound field of uniform intensity the pressure on the diaphragm may
depart rather widely from a constant value in certain frequency
ranges. Several articles * have recently appeared calling attention to
this discrepancy between the pressure and field calibrations and
pointing out that a pressure calibration of a microphone may not be
entirely representative of its performance under the conditions which
exist in a studio.

7“ Master Reference System for Telephone Transmission,’’ W. H. Martin and
C. H. G. Gray, Bell System Technical Journal, July 1929, pp. 556-559.

’“*The Use of a Wente Condenser Transmitter to Measure Sound Pressures in
Absolute Terms,”’ A. J. Aldridge, P. O. E. E. Journal, Oct. 1928, pp. 223-225.
‘Effect of the Diffraction Around the Microphone in Sound Measurements,”’ S. Bal-
lantine, Physical Review, Dec. 1928, pp. 988-992. ‘‘Measurements of Sound
Pressure on an Obstacle,’’ W. West, Just. Elec. Eng. Journal, 1929, pp. 1137-1142.

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The di.. ence between the pressure and field calibrations is due to
several factors. In the first place the sound is diffracted around the
microphone differently at different frequencies. At frequencies where
the wave-length is large as compared with its external dimensions the
pressure is the same as that of the undisturbed wave. At the higher
frequencies where the microphone is large in comparison with the wave-
length of the sound, the pressure is twice that developed at the lower
frequencies. In the 394-type microphone the effect of diffraction

FREQUENCY IN CYCLES PER SECOND
Fig. 4—Pressure calibration of the 394 type condenser microphone.

first becomes noticeable in the region of 1200 cycles and reaches a
maximum of 6 db at approximately 2200 cycles. The second factor
which causes a difference between the pressure and field calibrations is
acoustic resonance in the shallow cavity in front of the microphone.
This causes the pressure actuating the diaphragm to be higher than
that of the incident sound wave in the frequency region of 1500 to
5500 cycles. The maximum increase in pressure occurs at approx-
imately 3500 cycles. If the sound source is so located relative to the

Fig. 5—Field calibration of the 394-type condenser microphone for a direction o
approach of sound normal to the diaphragm.

microphone that the waves approach from a direction normal to the
diaphragm and reflection from surrounding walls and objects is
negligible, the combined effect of diffraction and resonance is to
produce a maximum departure from flatness of approximately 12 db
as is shown by the field calibration Fig. 5." If the sound wave travels

® These curves are taken from unpublished work of P. B. Flanders of the Bell
Telephone Laboratories, Inc.

along the diaphragm the effective pressure is reduced at the higher
frequencies due to difference in phase. Hence, if the direction of
approach of the sound wave is parallel to the plane of the diaphragm,
the departure from flatness is materially reduced. This is brought
out quite clearly by the field calibration for sound approaching from
a direction parallel to the diaphragm, Fig. 6.°

The discrepancy between the pressure and field calibrations of the
condenser microphone involves two important assumptions, namely,
a plane sound wave and no reflection from walls or surrounding objects.
When the microphone is used in a studio much of the sound reaches
the diaphragm by way of reflection from the walls of the room. The
requirement of no reflection is therefore not met and the influence of
the acoustic properties of the reflecting surfaces is added to the char-
acteristics of the microphone. The effect of the diffusion of the

Fig. 6—Field calibration of the 394-type condenser microphone for a direction of
approach of sound parallel to the diaphragm.

sound field and the tendency for most materials to be more absorbent
for sounds of high frequency appears to cause the response under
studio conditions to be more nearly like that obtained when the sound
approaches in a direction parallel to the diaphragm and make the
departures from the pressure calibration less marked than the field
calibration for a direction normal to the diaphragm would indicate.
This perhaps accounts in part at least for the instances in which a
corrective network designed to compensate for the field calibration
normal to the diaphragm failed to effect a material improvement in
quality.

The acoustic conditions under which a microphone is used cover a
wide range. It would therefore be difficult if not impossible to adopt
a set of conditions for use in connection with a field calibration of the
condenser microphone, which would be known to be representative
of those encountered in practice. The pressure method of calibration

on the other hand is definite, simple, and capable of being accurately
duplicated in different laboratories. In view of this situation it would
seem advisable to retain, at least for the present, the thermophone or
pressure method of calibration for general use. In cases where precise
quantitative measurements are required a field calibration of the
microphone should of course be secured under the conditions of actual
use. Various methods of making such a calibration have been pro-
posed. The Rayleigh disc has been used extensively in this work
thus far but there are certain very definite limitations to the extent
to which it can be applied. An interesting discussion of. the use of
the Rayleigh disc may be found in papers by E. J. Barnes and W.
West,’ and L. J. Sivian."

It would seem reasonable to expect that future design work would
be directed toward reducing transition, resonance and phase difference
effects toa minimum. The results of work along this line have been
reported by S. Ballantine ' and D. A. Oliver.’* In both instances
the mechanical design is such that the resonant cavity in front of the
diaphragm is eliminated and the housing is spherical or streamline
to reduce the diffraction effect. There has as yet been little oppor-
tunity to determine the extent of the practical improvement effected
by these changes in design and the whole discussion continues to be
somewhat academic in character.

Bell’s original microphone was essentially a generator and hence
was limited in its output to the maximum speech power available at
its diaphragm. The demand for telephonic communication over
longer distances led to the early introduction of a carbon microphone.
In this instrument the resistance of the carbon element is caused to
vary in response to the sound pressure on the diaphragm and produces
changes in the current supplied from an external source of electrical
potential, which are fairly faithful copies of the pressure changes which
constitute the sound wave. The carbon microphone is therefore in
general an amplifier in which a local source of power is controlled by
the acoustic power of the sound wave.

The carbon element or ‘“‘button” of the first microphones (Edison,
1877) was made from plumbago compressed into cylindrical form.

“The Calibration and Performance of the Rayleigh Disc,” E. J. Barnes and
W. West, Inst. of Elec. Eng. Journal, 1927, Vol. 65, pp. 871-880.

“Rayleigh Disc Method for Measuring Sound Intensities,” L. J. Sivian,
Philosophical Magazine, March 1928, pp. 615-620.

2 Contributions from the Radio Frequency Laboratories No. 18, 5. Ballantine,
April 15, 1930.

8“ An Improved Microphone for Sound Pressure Measurements,” D, A. Oliver,
Journal of Scientific Instruments, April, pp. 113-119,

This type of button was relatively insensitive and shortly after its
introduction the suggestion (Hunnings, 1878) was made that the
space between the diaphragm and the fixed electrode be “partially
filled with pulverized engine coke,” ' in order to increase the number
of contact points and render them more susceptible to the forces
developed by the motion of the diaphragm. When at its best the
Hunnings transmitter was fairly efficient but at times was erratic in
its performance due in part to the nature of the microphonic material.
In 1886 Edison '° proposed the use of granules of hard coal which had
been heat treated. ‘This was an important advance, for carbon made
from anthracite coal is used not only in the microphones which are
being considered in this paper but in commercial telephone trans-
mitters as well.

As in the case of the condenser microphone, the displacement of the
diaphragm of the carbon microphone must be substantially constant
at all frequencies if uniform response is to be obtained. In the early

rather prominent irregularities in response. Air damped stretched
diaphragms offered one solution of this problem. During the World
War instruments of this type were developed and applied to the
problem of locating airplanes. In 1921 double button stretched
diaphragm microphones were made available for use with the public
address equipment installed for the inaugural address of President
Harding and the excercises at Arlington on Armistice Day.'® The
carbon microphones employed in sound picture recording are of the
stretched diaphragm double button type. The electrical output
from this type of microphone is not only of substantially uniform
intensity over a wide frequency range but due to the ‘“‘push-pull”’
arrangement of the buttons is comparatively free from harmonics.
A typical example of the present day carbon microphone is shown in
the photograph, Fig. 7. Fig. 8 is a cross-sectional view of the same
type of microphone.

The diaphragm is made from duralumin .0017 inch in thickness and
is clamped securely at its outer edge. The clamping surfaces are
corrugated and emery cloth gaskets are provided to prevent slipping.
The stretching of the diaphragm is done in two steps. The initial
stretching ring is first advanced by means of six equally spaced screws
until the diaphragm is smooth and free from irregularities. The inner
or final stretching ring is then adjusted to a position which gives the

: diaphragm a resonant frequency of 5700 cycles per second. The
method employed in making the determination of the resonant
frequency is substantially the same as that used in connection with the
assembly of the condenser microphone, with the exception that the

A spacing washer .001 inch in thickness separates the diaphragm
from the damping plate. A single concentric groove is provided in
the damping plate.

The buttons are of the conventional cylindrical type but are provided
with a novel form of closure to prevent carbon leakage at the point
where they make contact with the diaphragm. The closure consists
of twenty-seven rings of .0004 inch paper clamped firmly together at
the outer edge and spreading apart at the inner edge to form a structure
which effectively seals the junction between the diaphragm and the
buttons without adding materially to the mechanical impedance.

As has already been pointed out the granular carbon is made from
selected anthracite coal. The size of the granules is such that they
will pass through a screen having 60 meshes per inch but will be re-
tained on a screen having 80 meshes per inch. Before heat treatment
the raw material is treated with hydrofluoric and hydrochloric acids
to reduce the ash content. Each button contains .060 cc. of carbon,
i.e., about 3000 granules.

The bridge which supports the button on the front of the diaphragm
partially closes the acoustic cavity on that side. It is essential,
therefore, that it be so proportioned as to have a minimum reaction
on the response of the microphone and yet provide the required degree
of rigidity. It was this consideration that led to the smooth stream
line contour now employed.

FREQUENCY IN CYCLES PER SECOND
Fig. 9—Pressure calibration of the 387-type carbon microphone.

Referring to Fig. 9 it will be observed that the adoption of an air
damped stretched duralumin diaphragm for the carbon microphone
has resulted in an instrument having a substantially uniform response
over a wide range of frequencies. The arrangement of the apparatus
employed in securing the data from which this curve was plotted is
shown in the photograph, Fig. 10. The microphone under test was
mounted in a highly damped room at a distance of six to eight feet
from a source of sound which consisted of two loud speaking receivers,

One of the receivers was the conventional form of moving coil direct
radiator and was used to provide sound in the lower frequency range.
The other was a special moving coil receiver with a short horn so
designed as to serve as an efficient source of sound up to 10,000
cycles.!7 To reduce the effect of standing waves the mounting for the
receivers was so constructed that they could be rotated through a
circle approximately five feet in diameter and always face the micro-
phone under test. Before starting the test of the carbon microphone
the receivers were calibrated by placing a calibrated condenser micro-

phone at the point where the test instrument was to be located and
determining the receiver current required to produce a pressure of one
bar (one dyne per square centimeter) on the microphone diaphragm.
The condenser microphone was then removed and the test microphone
substituted. The open circuit voltage developed by the microphone
when supplied with a direct current of .025 ampere per button was then
measured. The data obtained in this way are essentially a ‘“‘ pressure

calibration” of the microphone and in interpreting them in terms of
“field”’ performance the same factors must be taken into account

_ “An Efficient Loud Speaker at the Higher Audible Frequencies,” L. G. Bost-
wick, Journal of the Acoustical Society, Oct. 1930, pp. 242-250.

which have been discussed in considerable detail in connection with
the condenser microphone.

The circuit employed in measuring the response of the carbon micro-
phone is shown on Fig. 11. Two steps are involved in the calibration
of the sound source. With the output terminals of the microphone

' circuit and the sound source short circuited and the polarizing voltage

for the condenser microphone removed, the attenuator is adjusted
until the voltage applied to the measuring circuit is that developed by
the condenser microphone when a sound pressure of one bar is im-
pressed on its diaphragm. A record is made of the reading of the

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output meter in the measuring circuit. The polarizing voltage is
then applied to the condenser microphone. After the output terminals
of the attenuator have been short circuited an alternating current of
a known frequency is supplied to the sound source and the magnitude
of this current adjusted until the meter reading is the same as that
previously obtained with the attenuator. This completes the cali-
bration of the sound source for that frequency. After the carbon
microphone has been placed in the position previously occupied by
the condenser microphone, the polarizing voltage is once more removed
from the condenser microphone and the output from the carbon
microphone circuit impressed on the measuring circuit. The reading

of the output meter is recorded. The sound source and carbon
microphone circuit are then short circuited and the output from the
attenuator again applied to the measuring circuit. The attenuator
is adjusted until the reading of the output meter is the same as was
previously obtained with the carbon microphone in circuit. In this
way the voltage applied to the measuring circuit when the carbon
microphone is in operation is determined. The open circuit voltage
developed by the carbon microphone may then be computed from the
voltage and the constants of the microphone circuit. At the locations
where these measurements were made a certain amount of interference
from 60-cycle circuits and low frequency acoustic disturbances was
encountered. ‘The high-pass filter in the measuring circuit was intro-
duced to facilitate the measurements under these conditions. The
adjustable low-pass filter was used to confine the measurements to
the fundamental frequency. Only that portion of the apparatus to
the left of the dotted line was mounted in the damped room.

The two buttons of the carbon microphone are identical in their
dimensions and if the granular carbon is in the same mechanical state
have substantially the same electrical characteristics. They are also
practically free from the cyclic variations in resistance known as
‘‘breathing’’ which result from the temperature changes caused by
the power dissipated in the granular carbon. It is, however, a matter
of every day experience that a given mass of granular material will
occupy different volumes, depending upon the configuration of the
particles. In the case of microphone carbon this change in configura-
tion of the granules results in changes in the contact forces of sufficient
magnitude to affect the resistance and sensitivity. If these changes
occur in unequal amounts in the buttons electrical unbalance results.
When complete balance exists the electrical output is free from all
harmonics introduced by the circuit. Hence, in using the microphone
care should be taken to see that a fair degree of balance between the
buttons is maintained.

The performance of a carbon microphone may be affected adversely
by cohering of the granules. Severe cohering is accompanied by a
serious reduction in resistance and sensitivity which persists for an
extended period unless the instrument is tapped or agitated mechan-
ically. One of the common causes of cohering is breaking the circuit
when current is flowing through the microphone. Experiment has
shown that the insertion of a simple filter consisting of two .02 mf.
condensers and three coupled retardation coils each having a self-
inductance .0014 henry, will effectively protect the microphone button
from cohering influences without introducing an appreciable trans-

mission loss. This filter may be located in the base of the mounting
or in a container fastened to the back of the microphone.

Aging of granular carbon may result from changes in the contact
surface caused either by mechanical abrasion or overheating due to
excessive contact potentials. Aging is usually accompanied by an
iicrease in resistance and loss in sensitivity. Care should therefore
be exercised in the use of the carbon microphone that it is not sub-
jected to unnecessary vibration which would cause the granules to
move relative to one another and abrade the surfaces. The use of
abnormally high voltages should also be avoided.

The quality of transmission obtained with the double button carbon
microphone compares favorably with that secured with a condenser
microphone. ‘The carbon microphone also requires less amplification.
There is, however, one characteristic which limits its use, namely
carbon noise. The level of the noise is much higher than that due to
thermal agitation within the carbon granules '* and appears to be
caused by heating at the contacts between the granules. A certain
amount of gas is contained in the pores in the contact surfaces. When
current passes through the button, a sufficient increase in contact
temperature takes place to cause a portion of this gas to be driven off
and produce the non-periodic changes in resistance which give rise
to carbon noise.

In conclusion it may be stated that the condenser and carbon types
of microphones have been developed to a point where there is little to
choose between them from the standpoint of quality of transmission.
The design from a mechanical standpoint has also been carried to a
point where little difficulty should be experienced intheir use if reason-
able precautions are exercised. Although requiring less amplification
than the condenser microphone the extent to which the carbon micro-
phone is used at present is limited by the higher noise level obtained.
- The condenser type of microphone has therefore been adopted for
most of the recording work in the sound picture field.

18*Thermal Agitation of Electricity in Conductors,”’ J. B. Johnson, Physical
Review, July 1928, pp. 97-109.

Certain Factors Affecting the Gain of Directive Antennas *
By G. C. SOUTHWORTH

This paper analyzes the performance of antenna arrays as influenced by
certain variables within the control of the designing engineer. It starts with
an extremely simple analysis of the interfering effects produced by two
sources of waves of the same amplitude. This is followed by a short dis-
cussion of a paper by Ronald Foster, which considers two antennas and also
16 antennas when arranged in linear array. Two antennas separated in
space by 14 wave-length and in phase by '4 period give sensibly more
radiation in one direction than in the opposite. This, for convenience, has
been called a unidirectional couplet. A number of these couplets may be
arranged in linear array, thereby giving an extremely useful directive
system. Diagrams are shown for such arrays as affected by the number and
spacings of the individual couplets. The gains from such arrays are
calculated and data are given showing fair agreement between calculation
and observation.

Directional diagrams for arrays of coaxial antennas indicate that some-
what less gain may be expected from this form than when the elements are
spaced laterally. Combinations of these two types of arrays give marked
directional properties in both their horizontal and vertical planes of refer-
ence, This principle has been used rather generally in short-wave com-
munication. This paper also discusses effects resulting from combining
two or more arrays. In one case the space between two arrays tends to
emphasize spurious lobes. The directional diagram of such a combination
may be rotated within limits by changing the phasing between adjacent
arrays or sections of an array. In all of the above cases the influence of
the earth is ignored.

A mathematical appendix gives general equations for calculating di-
rectional diagrams of linear arrays. Special cases of these equations apply
to the figures included in the main part of the text. General equations are
also given for calculating the gains of arrays. Similar equations permit the
areas of diagrams to be calculated. An extended bibliography on antenna
arrays is appended.

engineer has aspired to a directive system whereby radiation
might be projected from one point to another with a maximum
of efficiency and a minimum of interference with adjacent stations.
Also, he has aimed at similar directivity at the receiver to improve
the signal-to-noise ratio and otherwise discriminate against un-
desirable signals. It was recognized at a very early date that directive
radio based on wave interference was feasible provided sufficiently
short waves could be utilized, and as a result many interesting sug-
gestions to this end were made. However, as is well known, the early
development of the radio spectrum proceeded in the direction of long

waves rather than short waves, thereby deferring many of the applica-
tions of these suggestions.

The principle of wave interference on which most short-wave
systems of directive radio are based has probably been known for
several centuries. However, the first thorough treatment of this
subject was by Sir Thomas Young,' who, together with Fresnel,
securely established the wave theory of light in the early part of the
last century. Even Hooke and Huygens, who had offered the wave
theory over a century earlier, failed to recognize the full significance
of interference.

When Hertz started his celebrated experiments to verify Maxwell's
theory he was, of course, in full knowledge of these phenomena and
their explanation, and invoked their use in proving the existence of
electric waves. It is interesting that in some of his experiments he
made use of parabolic mirrors for both transmitting and receiving,
having directional characteristics very similar to those sometimes
used in present day radio practice. It is also of interest that he found
that parallel wires stretched over a frame were quite as effective as a
reflector as a continuous sheet of metal of similar dimensions, pro-
vided the wires were kept parallel to the lines of electric force of the
arriving wave. He apparently did not investigate the effect of varying
the spacing nor the length of the parallel wires, nor did his subsequent
experiments otherwise tend toward the present day antenna array
technique.

This paper treats in an elementary way certain aspects of the
antenna array problem, principally as regards the manner in which
calculated directivity is affected by the number and spacing of the
individual antennas which go to make up the array. The theory is
applicable only to those forms of directive antennas which may be
resolved into a series of individual sources. It does not apply to the
so-called wave antenna. However, principles are included which have
for some time been in general use in. combining two or more such
antennas.

Extensive study has been given to directive antenna systems for
use in transoceanic radiotelephony. Papers dealing with this general
subject have appeared from time to time. Further work is in prog-
ress. Papers by E. J. Sterba and also by E. E. Bruce and H. T. Friis of
the Bell Telephone Laboratories are in preparation which will include

2R. M. Foster, ‘Directive diagrams of antenna arrays,”’ Bell Sys. Tech. Jour.,
292, May, 1926. Austin Bailey, S. W. Dean, and W. T. Wintringham, “‘ Receiving
system for long-wave transatlantic radiotelephony, Proc. J. R. E., 16, 1694, December,
1928. J.C. Schelleng, ‘‘Some problems in short-wave radiotelephone transmission,”
Proc. I. R. E., 18, 913; June, 1930.

GAIN OF DIRECTIVE ANTENNAS 65
certain calculated data similar to those contained in the present paper,
and also experimental results obtained from tests on actual antennas of
various sizes and proportions.

In the early part of the following discussion each antenna is con-
sidered as a spherical source of waves which radiates equal power in
all directions. Furthermore, it assumes that the current in each

Fig. 1—Interference pattern. ‘Two equiphased sources spaced one-half wave-length.

individual source, in a given array, is the same and is not materially
affected in either magnitude or phase by its proximity to other sources.
The fair approximation to which these calculated results are realized
in practice bespeaks the justification of these assumptions.

The various steps by which present day directional radio has been
developed are extremely interesting, but they are so involved in the
development of radio itself that their enumeration is considered out-

side the scope of this paper. However, bibliographies are cited below
covering some of their important phases.

The interference patterns resulting from a number of individual
sources of waves, such as antennas, are dependent on both their
spacial arrangement and the magnitudes and relative phases of their
forces. This makes possible an almost unlimited number of com-
binations of which only a portion have thus far found use in com-

Fig. 2—Interference pattern. Two sources separated in space by one-fourth wave-
length and in time by one-fourth period.

munication. This paper will restrict itself mainly to some cases which
are already finding general application. As a suitable introduction
to this subject, a very simple case of wave interference is discussed in
the following paragraph.

Figs. la and 2a depict in a rough way the interference resulting
from two independent sources of spherical waves of the same ampli-
tude. In the first case they are spaced 14 wave-length but are assumed
to be oscillating in phase. In the second case the two sources are sepa-
rated in space by 14 wave-length and in phase by 14 period. Crests

GAIN OF DIRECTIVE ANTENNAS 67
and troughs are represented respectively by solid and dotted lines.
At points where either two crests or two troughs arrive simultaneously
the resultant wave is greatly enhanced, whereas at certain other points
crests and troughs arrive together, thereby neutralizing each other's
effects. At certain intermediate points these interfering effects are
only partially complete. Accompanying each figure is a directive
diagram (1b and 2b), plotted in polar coordinates, which shows the
effectiveness of the wave in each direction. The circle drawn outside
each diagram indicates the effect if the radiation had proceeded trom
a single non-directional source similar to each of the above. The
ratio between the areas of the circle and the inscribed diagram gives
roughly the power improvement of such a device as manifested in the
intensity of the radiated wave. A more exact calculation of this
improvement requires an integration of the force components over a

Most directive antenna systems now in general use for short
waves may be regarded as special applications of the linear array.
This type consists of two or more antennas all having currents of equal
amplitude, equispaced along the same straight line. The properties
of such arrays have been treated very generally by Foster,? whose
paper included several hundred directive diagrams, taken in a bi-
secting plane perpendicular to the axis of each antenna of the array,
and typical of the results which may be expected from two antennas
and from arrays consisting of 16 antennas. A portion of these dia-
grams have been reproduced in Figs. 3 and 4 below. The same
principles are applicable to both transmission and reception.

In Fig. 3 are shown diagrams resulting from two antennas as the
separation is increased from 0 to 1 wave-length in steps of 14 wave-
length and the phase increased from 0 to '% period in steps of 14
period. The line or axis of the array is assumed to be horizontal and
the specified phase difference is such that the current in the right-
hand antenna is lagging for a transmitting system and leading for a
receiver. It will be noted that for phase differences of both 0 and 447
the diagrams are symmetrical about both the horizontal and vertical
axes of the figure, whereas for other phases the figures are asymmetrical
about the vertical axis except for certain limiting cases. Of these
asymmetrical diagrams, that corresponding to phase and spacial sepa-
rations both of 14 (Fig. 3b) is of particular importance and forms
the basis of the so-called reflector effect. This particular combination
of two sources is referred to later as a unidirectional couplet. In

3In this, and in other cases in this paper, radiation is referred to as unidirectional
when sensibly more power is propagated in one direction than in others.

passing it is also of interest to note that the diagram of the coil or
frame aerial as generally used is intermediate between Figs. 3c and
3d. Its diagram would not differ essentially from its neighbors, Figs.
3d, 3e, or 3f, except for scale. This scale may conveniently be regarded
as a measure of the impedance of the device, or possibly its radiation
efficiency, but not necessarily a measure of its usefulness.

Fig. 4 shows similar diagrams resulting from 16 antennas for vari-
ous phase and space relations. As in Fig. 3, diagrams in the top and
bottom rows corresponding respectively to phases of 0 and 1% T are
symmetrical about both the horizontal and vertical axes. The dia-
grams in the top row are in general bidirectional, while the bottom row
has one bidirectional diagram corresponding to phase and space
differences both equal to 1%. It is of interest that for the most part
cases where the phase and space separations are numerically equal
correspond to unidirectional diagrams. However, these diagrams are
only moderately sharp and thus far such arrays have not been used
extensively in practice.

Referring again to the diagrams in the top row corresponding to 16
antennas all driven in phase, we note that directivity becomes progres-
sively sharper as the spacing is increased until in the vicinity of 15/16X
appendages develop which soon surpass in magnitude the desired lobes.
This effect is present in the commercial array, and limits, as we shall
later see, the gain that may be derived from a given number of elements.
The diagrams shown in Fig. 4 for 16 antennas are typical of others
where the number of antennas in linear array is fairly large.

One type of array now in commercial use consists of two parallel
linear arrays of equiphased elements where the two parallel arrays are
spaced 14 wave-length and differ in relative phase by 14 period. It is
convenient to regard such a device either as two independent linear
arrays, each having a directional characteristic as shown in the top
row of Fig. 4, or as an array of couplets, each couplet of which has by
itself a heart-shaped characteristic. Both antennas of the couplet may
be independently driven at their prescribed phase separation of 14
period, or one may derive its power from that radiated by the other, in
which case the proper phase relation is automatically approximated *
and the same practical result is obtained. In the latter case one is

‘The problem of the reflecting antenna has been considered by Wilmotte and
McPetrie, Jour. I. E. E., 66, 949, Englund and Crawford, Proc. I. R. E., 17, 1277;
August, 1928, and Palmer and Honeyball, Jour. I. E. E., 67, 1045. Their conclusions
indicate that the optimum separation between a single antenna and its reflector to
give maximum forward radiation is roughly \/3. However, it appears that when
several antennas and reflectors are involved a separation more nearly \/4 is optimum.

frequently known as the driven antenna and the other the ref’
This viewpoint is perhaps only a convenience and ma
together correct. An array of the above type transmits). . ves
best in a direction at right angles to its principal dimeusion. This
type is, therefore, frequently known as a broadside array.

In Fig. 5 is plotted a series of diagrams in a bisecting plane normal
to the axis of each antenna of the array for different broadside arrange-
ments such as are used commercially. They are systematically ar-
ranged horizontally in the order of the number of couplets in the array,
and vertically with the increased spacing between adjacent couplets.

Several different forms of such directive diagrams are possible,
which may be plotted in either polar or rectangular coordinates. In
one form all diagrams are roughly of constant area and relative gains
from various antenna systems are expressed in terms of the principal
radius vector. In the second form the length of the principal radius
vector remains constant and the relative gain is roughly inversely
proportional to the area of the diagram. The second of these forms has
been adopted in this paper largely because of the relative simplicity of
the equation of the diagram and the facility with which properties of
antennas may be determined.

In the lower left-hand corner of Fig. 5 will be found a plan showing
the arrangement of the elements relative to the important direction of
transmission. At its right is the general equation of these diagrams.
This formula is also given as equation (14) of the appendix where the
analytical theory of arrays is developed. Below each diagram is the
ratio of the area of the circumscribed unit circle to the area of the hori-
zontal diagram. Here also will be found the ratio of the area of the
subordinate loops to the area of the main loop. The total area may be
measured approximately with a planimeter or calculated more accu-
rately by equation (32) in the mathematical appendix. In making up
Fig. 5 each diagram was accurately plotted on standard polar coordin-
ate paper from perhaps a hundred calculated points. This was then
reduced photographically and the several diagrams were assembled.*

Inspection of the diagrams shows that increasing the number of
couplets increases in all cases the sharpness of the main loop and
hence the gain of the array. However, increasing the separation be-

5 The diagrams used in this paper were calculated by a group of the Department
of Development and Research of the American Telephone and Telegraph Company,
under the direction of Miss E. M. Baldwin. Most of the material was checked by

Mrs. Isabel Bemis, who assembled it in its present form and prepared the attached
bibliography.

Fig. 4—Directive amplitude diagrams for an array of sixteen antennas. Separation in wave-lengths (A) along the top. Phase difference in pe

S = RATIO OF AREA OF UNIT CIRCLE TO THAT OF DIRECTIONAL DIAGRAM
R = RATIO OF AREA OF SUBORDINATE LOOPS TO THAT OF MAIN LOOP
A™FRACTION OF WAVE LENGTH SPACING BETWEEN ELEMENTS

tween couplets increases the gain only up to a certain point, after
which the formation of parasitic lobes decreases the effectiveness of the
array. The trend of these gains may be illustrated more effectively in
graphical form.

In Fig. 6 calculated gain ratio is plotted against number of couplets
giving one graph for each separation considered. These ratios are not
based on the data given in Fig. 5, but were obtained from the integra-
tion of the equation of the directional diagram over an arbitrary
sphere by use of equation (27) below. It may be noted that for many
conditions the difference between these methods of calculating gain is
only moderate. These power ratios are for the most part linear,

indicating that such gains are proportional to the length of the array.
This is in keeping with the view that a receiving antenna can intercept
wave power more or less in proportion to its dimensions. It is also
interesting to note that the slope of the curve of \/2 is approximately
twice that for 4/4, so that 16 couplets spaced 14 wave-length give
approximately the same gain as eight couplets spaced 14 wave-length.
This again shows that the length of the array is the most important
criterion in determining its gain. In Fig. 7 the same data have been
plotted in decibels.

In Fig. 8 gains expressed in decibels are plotted against the separa-
tion between elements. This shows more definitely the trend of the

antenna gain to a maximum, after which spurious lobes become of
importance. Fig. 8 suggests that the spacing, giving optimum gain,
would be the desideratum in antenna design. However, this is not

Fig. 8—Antenna arrays. Calculated gains vs. lateral spacing between couplets.

necessarily the case, as we shall presently see. It has already been
pointed out that the over-all length of array, rather than the spacing
or the number of conductors per unit length, constitutes the most

important factor in determining the gain. Furthermore, minimum
area diagrams are frequently attended by fairly large spurious lobes
which are undesirable particularly on receiving antennas. Also the

8 ie) i2 14 16 18 20
LENGTH OF ARRAY IN WAVELENGTHS
Fig. 9—Approximate gains to be expected from arrays of couplets for spacings of
approximately A/4 and 4/2.

Fig. 10—Approximate gains to be expected from arrays of couplets for spacings of
approximately /4 and

cost of an antenna system of a given height is more or less proportional
to its length, and in many cases is not materially affected by the number
These considerations, together with the fact

that proper phases may often be most readily accomplished with
intervals of either 14 wave-length or 14 wave-length, have led to a
rather general adoption of these closer spacings.

In Fig. 9, approximate gain ratios from arrays of various lengths
have been plotted. These are most applicable for separations in the
vicinity of 144 and % wave-length. Fig. 10 shows the same data
plotted in decibels. Within these limits, it appears that the gain ratio
may be expressed by the simple formula G = KL, where L is the array
length in wave-lengths and K is approximately 5.6. The result
expressed in decibels is G’ = 10 logio(KL).

The degree to which the gains calculated above are approximated in
practice is indicated by the data given in the diagrams of Figs. 11 and
12 and in Table I.

* This antenna actually consisted of two arrays of four couplets each spaced
laterally by one wave-length. The resultant diagram of such an array is for all
practical purposes the same as that produced by a continuous array of nine couplets.

Fig. 11 shows a calculated diagram corresponding to certain
receiving arrays used in the transatlantic telephone service between
America and England. Several points are plotted on this diagram
which correspond to the relative strengths of signals received at vari-
ous angles. These points were obtained by observing the relative
received signal voltage, measured on a standard field-strength measur-
ing set connected to the array as an electric oscillator of constant
amplitude was carried around the array at a distance of perhaps 20
wave-lengths. The plotted data correspond to the case where the

reflector was ‘‘floating.’’ Although this arrangement most nearly
corresponds to the conditions assumed in the calculated curve, it is not
necessarily the most desirable adjustment to minimize noise arriving
from the rear. This diagram corresponds to the antennas designated
as 1-A, 2-A, and 3-A in Table I. These antennas consist effectively of
24 vertical couplets spaced horizontally at intervals of 14 wave-length.

In this table are given further data on the strength of signals
received on arrays, as compared with those received simultaneously on
a single element of similar structure and height above earth. The
different antennas represented involve varying conditions of wave-

Fig. 11—Calculated directional diagram. Twenty-four couplets spaced one-fourth
wave-length. Circles indicate experimental points.

length, height above earth, adjacent terrain, and types of support.
These details are not believed to be of sufficient importance for dis-
cussion here. Two different array lengths are represented. The rela-
tive gains were substantially the same when observed on a local source
of waves and when the signal came from a distant station. The last
array represented in Table I was one used for transmitting. To effect
the test, equal power was transmitted alternately from the array and
from a single element while comparative measurements of electric
field strength were made at a distance of approximately 3500 miles.
The datum given is the mean of perhaps 100 observations extending
over a total of eight hours on three different days. Two errors are

involved in the data of Table I. One is due to the doubtful magnitude
of a correction necessary to account for the various heights at which
the arrays were located above the earth and the second is the error of
measurement of gain as compared with the reference antenna. These
errors are approximately equal and together amount to + 1 db.

In order to test further the agreement between measured gains and
those calculated from the simple assumptions above, a receiving array
was assembled step by step and corresponding measurements made.
Certain precautions, such as to maintain impedance matches at points
of coupling, were observed. The resulting data were plotted as points
in Fig. 12. A smooth curve represents the corresponding calculated

Fig. 12—Relation of measured to calculated gain of receiving antenna array at
14,350 ke.

data. -It will be observed that the measured values are consistently
higher than those calculated at the lower end of the curve, and in this
region the agreement can hardly be regarded as satisfactory. How-
ever, limited time prevented a thorough study of the errors of measure-
ment. Consequently these limited data may not be regarded as any
adequate test of the theory.

It may be shown that two or more similar directive systems may
be combined to give a total directive effect, represented by the product
of the individual effect, multiplied by the group effect. This principle
is partially covered by equation (35) of the mathematical appendix.

Two cases are of special interest. First, it is sometimes desirable to
divide an array into two or more bays, in order to make room for a
supporting structure. This, of course, gives rise to a definite discon-
tinuity in the over-all array.

Fig. 13 shows a series of diagrams resulting from a typical case
of two such arrays, each having a length of 2% wave-lengths but
separated variously from 0 to 2 wave-lengths in steps as noted. These
diagrams, of course, do not take into consideration the reaction re-
sulting from proximity to an antenna mast, located in such an opening.
The most important résult is to emphasize the spurious lobes, as the
spacing between arrays is increased.

A second effect of grouping which is of considerable interest is that
of varying the direction of transmission by altering the respective
phases between two or more arrays or between sections of the same
array. In Fig. 14 a series of diagrams is shown for a typical case of
two 314 wave-length arrays, spaced one wave-length. All elements in
the same array are driven in phase, but the two arrays differ in phase
by various amounts, as noted. It will be observed that the possible
rotationai effect is very limited. The general equation for this diagram
is given by formula (36) of the mathematical appendix.

This effect was investigated further by assuming a continuous array
714 wave-lengths long, made up of 16 couplets spaced at intervals
of 1% wave-length. The results are depicted in Fig. 15. The top row
assumes that the array is divided into two sections of eight couplets
each. This gives similar but not exactly the same results as those of
Fig. 14. The array, however, might have been divided into other sec-
tions for purposes of phasing. The various possible combinations are
tabulated below:

Diagrams in rows two, three, and four show that, as the array
continues to be divided into smaller sections, the direction of trans-
mission is capable of greater variation without sensible loss of sharp-
ness. If the array be divided into two sections this range is limited
to perhaps 3 deg. as in the case depicted in Fig. 14. Although this is
very moderate, it is extremely useful in correcting for any errors in
the orientation of the supporting structure or possibly correcting for
deviation of the projected radiation caused by peculiarities of the ad-
jacent terrain.

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If the array is divided into four sections the rotation may extend
over a range of perhaps 9 deg., while for eight sections it may be 15 deg.
The final case of 16 sections of one couplet each permits of considerable
flexibility such as would be useful in operating with several distant
stations in the same general direction. It should be pointed out, how-
ever, that the problem of making 16 phase adjustments each time a
station wishes to change its direction of transmission is of considerable
magnitude. For the particular case illustrated above it appears that
the maximum rotation of the projected radiation is more or less pro-
portional to the number of sections into which the array is divided.
It may readily be seen from the two top rows of diagrams in Fig. 15
that continued addition of phasing amounts effectively to negative
rotation. This may also be seen from an analysis of the equation of the
diagram.

The successful use of an array of couplets to give unidirectivity
suggests that the use of more than two parallel linear arrays might
further be employed to advantage. Obviously many such combina-
tions are possible, but one of some interest has been investigated
below. As a concrete example of this variation of gain with arrange-
ment of arrays, a series of diagrams for 36 elements has been plotted
in Fig. 16. The condition of spacing and phase intervals between

columns of each of 14 has been chosen. The horizontal character-
istic is given for separations between rows of both 4 and 14 wave-
length. The vertical characteristic common to these two separations
is also shown. ‘The equation of the diagram is given in formula (17)
of the mathematical appendix below.

It will be observed from Fig. 16 that the horizontal directivity is
for the most part only moderate, but approaches a maximum for the
condition where a long broadside array prevails, whereas the vertical
directivity is increased by increasing the number of columns in the
field. A substantial loop will be found near the rear of diagrams corre-
sponding to an odd number of columns. It is of further interest that,
as far as horizontal directivity alone is concerned, the optimum may
be derived either from a single array of 36 elements or from 18 couplets.
Considerations of both minimum interference and total gain, however,
‘ make the latter preferable. These conclusions may also be reached by
more direct analysis.’

7 Wilmotte, “General considerations of the directivity of beam systems,” Jour.
I. E. £., 955.

im = RATIO OF AREA OF SUBORDINATE LOOPS TO THAT OF
MAIN LOOP
ip’=RELATIVE PHASING BETWEEN SECTIONS OF AN ARRAY

$=540 $=540 $=34.7 $=270 $010.2
S10 $22.0 S=2.2
— ARRANGEMENT OF ARRAY —— — EQUATIONS OF DIAGRAMS — ‘
COLUMNS Sin (NTA Sin Sin (mn [cos 6-1
N SIN (TA sin 6) 1 sin (cos ¢-1))
eoeeee TRANSMISSION N SIN (1A SIN SiN [sin @-
Ae S =RATIO OF AREA OF UNIT CIRCLE TO THAT OF
S’= RATIO OF AREA OF TANGENT CIRCLES TO THAME
Jal. sin (n + R =RATIO OF AREA OF SUBORDINATE LOOPS TO
NSIN (TA SING) SiIN(Z [SIN O+ A = FRACTION OF WAVE LENGTH SPACING BET

SIN (1 A SIN 6) SIN SIN 1
( ) Gt ) S =RATIO OF AREA OF UNIT CIRCLE TO THAT OF DIRECTIONAL DIAGRAM

S’= RATIO OF AREA OF TANGENT CIRCLES TO THAT OF DIRECTIONAL DIAGRAM
N (NWA 0) (n R =RATIO OF AREA OF SUBORDINATE LOOPS TO THAT OF MAIN LOOP
SIN (T A SIN@) 1 SIN ( [sin @ + ) A=FRACTION OF WAVE LENGTH SPACING BETWEEN ELEMENTS IN SAME COLUMN

\ S = RATIO OF AREA OF ONE TANGENT CIRCLE TO THAT OF DIRECTIONAL DIAGRAM
R= RATIO OF AREA OF SUBORDINATE LOOPS TO THAT OF MAIN LOOP
A= FRACTION OF WAVE LENGTH SPACING BETWEEN ELEMENTS

Fig. 17—Vertical plane diagrams due to couplets of coaxial antennas—number of couplets versus separation in wave-lengths.

Thus far the discussion has centered mainly around directivity
produced by placing vertical antennas in horizontal array. Added
gain may be had also by incorporating directivity in a vertical plane.’
This is frequently accomplished by arranging individual antennas one
above another with their axes collinear, and is sometimes known
as stacking. The fundamental principles of analysis are the same as
those already utilized. However, an approximate correction must be
allowed to account for the fact that the radiation from a linear oscilla-
tor increases from zero along the axis to a maximum in a plane per-
pendicular to the axis. The directional characteristic in planes passed
through and parallel to such a radiator is approximated by two
tangent circles.

Fig. 17 shows a series of directional diagrams indicating the re-
sults of stacking unidirectional couplets. The diagrams shown refer
to the plane passed through the axes of the two linear oscillators com-
prising the couplet. On each diagram is a unit circle corresponding to
a single point source. Inscribed are the two tangent circles, represent-
ing the vertical directional characteristic of a single linear source.
Inside one of the tangent circles is the final directional diagram of the
stacked array. The ratio of the area of the tangent circles to that of
the characteristic diagram is given under each figure. This may be
regarded as a rough measure of the relative gain. These diagrams are
arranged horizontally in order of increasing number of couplets and
vertically in order of separation. It frequently happens in practice
that each radiator is approximately 14 wave-length long so it is con-
venient to utilize a vertical spacing interval also of 14 wave-length.
Consequently the second row of diagrams is probably of greatest
practical interest. In calculating these diagrams earth effects have
been ignored.

In Figs. 18 and 19, the gain in decibels to be expected from stacking
couplets has been plotted against number of couplets and fractional
wave-length spacing. These values, like those for Figs. 7 and 8 above,
were calculated by integrating the equation of diagram over a sphere
of arbitrary radius. This was accomplished by use of equation (30)
below. On account of the limited data at hand, Figs. 18 and 19
should be regarded only as a convenient method of illustrating the
trend of the variables. These indicate that somewhat lower corre-
sponding improvements result from stacking than from increasing the
length of an array.

The gains of arrays combining both horizontal and vertical direc-
tivity may not be simply calculated by adding the gains (expressed
in decibels) corresponding to elements arranged respectively along the
two principal coordinate axes. However, they may be calculated
except for earth effects by means of equation (26) below. Some cal-
culations of this kind have been made and the data are tabulated below.
They assume a total of 36 couplets which are arranged variously as
noted. In the first case all 36 couplets are arranged as a simple
horizontal array. The second case assumes that they are arranged in a

TABLE II
Number of Couplets Number of Couplets Gain over Single
Along Horizontal Along Vertical Half-Wave Element

Fig. 20—Approximate three-dimensional diagram. Linear antenna array with
reflector. Aperture two wave-lengths by eight wave-lengths.

broadside rectangle two elements high and 18 elements wide. This
combination may be regarded as two arrays of 18 couplets arranged one
above the other. The third case similarly assumes three arrays of
12 couplets each. A separation between couplets of 14 wave-length
has been assumed throughout. The most economical arrangement of
sfich an array depends not only on the relative costs of real estate
and towers but also on feed-line losses and effects due to the proximity

of the earth. The latter have specifically been omitted in this dis-
cussion.

Fig. 20 shows roughly the calculated directional characteristics of
a typical stacked array incorporating both horizontal and vertical
directivity. The planes passed through the diagram serve only as
convenient references to assist in visualizing the horizontal and vertical
diagrams. Earth effects of course, have been ignored.

A general case of linear arrays which includes those used exten-
sively in short-wave radio work, consists of a number of sources equi-
spaced and equiphased along each of the three principal coordinate
axes such that the space between sources is made up of rectangular
parallelopipeds with the individual sources located at each corner.
This may be regarded as N paraliel planes each made up of N parallel
columns where each column is made up of m individual radiating ele-
ments. The arrangement is made more evident by Fig. 21. The

usual conventions for representing three-dimensional space have been
adopted. We may designate the spacing between elements along the
x, y, and z axes, respectively, by a\, AX, and Ad and their corresponding
phase displacements between adjacent elements along the three princi-
pal axes by b7, BT and BT.

Similarly the time phase of any particular element relative to the
origin is
dawn = [(n — 1)b + (N — 1)B + (N — 1)B]T. (2)

The instantaneous value of the electric field at any remote point P
due to one of these sources is given by

where n’ =
The resultant interfering effect at a point P due to ’ such sources
all of equal amplitude is given by

Reducing to common voltage level and including a term sin 6 to cover
the case of radiation from linear oscillators we have for the equation
of the directional diagram

It will be recognized that this equation is made up of four factors.
The first three account for the effects of the disposition of elements
along the x, y, and zg axes, respectively, while the fourth, of course,
accounts for the direction of radiation from a linear oscillator. This
is an equation giving magnitudes only. In plotting polar diagrams
from this equation negative signs have no physical significance, and are
plotted in a positive sense.

An examination of this equation shows that there are many possi-
bilities which allow radiation in preferred directions, and at the same
time limit it in others. Some of these are discussed below.

This corresponds to the practical case of transmission along the
x axis from an antenna curtain and reflector made up of N vertical
columns of N elements each.

GAIN *F DIRECTIVE ANTENNAS 87
The equation for the diagram in tie (XY) plane may be had by
placing @ = 2/2 giving

which is the equation of the diagrams in Fig. 5 above. The corre-
sponding equation for the principal vertical section may be had by
placing ¢ = 0 and ¢ = r giving

which is the equation for the diagrams of Fig. 17.
The diagram of a single linear array of point sources is specified
by the first term of equation (12) where @ = 7/2 or

The diagrams of Figs. 3 and 4 above may be calculated from equation
(1€) by placing nm = 2 and n = 16, respectively. This also agrees with
Foster’s equation (1), page 367.?

The diagram of a field of coplanar linear arrays such as depicted
in Fig. 16 above follows from equation (12) by placing N = 1, a = {,
b= —jfandB=0.

The flow of power through each unit area due to an advancing
electric wave is given by the Poynting vector as

The radiated powers of these two systems might be so adjusted
at the source as to give equal fields at any point along a preferred
direction. <A ratio of these powers, therefore, would be a convenient
measure of the relative directional properties of the two arrays. This
“test ratio’’ may conveniently be set up in terms of the equations of

If we assume all comparisons are to be made with respect to a single
linear oscillator the denominator reduces to 87/3, so

This ratio may conveniently be expressed in decibels. In which
case G = 10 logio 1/7 is sometimes called the gain of an array.

If we are interested in the solid array shown in Fig. 21, where
n-N-N linear oscillators, each having respective space and phase
separations of ad, bT; AX, BT; and AX, BT, are arranged progressively
along the three principal coordinate axes, this becomes

This integration has been carried out by R. M. Foster who has very
kindly placed the results at the writer's disposal. Only the final
result is given herewith:

This, like equation (13), corresponds to the practical case of
transmission from an antenna curtain and reflector each made up
of N vertical columns of N elements, all driven in the same phase.

(2) If we assume that no stacking is involved, then N = 1 and we
have for the test ratio for NV couplets

(3) If we wish to apply equation (25) to the case of a single array
of .N linear oscillators driven in phase we have n = N = 1 and B = 0,
so

which differs from equation (27) by a factor of two. This indicates
that an array of N equiphased linear couplets gives twice the field in
the preferred direction as received from N equiphased linear elements
radiating the same power.

(4) Applying equation (25) to the extremely simple case of one
couplet, nm = 2,a = 3,6 = —jand N = N = 1 and

(5) We may calculate the test ratio for a single stack of linear
couplets (earth effects not considered) by placing N = 1,” = 2,a = }
b=- eed 0 and get

GAIN OF DIRECTIVE ANTENNAS 91
This equation was used in calculating the data given in Figs. 18 and 19.

(6) The test ratio for the case of the rectangular array of nV
elements discussed - conanetinm with Fig. 16 may be calculated by
placing N = 1,a = 1,6 = —}and B = 0. In which case

In general, the areas of directional diagrams may be calculated
from their equations by the usual integration methods. The special
case of N couplets in horizontal array, such as used rather generally
in practice and shown in Fig. 5 above, is of sufficient importance to be
given here. The area of the diagram in the (XY) plane is

N-1
= (N — K) Jo(2xKA) cos |. (32)
=1
This equation was used in calculating the data given in Fig. 5.
The area of diagrams in the horizontal plane due to a single array
of N oscillators is given by the equation:

This differs from equation (32) by a factor of two and indicates that
regardless of whether the gain is reckoned by an integration over a
unit sphere or in terms of the area of the horizontal diagram the effect

of the reflector is to double the radiated field in the preferred direction.
Placing N = 1 in equation (32)

Each element of a generalized linear array, such as shown in Fig. 21,
may be replaced by a generalized array, thereby producing an array
of arrays. It may be shown that the resultant is given by an array
factor, representing the characteristics of individual arrays, times
other factors representing the relative position of the individual arrays
in the array of arrays. A derivation analogous to that beginning on
page 22 results in the equation

where a’\, A’ and A’) are the coordinate spacings between arrays and
b'T, B'T, and B’T are the corresponding phase intervals, and r repre-
sents the characteristics of one of the individual arrays. If each array
is of the type shown in Fig. 5, r is given by equation (14) above.
Placing n’ = N’ = 1 and N’ = 2 also n = 2 and B = 0, the above
equation reduces to

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Several methods have been used or proposed for the calibration of the
Wente condenser transmitter. The methods falling under the two classifi-
cations conveniently designated “constant pressure’? or ‘“‘pressure”’
calibration and ‘‘constant field” or “‘field”’ calibration are most useful and
amenable to measurement. Which of these two calibrations is more
significant depends on the particular use made of the transmitter. In the
following pages the methods now used or proposed are reviewed and the
advantages or disadvantages of each from the standpoint of transmitter
application are discussed.

diaphragm resonance was well above 10,000 c.p.s. The new design
(Western Electric No. 394-Type), developed by Wente, has an
effective resonance at approximately 5,000 c.p.s. It is about ten
times more sensitive (on a voltage-pressure basis), and more immune
from effects of humidity and of barometric changes. The important
external dimensions of the instrument are shown in Fig. 14.

The response of the transmitter is defined as the ratio of the electro-
motive force generated to the acoustic pressure acting on the trans-

Fig. 1B—Contour dimensions of condenser transmitter used for field calibration.

mitter. That ratio [[R(f) = e/p], as a function of frequency, gives the
calibration. Where and how is the acoustic pressure to be measured?
This can be done in any one of a number of ways, all of which in
general lead to different calibrations. The two calibrations most
useful and amenable to measurement are when the pressure is uniform
over the diaphragm and measured at the diaphragm and when the
pressure is the pressure in a progressive plane wave, undistorted by the
transmitter or any other obstacles; when the electromotive force is
measured the distortion of the sound field must be due to the trans-
mitter alone.

It is convenient to designate the former as ‘constant pressure”’ or
‘‘pressure”’ calibration, the latter as the ‘‘constant field”’ or “field”
calibration. In general the field calibration will depend on the angle
of wave incidence. Incidence normal to the diaphragm gives the
“normal field’’ calibration. Where no confusion can arise, “field”’
calibration will be used to imply normal incidence. The pressure and
field calibrations tend to coincide when the transmitter dimensions are
small compared to the sound wave-length and when there are no
appreciable impedances between the diaphragm and the sound field in
front of it. Neither condition obtains for the No. 394-Type Trans-
mitter, except at very low frequencies.

Which of the two calibrations—‘‘pressure’’ or “‘field’’—is more
significant depends on the particular use made of the transmitter.
Thus in the receiver testing machine, where the sound is substantially
uniform throughout a small chamber closed by the transmitter
diaphragm and by the receiver under test, the pressure calibration is
important. When the transmitter is used to pick up sound in the
open air at a distance from the source, the field calibration applies.
For other cases, neither calibration is directly applicable, this being
discussed at the end of the paper.

For the several methods available for constant pressure calibration,
the pressure may be applied either acoustically or electrically. In the
acoustical group are the following methods:

4. Compensation methods.
a. Electrodynamic compensation for acoustic pressure.‘
b. Electrostatic compensation for acoustic pressure.®

In the electrical group for pressure calibration are the following
methods:

6. The back electrode (backplate) serving as the driving electrode.*: ¢
7. An auxiliary third electrode driving the diaphragm.

1. Thermophone.—The alternating pressure generated in the chamber
of which diaphragm D (see Fig. 2) is one wall, is computed from the
physical constants of the thermophone 7, and of the gas (hydrogen)
filling the chamber. A computation similar to that in reference? is
discussed in Appendix I and II. The difference is in the manner in

which the heat conductivity of the walls is taken into account. Alsoa
slight correction for the yielding of the diaphragm is introduced, which
was superfluous with the earlier, less sensitive model. An important
advantage of the thermophone method is that it is not necessary to
have the heating element parallel to the diaphragm. This makes it
applicable to transmitters with curved or corrugated diaphragms. In
such cases it is difficult to provide the accurately parallel and narrow
spacing between the diaphragm and driving or compensating electrode,
required in electrostatic methods.

2. Pistonphone.—The pressure is generated by means of a recipro-
cating motor-driven rigid piston as shown in Fig. 3. The piston
amplitude is computed from the dimensions and the angular velocity
of the cam driving it. The motor drive makes the method suitable for
relatively low frequencies, up to about 200 c.p.s.

3. Resonating Tube.—The pressure at the diaphragm end of the
tube (see Fig. 4) is computed from a measurement of the air particle
velocity at a pressure node. That velocity is obtained by observing
the deflection of a Rayleigh disk, R. D., placed in the tube. The
sound source R is shown as a moving coil receiver.

4. Compensation Methods.——The pressure in the chamber is de-
termined by measuring the force required to prevent motion of a

small auxiliary diaphragm D2, Fig. 5. With the sound pressure so
determined the corresponding electromotive force of the transmitter is
measured. The rest condition of D2 is indicated by absence of sound
in an exploring tube communicating with the space back of D2 or by
absence of frequency variation in a high frequency circuit in which D,
is made one plate of a condenser controlling the oscillation frequency.

4a. Electrodynamic Compensation for Acoustic Pressure-—The com-
pensating pressure is provided by sending a current of adjustable
frequency, amplitude and phase through D; placed in a steady magnetic
field.

4b. Electrostatic Compensation for Acoustic Pressure -—The same end
is attained with a potential difference of adjustable frequency, ampli-
tude and phase applied between D; and a fixed electrode parallel to it.

In particular the transmitter diaphragm and backplate may serve as
Dz and the fixed electrode. This, however, requires caution. The air
gap is so small (approximately 2.5 X 10-* cm.) that unavoidable
variations in its value will in general cause appreciable variations in the
value of the electric driving force over different parts of the diaphragm.
The non-uniformity of the air gap is due to mechanical imperfections
and to the electrostatic pull of the polarizing voltage. Furthermore,
in transmitters of the type here considered, the backplate diameter is
substantially smaller than that of the diaphragm, and hence the
compensating electric force is not effective in a peripheral portion of
the diaphragm.

The electric force in this case is provided by iiserting between the
diaphragm and the backplate a steady potential difference, Vy, and a
much smaller alternating potential difference, V; sin wf, in series. One
of the resultant force components is aVoV, sin wt which has the same
frequency as the sound source (e.g. a thermophone). The amplitude
and phase of the electric force are adjusted until it balances the
acoustic pressure on the diaphragm. This gives the value of the
acoustic pressure, provided a is known. The compensating electric
force is then removed, and the output of the transmitter due to the
acoustic pressure is measured. Thus the pressure calibration is
obtained. The value of @ is given by a measurement of the value of
Vo required to balance a known static gas pressure established at the

face of the diaphragm. This must be done for each instrument to be
calibrated.

5. Membranephone.—In principle this method is similar to the
pistonphone. An acoustically driven membrane .\/ (see Fig. 6)
replaces the motor-driven piston. From the volume displacement,
AV, of M the pressure on the transmitter diaphragm D is computed.
The value of AV is given by a measurement of the alternating variation
in capacitance between M/ and an auxiliary perforated electrode G.
The range of the method is from the lowest frequencies up to those at
which the linear dimensions of the chamber become comparable with
the sound wave-length (A). As with the thermophone, that upper
limit can be extended through the use of hydrogen instead of air.

The computation of AV is given in Appendix III. It will be noted
that the computation is independent of the mode in which the mem-
brane vibrates. However, for frequencies above the first resonance of

the membrane the requirement as to smallness of chamber dimensions
relative to \, becomes much more stringent than in the thermophone
case.

Methods Employing Electrical Drive —Since the driving forces in this
group are electric the pressure on the diaphragm is affected by the
acoustic load on the front face of the diaphragm. To obtain the true
pressure calibration that acoustic load must be known. Practically
this is taken care of by making that load sufficiently small, rather than
accurately determining its value.

6. The Back Electrode Serving as the Driving Electrode—The alter-
nating potential difference, V; sin wf, is impressed in series with the
steady potential Vo, see Fig. 7. This gives a driving force component
aVoV, sin wt. The corresponding alternating variation in the trans-
mitter capacitance is determined by having that capacitance control
the frequency of a high frequency oscillator circuit. Absolute values
are obtained by means of a static pressure calibration as in Method 4.

In this case, however, that does not give the force acting on the dia-
phragm unless the air impedance between the diaphragm and back-
plate is negligible in comparison with that of the diaphragm itself.
Hence the method does not apply to the No. 394-Type Transmitter.
The same consideration as to non-uniformity of the driving force over
the area of the diaphragm which was mentioned in connection with
Method 40, applies to this case.

7. Auxiliary Third Electrode Driving the Diaphragm.—Here an
auxiliary electrode .V and a circular metal screen furnishes the electro-
static drive (see Fig. 8). It has nearly the same diameter as D and is
parallel to it. The gap between ./ and D is about thirty times greater

Yo
Fig. 7-—Electrostatic method—Back electrode serving as driving electrode.

than between D and the backplate. Hence the electric force on D is
uniform over the surface of D, and its absolute value can be computed
with some accuracy. The calculation is given in Appendix IV. Care
must be taken to avoid acoustic loading of D in a manner that would
materially change its impedance. With this possibility guarded
against, this method admits of an absolute transmitter calibration
from 20 to 20,000 c.p.s. A comparison of a calibration so obtained
with that given by a thermophone for the same transmitter,* is shown
in Fig. 9. The two are quite independent. The discrepancy between
the two up to about 6,000 c.p.s. is regarded as being within limits of
experimental error. The acoustic load imposed on the diaphragm by

* This particular instrument happened to be about 4 db less efficient than the
average No, 394-Type Transmitter,

the calibrating apparatus, while relatively small in either case, is not
the same for both methods. At higher frequencies other factors
contribute. At the highest frequencies, say above 10,000 c.p.s., the
pressure on the diaphragm probably is more uniform in the present
method than in Method 1.

For constant field calibration methods it is difficult to provide a
plane progressive wave over a sufficiently large wavefront. Instead a
small source in a chamber lined with highly absorbing material is used.
The resultant progressive spherical wave, at sufficient distance from

Fig. 8—Electrostatic method—auxiliary third electrode driving diaphragm.

the source, gives approximately the desired sound field. The measur-
ing device must give the absolute value of the undistorted field in-
tensity. We shall not consider the thermal, optical and sound radi-
ation pressure methods possible, on account of the experimental
difficulty which they present. One other absolute method is more
readily available:

The Rayleigh Disc, which on certain assumptions gives the absolute
value of the particle velocity in the sound wave. In the sound field
presupposed for the field calibrations, the corresponding sound pressure
is easily computed.$

Another procedure is to measure the sound pressure with the aid of a
“search transmitter.’’ This is a transmitter whose dimensions are so

small relative to the sound wave-length that its pressure calibration, as
obtained say by Method 1, may be taken to coincide with its field
calibration.

The normal field calibration of a No. 394-Type Transmitter is
shown in Fig. 10. The contour of the particular instrument used is

shown in Fig. 1B. It was suspended from a thin rod clamped to the
metal band B. The measurements were made with a Rayleigh disc

(0.5 cm. diameter, 2.46 second period), using the modulated sound
method.’ The transmitter was placed 32 cm. from the sound source, a
1-cm. diameter tube attached to a loud-speaking receiver. The data
obtained for frequencies below 500 c.p.s., are believed to be not so
reliable as the rest because of appreciable reflections from the chamber
walls.

FREQUENCY IN CYCLES PER SECOND
Fig. 9—Comparison of two pressure calibration methods.

For purposes of comparison, the pressure calibration (Method 7) of
the.same instrument is shown. At the lowest frequencies the two
calibrations nearly coincide, as might be expected. At high fre-
quencies, say from 1,000 c.p.s. upward, the divergence of the two is
quite marked. It has been pointed out by several writers that the
difference may be regarded as due to two effects. First,!® as \ de-
creases, the transmitter tends to cause a doubling of the pressure in
front of it as would a rigid wall. Second," the recess in front of the
diaphragm (Fig. 1) introduces a broad resonance which has its maxi-
mum approximately at 3,500 c.p.s. An estimate of this effect is given
in Appendix V.

The observed differences between the field and pressure calibrations,
from 500 to 8,000 c.p.s. are in fair agreement with those computed for

the two effects given above. The computations are based on as-
sumptions as to the transmitter contour which are quite removed from
the actual case. Thus for the first effect it has been suggested that the
transmitter may be replaced by an “equivalent” rigid sphere of equal

Lt
20 50 100 500 _—1,000 5,000 10,000 20,000
FREQUENCY IN CYCLES PER SECOND
Fig. 10A—Pressure and field calibrations of No. 394-type condenser transmitter.

Fig. 10B—Field calibrations of No. 394-type condenser transmitter for normal and
random incidence.

volume or of equal diameter.'® The data in Fig. 10A are best fitted
by assuming a sphere of 9 cm. diameter, i.e., a diameter even larger
than that of the transmitter. For the second effect the assumption is
made that the face of the transmitter acts as an infinite wall, and that

the air particles in the recess aperture all move in phase and normally
to the diaphragm.

At still higher frequencies the doubled pressure effect largely persists
and superposed on it are a number of rather complicated diffraction
effects. These involve radial wave propagation across the diaphragm
recess while the above two effects are due to normal plane waves.
The marked dip at 11,200 c.p.s. corresponds to a sound wave-length
such that

So far normal incidence of the sound wave has been assumed. For
other directions of arrival, substantially different field calibrations are
obtained. Since the transmitter is symmetrical about any diaphragm
diameter, the effect of direction may be given in terms of the azimuth
angle of incidence. A set of azimuth curves for various frequencies
are given in Fig. 11, all expressed relative to the normal field cali-
bration. In general, the higher the frequency the greater the effect of
azimuth. Fora large range of angles that effect is as great as or greater
than the difference between the pressure and the normal field cali-
brations. It is interesting to note that the anomalous azimuth curve
at 11,200 c.p.s. corresponds to a pronounced dip at that frequency in
the normal field curve.

We now consider the bearing of field calibrations upon the response
of the No. 394-Type Transmitter under one or two conditions of
actual use.

First, consider the case of a person speaking directly toward the dia-
phragm. The normal field calibration approximately applies, provided
the distance is not great enough for reflected waves to be comparable
with the direct wave and the distance is not so small that the transmitter
reacts back on the source (the voice), or that pronounced standing
waves are set up between the transmitter and the head. Outdoors and
in a well damped room distances ranging say from 6 inches to 3 feet are
likely to be within the above limits for the important voice frequencies.

On the other hand, for much of indoor work the distances from the
microphone to the source and to the several reflecting surfaces are such
that waves reaching the microphone by reflections are comparable
with and often predominate over the direct sound. Besides, the
microphone often is so placed that the direct sound strikes it more

nearly at a 45-degree or 60-degree angle rather than normally. Ina
29-foot X 29-foot X 13-foot room having a reverberation time of 1

second, the reflected waves reaching the microphone at 12 feet from a
small source contribute much more to the microphone output than
does the direct sound. To illustrate the effect of these reflections, the
curve (b) in Fig. 10B has been plotted. It is based on the data of Fig.
10 and Fig. 11, and assumes that the transmitter is acted upon by
progressive plane waves arriving with equal intensity from all directions
in space. Their phases are taken to have random distribution. At
any one frequency the response of the transmitter is then proportional

where A(°) is the azimuth factor taken from Fig. 11. The result is
seen to be intermediate between the pressure and the normal field
calibrations, for frequencies up to about 8,000 c.p.s. Under these
circumstances it is immaterial which way the diaphragm faces, but
this holds only for sustained sounds. For sounds of short duration,
the peak amplitudes in the microphone output often are of particular
interest. They will be more nearly given by that single field curve
corresponding to the azimuth with respect to the sound source in which
the transmitter happens to be.

The above discussion of directional effects is simplified by the fact
that the No. 394-Type Transmitter is symmetrical about any dia-
phragm diameter. Hence a single parameter—azimuth angle—is
sufficient. The amplifier mounting cases usually employed destroy
that symmetry. The directional effect becomes much more compli-
cated since it involves two parameters, e.g. two direction cosines of the
diaphragm axis. It has been suggested * that this complication can
be done away with by placing the transmitter and its amplifier case in
a rigid hollow sphere, only the transmitter front being exposed. If
the front contour of the instrument be designed closely to conform to
the rest of the sphere, and if the diaphragm subtend a sufficiently
small angle at the center of the sphere, the directional effect can be
computed.'*

The simplest directional properties, i.e. uniform response for all
directions of incidence, require a transmitter whose linear dimensions
are small (say < 14\) relative to the shortest sound wave-length to be
picked up. For a frequency range extending to 10,000 c.p.s., this
means a transmitter less than 0.85 cm. in diameter. In general such
restriction on the permissible size adds to the difficulties of construction
and operation of the instrument. It is not intended to imply that non-

directivity of the transmitter is always desirable for pickup systems of
highest quality.

A complete description of the performance of the microphone as an
electro-acoustic converter is extremely complex. It involves the
microphone, the sound source, their relative positions, and the sur-
rounding acoustic configuration. Furthermore, it is limited to sound
sustained long enough to allow the reflection pattern to attain a steady
state. Therefore, in order to obtain a reasonably simple and useful
statement of the transmitter response, the field calibration is made
under the ideal acoustic conditions stated in part A. Even then the
field calibration (including, of course, the azimuth measurements) is far
more difficult and laborious than the corresponding pressure cali-
bration. For some important purposes the pressure calibration is
sufficient, even though the transmitter be intended for use in an
‘‘open’’ sound field. An instance is the specification and comparison
of instruments having similar contours. The difference between the
field and pressure calibrations, once determined for an individual
instrument, applies to all others. That is, provided the acoustic
impedances of their diaphragms are not too widely different, which
usually is the case. Therefore the response of any instrument, as a
function of frequency, age, barometer pressure, temperature, etc., is
given by the pressure calibration. The thermophone method (Method
1) is particularly suitable for rapid and reproducible determinations of
the pressure calibration.: That is the method employed for the
specification of No. 394-Type Transmitters, and of others having
similar contours, in the Master Reference Systems '*® for Telephone
Transmission in Europe and in this country.

I am indebted to Messrs. R. T. Jenkins, H. T. O’Neil and E. M.
Little of Bell Telephone Laboratories for much of the material used in
this paper.

APPENDIX I
The pressure generated by the thermophone is slightly reduced by
the heat conductivity of the chamber walls. That conductivity is so
' great as compared with that of the gas, that zero temperature variation
at the walls may be taken as one of the boundary conditions of the
problem. This results in a solution nearly identical with that of eq.
(7), p. 336, in the original derivation.2, The correction factor given
there on p. 340, which takes care of the wall conductivity, is now
found to be more nearly unity. The difference between the two
solutions is shown in Fig. 12 for a special case typical of condenser
transmitter calibrations. As might be expected, it is greater the lower
the frequency.

Fig. 12—Pressure generated by a thermophone in a transmitter calibration chamber.

In the thermophone theory the walls of the chamber were treated as
being rigid. Actually the transmitter diaphragm presents a small but
finite admittance in shunt with the elastic admittance of the gas in the
chamber. The correction factor M due to this, is approximately

Vo = volume of thermophone chamber,
v = volume displacement of diaphragm per bar,

phase angle of above displacement with respect to the pressure on
the diaphragm.

At low frequencies cos @ may be taken as nearly unity, and v can be
approximately computed as below

C,; = capacity between diaphragm and back plate without
polarizing voltage.

For the 394-Type Transmitter, up to about 2,500 c.p.s., 1 is nearly
0.92. Above that the correction decreases owing to decreasing cos @,
and becomes negligible at 5,000 c.p.s. For still higher frequencies the
correction becomes negative but remains small due to the increasing
diaphragm impedance.

Schematically the membrane phone is shown in Fig. 6. D is the
diaphragm of the transmitter to be calibrated; ./, a stretched membrane
acoustically driven from the receiver R; G, a perforated plate. Let
V = volume between D and M; yo = normal separation between G
and M; Cy = normal capacitance between G and M.

Then, if yol1 + K(S)- sin wt] represents the G.V/ separation when J/
is driven by R, the resultant capacitance variation is:

Taking A(S) << 1, but without restrictions on the variation of
K(S) over the surface of J,

The above presupposes: (1) V/S << A, vS << A; (2) acoustic
admittance of D is very small compared with that of V; (3) adiabatic
compression. If necessary, corrections for deviations from (2) can

be made in accordance with Appendix II. The correction for (3) is
found to reduce the pressure in the ratio

The upper frequency limit imposed by condition (1) can be raised by
filling V with hydrogen. For the No. 394-Type Transmitter, and with
Ra No. 555-W Western Electric Receiver, an air-gap yo = 0.075 cm.
corresponds to easily measurable values of e; and @:. J was a 0.001
inch duralumin diaphragm, stretched to 5,000 c.p.s. resonance fre-
quency. It was found that the upper frequency limit of the method is
determined by ./ breaking up when vibrating in one of its higher
natural modes. This tends to produce a non-uniform pressure on D,

and the above condition must be met much more perfectly than in the
thermophone case.

means for keeping the relatively large driving voltage out of the
transmitter output circuit.
In terms of Fig. 8 the sensitivity of the transmitter is given by

9 10° X 8y2-
where VV2 = Vi, measured in volts, i,Vv2 = i;, h = separation
between 1/ and D. Thee.m.f. eis measured by means of the potential
attenuator P, carrying a known current 7,, and having an input
resistance 7g. At any one frequency two quantities must be measured:
V, say with an electrostatic voltmeter, and a, the setting of the
attenuator in decibels. The current 7, must be known but can readily
be kept constant at all frequencies if a heterodyne oscillator be used as
the source.

Two corrections must be applied. First, the auxiliary electrode is
perforated. Hence not all of its area is electrostatically effective.
Second, p in the above is the electrostatic force per unit area, rather
than the acoustic pressure on the diaphragm. The two are different,
in general, because of the acoustic load (Z,4) on the front face of the
diaphragm. The value of Z2 is affected by form of C, the auxiliary
electrode, and by the acoustic impedance beyond it in the chamber G.

It is best to have Z, as small and as free from reactance as possible.
This is accomplished by using stretched fine metal gauze. Copper
gauze, 300-inch mesh, is quite good. It terminates in the tube 77,
l4-inch iron wall, which is filled with several layers of loose cotton
batting and hairfelt. The effectiveness of the arrangement was
judged by the fact that altering the size of G did not appreciably affect
the calibration.

While the screen electrode provides a practically uniform electro-
static pressure over the surface of D, it is rather complicated to
compute the effective absolute values of and of the electrostatic
area. This is more easily done by comparing it at low frequencies (say
at 100 c.p.s.) with a steel plate electrode in which the perforations
take up about 12 per cent of the total area. The surface facing D is
carefully machined so that / is uniform and known within less than

+2 percent. This is for absolute values of in the range 0.075—0.080
cm. The acoustic load which this electrode imposes on D, with G
removed, is negligible at low frequencies. A lower limit on the
electrostatic correction for the perforations is made by adapting the
calculation given by Maxwell (‘‘ El. Mag.,"’ 3d ed.) for rectangular
grooves in one plate of a parallel plate condenser. The above value of

The R’ actually used was the mean of the above two values. The
value of R obtained with the screen electrode is shifted up or down to
make it coincide with R’ given by the perforated electrode, at 106 c.p.s.

APPENDIX V
For frequencies below about 5,000 c.p.s. the difference between the
pressure and normal field calibrations is mainly due to two effects: (1)
reflection from the transmitter face and from the diaphragm; (2) air
resonance caused by the recess in front of the diaphragm.

Consider Fig. 14. Assume that in the circular aperture PQ, the
air particles are all moving in phase and parallel to AP. Then we may
treat PQ as a rigid massless piston in the wall RS. If RS/d is large
enough, the pressure on PQ held motionless will be double that of the
field pressure. The motional impedance of PQ imposed by the air
above PQ is given by Rayleigh (Sound, vol. IT, §802).  Perunit area it is

Let Rr/Rp represent the ratio of “ field ’’ to “ pressure ” calibration.
Using the expression for plane wave propagation in a tube (e.g.
Crandall, Theory of Vibrating Systems and Sound, p. 99) we have at
once:
Rr 1

where Z, is the equivalent impedance per unit area of the transmitter
diaphragm, and/ = AP. On substituting numerical values, Rr/Rp is

found to have a maximum value of nearly 3.3 at w/2I1 = 3500 c.p.s.
This means that the air resonance adds a factor of 1.65 to the ordinary
doubling of pressure caused by a plane wall. A substantially similar
calculation has been given by W. West.

The observed Re/Rp is a maximum at 3,500 ¢.p.s. but its value is
somewhat larger—nearly 3.65.

. L. J. Sivian, A Telephone Transmission Reference System, Elect. Comm., Oct.
1924. W. H. Master and C. H. G. Gray, Master Reference System for
Telephone Transmission, Bell Sys. Tech. Journ., July 1929.

This paper discusses the rating of the transmission performance of
telephone circuits on the basis of the rate of repetitions in telephone conver-
sations and presents the rating method set up on this basis, which is being
adopted in the Bell System for determining and expressing the data for
the transmission design of the telephone plant.

circuits is of course an essential in specifying the grades of trans-
mission service to be furnished, in designing, constructing and main-
taining telephone systems to provide the desired grades of service
economically and in the development of the various elements of the
telephone system which affect its transmission. As the art of tele-
phone transmission has developed and greater refinements have become
possible and desirable, changes have naturally been made in the
methods of specifying and rating transmission performance. Since
many such changes have been made in recent years, it seems opportune
to discuss the rating of transmission performance and to set forth the
rating method which is now being adopted in the Bell System for
determining and expressing the data for the transmission design of the
plant. In this connection, various methods which have been em-
ployed for measuring the transmission performance of telephone circuits
will be discussed to indicate their application and also their relation to
the new rating method. It is the purpose here to discuss this rating
matter primarily from the qualitative standpoint rather than to present
in quantitative detail the various relations involved in rating telephone
transmission. Obviously, the determination of many of these relations
presents sufficient material for separate treatment.

In carrying on a telephone conversation three major functionings are
involved, namely, that of the talker in formulating his ideas and utter-
ing words to convey these ideas, that of the telephone circuit in taking
the sounds of these words and reproducing them at another point, and,
lastly, that of the listener in hearing and recognizing these reproduced
sounds and in comprehending the ideas which they are intended to
convey. It is evident that all three of these functionings affect the
success of the telephone conversation. Since, however, the function-
ings of the talker and listener are common to both direct and telephone
conversations it might seem that the consideration of the transmission

performance of a telephone circuit could be limited to the functioning of
the circuit itself. In line with this, there has been some tendency to
confine such considerations to relations between the sounds reproduced
by the telephone circuit and the sounds impressed upon it. The per-
formances of the talker and listener, however, are materially affected in
certain important respects by the telephone circuit, and determinations
of the relative merits of the transmission performances of different
telephone circuits, must therefore go farther than the performances of
the circuits themselves and take account of the combined action of the
talker, circuit and listener.

In view of this reaction of the telephone circuit on the talker and
listener, attention is directed to the pertinent characteristics of their
performances in both face-to-face and telephone conversations.

In direct or face-to-face conversations both the talker and listener
more or less subconsciously adjust their actions in many respects to
each other and to their circumstances. The loudness of talking is
placed initially at a level which experience has shown to be suitable for
the conditions and for the particular listener. If the listener indicates
verbally or by his expression that he is understanding easily or with
difficulty, some further adjustment may be made in the loudness of
talking. Since the talker judges his own talking level largely by the
loudness with which he hears his own voice, this level will be a function
of the amount of reverberation in the place in which he is talking.
Apparently for a wide variation of loudness in the customary talking
range, the speaker is not in general conscious of the amount of energy
which he is expending. Noise at the place of conversation also plays a
part in determining the talking levels since it makes louder talking
necessary in order to permit a given degree of understanding on the part
of the listener.

Along with an adjustment in talking level, the talker may improve
his enunciation if difficulty of understanding is expected or indicated
by the listener. There may also be a change in the manner of express-
ing the ideas in avoiding words which experience has shown are difficult
to understand and an idea may be stated in more than one way in order
to insure its comprehension. Other adjustments on the part of the
talker may be determined by his opinion of the mental acuity of the
listener, by the familiarity of the listener with the matter under dis-
cussion and by the interest in it. These factors affect the way of ex-

pressing an idea, the kind and number of words used and hence the
time taken.

The listener also adjusts himself to the conditions by an amount
which is determined somewhat by his interest in the matter under
discussion. He may strive to comprehend the transmitted ideas and
require few repetitions by the speaker or he may refrain from exerting
himself and so tend to evoke greater effort on the part of the talker.
At times he may pretend not to understand in order to get confirmation
of a statement or to gain time in replying to a question. 5

In view of these factors and of the normal variations of different
talkers and listeners in all these respects, the portion of the questions
and statements of conversation which is correctly understood and the
time required to interchange certain ideas may vary widely for different
conversations even when they are carried on under a fixed set of local
conditions. If it were desired to determine a measure of the conversa-
tional satisfactoriness of these conditions, in addition to some quantita-
tive method for rating each conversation, there would be required,
therefore, observations on a large number of conversations between
different people in order to take account of the variables due to the
material of conversation, the people, and their abilities and desires to
accommodate themselves to the conditions.

In telephone conversations, there are adjustments between talker
and listener as is the case in direct conversations, but there are certain
definite differences in this regard because of the interposition of the
telephone circuit between the participants. Here also, the speaker
tends to adjust his talking level to the loudness with which he hears his
own voice. In this case, however, he hears his own voice not only
through the air path, but also through the “‘sidetone”’ of the telephone
set, that is, through the electrical path from his own transmitter to his
own receiver. When this electrical path is more efficient than the
acoustic path, the sidetone will tend to control the talking level. It
has been found that varying the sidetone of the set has on the average
a definite effect on the talking volume of the speaker, the talking
volume being lowered as the efficiency of the sidetone path becomes
greater.

In a telephone conversation, there is also a tendency for a person
in talking to adjust his volume on the basis of the loudness with which
he hears the person at the other end of the circuit. If the voice of the
other person comes through weakly, he judges that the connection re-
quires loud talking and acts accordingly. If the listener indicates that

he is not understanding, the talker may talk more loudly or closer to
his transmitter, and also make such adjustments in enunciation and in
setting forth his ideas as in the case of direct conversation. Also, the
loudness of talking may be affected by the room noise at the location
of the speaker, which noise incidentally may not reach the listener and
so play no part in his reception. Aside from cases where the room
noise at the far end is severe enough to be heard over the telephone
circuit, the speaker does not have definite knowledge of the room noise
at the listener’s end and therefore is not in a position to adjust his
manner of talking to this condition except in so far as the listener may
indicate difficulty in understanding.

In listening, the result is of course dependent upon the position of
the receiver with respect to the ear. The local room noise reaches the
ear to which the receiver is held both by the path between the ear cap of
the receiver and the ear, and also through the sidetone path of his tele-
phone set. Some telephone users have learned that this effect may be
reduced by holding the receiver tightly to the ear and by covering the
mouthpiece of the transmitter when they are listening.

It is evident that the success of telephone conversations depends not
only upon the performance of the telephone but also upon the perform-
ances of its users, the material of their conversations, the way in which
they talk into the transmitters and hold the receivers to their ears and
the room noise conditions. In addition, it is seen that the performance
of the telephone affects the performances of the users in such important
respects as the loudness of talking, the manner of presenting the ideas
and the amount of effort exerted to understand. Also, the effect of the
room noise is a function of the circuit characteristics. Furthermore,
the reactions of the circuit performance on those of the users are not
constant but may vary from person to person and from conversation to
conversation. In view of the random nature of these factors, which
are beyond the control of those who design and operate the telephone
system, the service performance rating of a telephone circuit should be
on a basis which takes adequate account of their ranges and combina-
tions in practice. This points to a rating based on a statistical analysis
of results obtained under service conditions.

To determine and specify these factors so that it may be known how
to duplicate the range of service conditions in laboratory investigations
would be a prodigious task. Moreover, the duplication of these condi-
tions under control is bound to introduce a large element of artificiality
which would vitiate the results or at least raise serious questions as to
their dependability.

The practical solution is to get sufficient data regarding the results
obtained over telephone circuits of different performance characteristics

by their normal users in carrying on regular conversations. This re-
quires a suitable quantitative method of rating conversations and
observations on a sufficient number of conversations over each circuit
condition to be investigated to constitute a reliable sample. This does
not mean necessarily that all the practicable circuit conditions have to
be observed in this manner but rather that sufficient data be so ob-
tained for the establishment of correlations with performance measure-
ments which are susceptible to laboratory determination. The funda-
mental point is that service performance ratings need to be based on
service results in order to take proper account of all the factors in-
volved.

The distinction has been made between two kinds of transmission
performance of a telephone circuit, namely, that indicated by relations
between output and input sounds and that indicated by the results ob-
tained by the users of the telephone in carrying on their conversations
under service conditions. Performance indications of the first kind
will be referred to as ‘‘transmission characteristics” of the circuit. The
second kind of performance may be termed ‘‘transmission service per-
formance.’ The distinction between these two kinds of performance
is an important one and should be kept clearly in mind.

The output sounds dealt with in transmission characteristics are not
only the reproduced sounds which correspond to the input sounds but
also the accompanying extraneous sounds which are delivered by the
circuit. Also, the output sounds to be investigated cover not only
those delivered by the receiver at the far end of the circuit but also
those reproduced by the receiver in the station set containing the
transmitter energized by the input sounds. The sounds from the near
receiver include both those transmitted through the sidetone path of
the set and those returned to the sending end by reflection at impedance
irregularities in the circuit. Due to the time required for propagation
over the circuit these latter sounds may be delayed with respect to the
sidetone and hence appear as echoes. Likewise, echo sounds may be
delivered at the far receiver.

Transmission characteristics do not in themselves show the service
performance as realized by the users of the telephone but are essentially
indications of the functioning of the circuit in reproducing sounds.
They provide, therefore, a means for investigating and specifying thé
performance of a telephone circuit without involving many, and in some
kinds of transmission characteristics any, of the actions of the talker
and the listener in conversation. With the establishment of proper

correlations between transmission service performance and transmis-
sion characteristics, these latter can of course be used to indicate
service performance.

In addition to specifying any kind or grade of circuit performance on
the basis of performance results there is the method, which has had
important practical application, of indicating performance in terms of
types of instruments and circuits and of the conditions of their use.
For example, a statement of the types of transmitters, receivers,
station sets and cord circuits and of the length and types of loops and
trunk, together with specific conditions of use, provides an indirect
specification of a performance. This method, which is extensively
used in many fields, may be termed the “
method.’’ An outstanding application of this method in telephone
transmission work is the Standard Cable Reference System! which
was so widely employed to provide a scale of performance. This
method has many present applications where physically determined
characteristics are unavailable or are difficult of definite determination
and specification. Also, the designation of instrumentalities is con-
venient in many cases because it provides a ready means of specifying
a practical combination of various kinds of transmission characteristics.
While this method is often expedient practically, taken by itself, it is
inherently cumbersome for the development of improved instrumental-
ities because of the lack of physical indication of the features to be
investigated.

The usefulness to the listener of the speech sounds reproduced over a
telephone circuit is a function of their loudness, of their distortion or
degree of departure from facsimile reproduction, and the magnitude
and character of the extraneous sounds or noise which accompany
them. Transmission characteristics are therefore directed primarily to
indications of the effects of the circuit and its parts on the reproduction
of sounds in these three respects. As already indicated, transmission
characteristics are determined not only for the path from transmitter
at one end to receiver at the other, but also for the sidetone and echo
paths.

Speech sound transmission characteristics, that is, expressions of the
relations between impressed and reproduced speech sounds, while they
have been extensively used, present some difficulty in quantitative
determination and specification because of their complex nature.
Also, the human element is involved in the persons used as generators

1“ Master Reference System for Telephone Transmission,” Martin and Gray,
Bell System Technical Journal, July 1929.

of the speeth sounds to be investigated and as observers to give indica-
tions of loudness and distortion and of their effects on the recognition of
the reproduced sounds. Two outstanding kinds of relations of this
type are those given by volume tests and articulation tests, which will
be discussed later. It has therefore been of great convenience to take
a further step and to study and specify the performance of telephone
circuits and their parts in terms of their functioning for single-frequency
sounds and currents. In this procedure, this functioning is investi-
gated for a number of different single-frequency sounds and currents,
so taken as to cover the range of frequencies transmitted by the circuit.
In the single-frequency transmission characteristics, the personal
element is eliminated and the measurements are made entirely on a
physical basis.

A great deal of attention has been given to the correlation of speech
sound and single-frequency transmission characteristics so as to enable
the former to be derived from the latter and so extend the application
of the type which is more readily susceptible to quantitative determina-
tion. Also, use has been made of easily specifiable multi-frequency
sounds and currents to permit the physical measurement to approach
more nearly speech sound conditions, of phonographic reproduction to
reduce the personal factor in the generation of speech sounds for meas-
urement purposes and of meter arrangements to simulate the ear rat-
ings of sounds, particularly from the standpoint of relative loudness.

As a result of the correlation of speech and single frequency charac-
teristics, extensive use has been made of determinations at selected
typical single frequencies to check the design, installation and main-
tenance of lines and other associated circuit elements.

The widely used volume test is essentially a means of specifying the
action of a telephone circuit or its parts, on the relation between the
reproduced and impressed sounds from the standpoint of their relative
loudness. In this test use has been made for many years of the
Standard Cable Reference System and recently of the Master Reference
System for Telephone Transmission? as references for comparison.
These reference circuits with their adjustable trunks provide a means
of obtaining different loudness ratios between input and output
sounds. By talking alternately over the reference circuit and the
one being investigated and adjusting the trunk of the reference
system until the output sounds of the two circuits are judged to be
equally loud, a specification of the loudness reproduction ratio is
obtained of the circuit under investigation in terms of the length of
the trunk in the reference system. The effect of a change in the

telephone circuit, such as the replacement of one receiver by another,
is measured in terms of the change required in the reference trunk
to give a loudness balance for the second condition. In this way,
measurements are also obtained of the effect on the loudness repro-
duction ratio of the various parts of telephone circuits. When the
circuits used commercially consisted of apparatus and lines similar to
those in the Standard Cable Reference System and the major con-
trollable factor was the loudness reproduction ratio, such measurements
constituted reasonably adequate means for indicating the comparative
functioning of circuits and apparatus.

The noise on a telephone circuit may be measured in various ways.
The method which has been most generally used is that of comparing
it with the controllable output of a fixed source of a complex wave shape
and adjusting this output until it and the line noise are judged to have
equal interfering effects.

With the availability of circuits and apparatus having widely differ-
ent distortion effects, the volume ratings became insufficient for indi-
cating the relative performances of commercial circuits. The earliest
method used in rating distortion effects was one in which observers
listening to transmission over the circuits, gave judgments as to their
relative merits. By so comparing various kinds and amounts of distor-
tion, two at a time, relative ratings can be established for placing them
in order of merit. This procedure was particularly useful in the early
days in working out the designs of transmitters and receivers, especially
from the standpoint of the location in the frequency range of their
points of maximum response. While such a judgment method has the
shortcoming of not providing quantitative ratings it has been found
that experienced observers can in general obtain results which are
relatively consistent with the results of more definite measuring
methods. Such judgment comparisons of distortion effects are fre-
quently used, particularly in exploratory work, and are still more or less
necessarily relied upon in setting limitations on circuit properties which
primarily affect the naturalness of reproduction.

To provide for the need of a method for measuring the relation be-
tween the reproduced and impressed sounds from the standpoint of
effects of different kinds of distortion, use has been made of the articula-
tion testing method.* In this method, which has been widely used in
recent years, lists of syllables, usually meaningless monosyllables, are
called over the circuits to be rated and the percentage of syllables cor-
rectly understood is taken as a measure of the circuit performance.

*“Articulation Testing Methods,” Fletcher and Steinberg, Bell Sys. Tech. Jour.,
Oct., 1929.

This testing method thus offers a means of indicating the distortion
effects of circuits in terms of the recognizability of the reproduced
sounds of speech. Probably one of its first applications* was in
determining the cutoff frequency to be used in the design of coil
loaded circuits.

The articulation testing method provides, of course, quantitative
measures in terms of the recognizability of the reproduced sounds of
speech not only of distortion effects, but also of the effects of the loud-
ness of these sounds and of the noise which may accompany them.
This method has provided a very powerful tool for investigating the
effects of changes in the reproduction characteristics of telephone
circuits on the recognition of the reproduced sounds and has been par-
ticularly useful in indicating the lines to be followed in reducing causes
of distortion in circuits and apparatus and in evaluating the impair-
ment caused by noise on telephone circuits. It has been recognized,
however, that while such measurements indicate the capabilities of the
circuits in reproducing recognizable speech sounds, they do not in them-
selves give direct measures of the degree of success which the users of
the telephone obtain in conversations where their actions are free from
the control which is necessary in articulation testing and where the
contextural relation of the words plays such a large part in their recog-
nition. ‘To make the results of this type of testing approach more
nearly the conversational results, words and sentences have been used
in place of the meaningless syllables but it is evident that even with
sentences, the control on the actions of the testers and on the ideas to be
communicated presents a condition which is quite different from those
of regular conversations.

All these ways of investigating and measuring the performance of
telephone circuits in reproducing sounds have useful applications in
present day transmission work. Frequently it is convenient to use
different methods for the various parts of a circuit in specifying the
complete functioning of the circuit in reproducing sounds.

From the standpoint of the users of the telephone circuit, the trans-
mission performance is measured by the success which they have in
carrying on conversations over the circuit. Different degrees of success
in this process may be taken as being indicated by the number of
failures to understand the ideas transmitted over the telephone and by
the amount of effort required on the part of the users to impart and
receive these ideas. Service performance is of course affected also by

RATING TRANSMISSION PERFORMANCE 125
accidental irregularities in circuit conditions such as interruptions and
cutoffs, but from the standpoint of transmission design, attention can
be concentrated on the results obtained when the circuit is in normal
operating condition. Since failures to understand and exertion of
effort are experienced also in direct conversations, their occurrence in
telephone conversations obviously cannot be entirely ascribed to the
functioning of the circuit. Variations in these factors for different
types of circuits can, however, be used as a measure of the effect of the
differences in the transmission characteristics of these circuits.

The repetitions required in a conversation can be noted but a deter-
mination of the effort factor presents dithculties. There is undoubtedly
the tendency in carrying on conversations, as in other activities, to
exert no more effort than is necessary to obtain what the participants
consider to be satisfactory results. This effort, however, will in
general be increased as the difficulty of conversing becomes greater
and so bears a relation to the increase in repetitions. Also, it is prob-
able that two dissimilar circuits which cause the same rate of repeti-
tions when used for the same service, will, on the average, call for the
same amount of effort by their users.

In line with this, the rate of occurrence of repetitions requested by
the users of a particular telephone circuit in carrying on their regular
telephone conversations can be used as a direct measure of the service
performance of the circuit. By determining the repetition rate for a
large enough number of different people at the two ends to take account
of the variability of their personal characteristics in talking and listen-
ing to the telephone and of the conversational material and conditions,
a rating can be placed on the service afforded. By making such obser-
vations on connections having different transmission characteristics,
relative ratings can be established for these various transmission
characteristics.

It should be recognized, however, that while the rate of repetitions
required can be used for relative ratings of the transmission service
performance of different circuits, such ratings in themselves do not give
a complete picture of the service from the users’ standpoint because
they do not show directly the amount of effort required. Some idea of
the effort exerted can be formed by the observers who are noting the
repetitions but this cannot be quantitative. In addition to the repeti-
tion rate and effort there is undoubtedly another factor which affects
the users’ opinion of the service. In conversing over a circuit having a
poor transmission performance, annoyance or irritation may be felt by
the users because the amount of effort required may be considered by
them to be unreasonable. These factors, by their smallness or large-

ness, may lead the users in the course of their conversations to make
favorable or adverse comments regarding the circuit performance.
These comments can be noted by the repetition observers and used,
together with any notations on effort and annoyance, to supplement the
repetition rating in arriving at a better picture of the service.

To provide for the transmission design of the telephone plant along
the lines of the previous discussion, ratings, termed “effective trans-
mission”’ ratings, are being determined which are based on the repeti-
tion rate in normal conversations. Circuits of different transmission
characteristics are considered to have the same effective transmission if
their repetition rates are equal when they are used for the same kind of
service. Furthermore, two changes in the transmission characteristics
of a circuit are taken as equivalent on the same basis. The effects of
such changes, however, are a function of the initial transmission char-
acteristics and it is therefore desirable to take as a basis for rating such
changes, a circuit which has characteristics typical in the various
respects of the ranges encountered in practice.

Asa standard reference circuit for determining an expressing effective
transmission ratings, it is proposed to use a modification of the Master
Reference System, inserting in this certain amounts of distortion,
sidetone and noise to give it transmission characteristics comparable to
those of present commercial circuits. Pending the development of this
standard reference circuit, however, use will be made of a circuit con-
sisting of station sets and instruments of kinds in general use, loops of
typical length and construction connected by typical cord circuits to a
trunk of specified transmission characteristics. For this latter it is
convenient to assume a trunk having a cutoff typical of the loading
systems in use and having a frequency characteristic which is flat below
the cutoff point. It is also convenient to assume that the attenuation
of this trunk can be varied uniformly for all frequencies below the cutoff
point. This circuit may also be assumed to deliver at the two ends a
typical amount of line noise and to have typical room noise at the
terminals. Such a circuit then specifies a complete combination of
transmission characteristics which are typical of the telephone plant in
commercial use and may be considered as a working reference circuit.
The transmission service performance of such a circuit in commercial
use can be changed by varying the attenuation of the trunk and this
attenuation, expressed in decibels with respect to some reference value,
can thus be taken as constituting a scale for expressing different grades
of service performance.

RATING TRANSMISSION PERFORMANCE 127
Starting with such a circuit, changes can be made in its transmission
characteristics such as varying the attenuation of the trunk and its cut-
off, varying the length and type of the subscribers’ loops, using different
types of transmitters and receivers in order to get different efficiencies
and kinds of distortion and changing the type of station circuit to get
different amounts of sidetone. By using circuits of these various char-
acteristics in commercial service and determining the repetition rates
obtained, a relation can be established between grade of service and
transmission characteristics both for different overall circuit combina-
tions and also for the various changes which can be made in such a
circuit. An outstanding advantage of selecting the type of circuit
which has been indicated, as a working reference circuit, is that it
readily permits direct comparisons of the service performance of the
working reference circuit, or of circuits having closely similar charac-
teristics, with the service performances of various types of commercial
circuits.

It is desirable to go one step further and to express the effects of
changes in various transmission characteristics all in terms of changes
in some one characteristic of the circuit. For this latter has been
chosen the attenuation of the trunk. In accordance with this, then,
the effect of such things as differences or changes in cutoff of the trunk,
line noise, room noise, transmitter and receiver volume efficiencies and
distortions, sidetone, and, in fact, of any transmission characteristics of
any part of the circuit can each be expressed in terms of an equivalent
change in the attenuation of the trunk on the basis of equality of effect
on service performance. Thus the ratings of all such effects can be
placed on a basis which makes them readily comparable. For the
practical range of variations in these factors it has been found that in
general the effects so expressed can be added together with a good de-
gree of approximation. Where this is not the case, interrelated sets of
effective transmission ratings can be supplied to cover the various
typical combinations which are likely to be found in practice. This
places the application of the ratings given by this method on a com-
parable basis with the application of the old volume ratings, that is,
the assignment of a number to each part of the circuit, which numbers
can be combined by algebraic summation in arriving at an overall
rating for any particular circuit.

In line with this, the effective transmission of a trunk, for example,
is rated in terms of an attenuation loss of so many db plus a rating in
db which expresses the effect of the range of frequencies transmitted
with respect to some range selected as standard, plus another rating
expressed also in db to take account of the noise on this trunk. Simi-

larly, loop loss curves can be drawn up for the combination of instru-
ments, set, loop and cord circuit such as has been used in the past on a
volume basis. On the new basis, these curves will include not only the
ratings of volume losses but also the ratings for the distortions in the
loop and instruments and the effect of the sidetone on transmitting and
receiving. In this manner, the transmission design of the plant can be
carried out in about the same manner as it has been on the volume rat-
ing basis but the effects of distortion, noise and sidetone can all be
included in these effective transmission ratings which are based directly
on service performance.

This in outline is the method of determining effective transmission
ratings which is now being worked to, its method of formulation and its
application. The complete discussion and description of these matters
involves innumerable details which, as already stated, it is not the pur-
pose to set forth here. From this outline it is seen that this method
provides the following outstanding things:

1. A scale for indicating different grades of effective transmission,
which scale is expressed in decibels and is directly correlated
with service performance by means of a typical circuit selected
as a reference. This permits the specification of grades of
service.

2. The use of this same scale as a means of assigning to each element
of practical telephone circuits an index, expressed in decibels,
which measures its contribution to the effective transmission of
the circuit, these indices being of such a nature that those cor-
responding to the elements in a circuit can be combined in a
simple way to give an overall performance index for that circuit.
Such a system of indices is necessary for plant design.

3. A means of correlating effective transmission service and circuit
transmission characteristics. This correlation is advantageous
in setting up the indices of (2) and in development and design
work in determining the desirability of possible changes in the
performance of the various elements.

The selection for the present of the typical practical circuit described
above, as a working reference circuit, has two important advantages,
which will be restated. First, by using a reference circuit having
typical transmission characteristics, the indices established for changes
in the various characteristics within the range of practical interest, are
directly applicable to the present plant and can be combined in a
simple manner to provide an overall circuit index. Second, and by no
means of minor importance in the earlier stages of the application of the

it is possible and practicable to make direct comparisons of the service

performance of the reference circuit, or circuits having closely similar
i characteristics, with the performances of various commercial circuits.
The maintenance of the first advantage will require, however,

changes in the working reference circuit as material improvements are

made in the transmission characteristics of the commercial circuits.
To obtain the second advantage means the use at present of carbon
transmitters in the working reference circuit. These are open to the
same objection here as they were in the Standard Cable Reference
System, namely, the difficulty of exactly specifying their performance
raises questions as to the reproducibility of their performance from
time to time. This was one of the major reasons for the replacement
of the Standard Cable Reference System as the basis for volume ratings
by the Master Reference System for Telephone Transmission with its
specifiable performance. To preserve the first advantage mentioned
and at the same time to obtain a reference system whose reproducibility
can be assured, it is the purpose, as more complete correlations are ob-
tained between transmission characteristics and service performance,
to associate with the Master Reference System, the means to make its
transmission characteristics meet the requirements necessary to retain
the first advantage. Meanwhile the Master Reference System will
continue its function as a reference for volume ratings.

To provide the basis for such a system of effective transmission
ratings as has been outlined, several series of tests have been made, the
most comprehensive of which has been under way for more than a year
between several hundred stations in the American Telephone and
Telegraph Company headquarters building and a similar number of
stations of the Bell Telephone Laboratories, between which there is a
large amount of intercommunication. The connections between these
stations are handled over special trunks in which the attenuation, cut-
off frequency and line noise can be varied. At the stations, different
types of instruments and station circuits have been employed. Ob-
servers are connected to each of these trunks who monitor the conversa-
tions over them and note the number of repetitions requested in each
conversation and also the duration of the conversation. In this way is
determined the repetition rate for a number of conversations between a
number of different people for the various combinations of circuit
characteristics so provided. Thus ratings are established directly of
such effects as those of trunk cutoff, noise on the trunk, different types

of transmitters and receivers and of variation of sidetone in the station
set. In addition to the observations of repetitions, measurements are
made of the talking levels on the trunks by means of volume indicators
to determine the reaction of the circuit performance on talking levels.

An illustration of the results of such observation is given in Fig. 1.
The curve shows the variation of the repetition rate with change in
trunk attenuation for connections having the same kinds of terminal
sets and loops at both ends. This then provides a means of expressing
different grades of service performance in terms of trunk attenuation in
this circuit.

On this figure is shown also the repetition rate obtained for trunks of
two different effective upper cutoff frequencies. The change in trunk

attenuation required to produce the same increase in repetition rate as
that obtained im going to point C, for example, from the corresponding
1,000-cycle attenuation point on Curve A, is taken as the rating in db of
the lower cutoff frequency represented by point C. This rating is
about 5 db. The rating of point B with respect to A is obtained in a
like manner to be about 1 db and correspondingly the rating of the cut-
off frequency of C with respect to B is about 4db. This illustrates the
manner of obtaining effective transmission ratings for any change from
the characteristics of the circuit of Curve A.

It is obviously laborious to cover the ranges of all the transmission
characteristics of circuits of this kind. The idea is to establish points
which will cover the practical range and to use the results of articulation

tests and other similar measurements for interpolating between the
points established by the repetition method. In this way it is planned
to put the rating of transmission on a basis which indicates the effect on
service of changes in the various parts of the circuit.
CONCLUSION

In concluding, it may be restated that the primary purpose here has
been to discuss the rating of the transmission performance of telephone
circuits and the method which is being adopted in the Bell System for
determining and expressing effective transmission ratings for the design
of the plant. The salient features of this method which should be
emphasized are the following:

1. In establishing the rating of the transmission performance of a tele-
phone circuit, its performance is taken as that obtained when
the circuit forms part of the combination of talker, circuit and
listener, where the talker and listener represent the users of the
telephone system in commercial service.

2. The ratings of the effective transmission of circuits are based on the
rate of repetitions required.

3. The ratings of effective transmission will eventually be referred to a
modification of the Master Reference System arranged with
typical distortion, sidetone and noise. For a working reference
circuit, use is made of a circuit which has transmission charac-
teristics typical of those encountered in service. The trunk of
this circuit is taken as adjustable in attenuation for the purpose
of providing a scale for specifying different grades of overall
transmission performance and also for expressing ratings of the
effect on transmission performance of the various transmission
characteristics of circuits and their parts.

Gutta percha and balata have proven eminently suitable for the in-
sulation of long deep sea telegraph cables, but their dielectric losses are too
high to meet the requirements of submarine telephone cables designed to
operate over long distances or of shorter cables employing carrier currents.

This paper describes a new material called paragutta which has been
developed to meet the present needs. It consists essentially of the purified
hydrocarbons of balata (or gutta percha) and of rubber together with minor
quantities of waxes to modify the mechanical characteristics. The puri-
fication of rubber partic ularly with respect to nitrogenous constituents
is necessary to effect electrical stability in water. A commercially usable
method of purifying rubber is described.

Evidence is furnished that paragutta has all of the desirable thermo-
plastic and mechanical properties of gutta percha while possessing such
superior insulation characteristics as to make it suitable for use on long
cables designed for transoceanic telephony. Its use is also advantageous
on shorter deep sea cables designed for carrier telephony as well as for
ocean telegraphs.

purposes but in recent years there has been an increasing use of
this type of cable for telephone service. Telephonic communication
requires cables of very much superior transmission quality to that
needed for telegraph. At the higher frequencies of voice transmission
the energy losses in the insulating material become a serious factor
and a radical improvement in submarine insulation is called for.

The longest existing deep sea cables operating at voice frequency
only slightly exceed 100 miles and the construction of a transoceanic
telephone cable with standard materials has been regarded as beyond
the practical limits of feasibility.

The installation and rapid expansion of transatlantic radio telephony
during the past few years have created a need for a deep sea telephone
cable to supplement this service, particularly during periods of at-
mospheric disturbances. In addition the development of carrier
telephony offers possibilities for increasing the traffic over shorter
submarine cables. For the shorter cable, the still higher frequencies
of carrier telephony make demands upon the insulating material
similar to those of long cables operating at voice frequency.

In view of these circumstances an extended study was undertaken
of the causes of losses and other electrical weaknesses of submarine
insulation and a search has been made for better materials. As a

By virtue of its superior electrical properties, the use of paragutta
in place of gutta percha for the insulation of telephone and telegraph
cables offers advantages either from the standpoint of improved trans-
mission or the economies in materials of construction which can be
made as a result of modified design.

Gutta percha and balata have been the standard materials for the
insulation of deep sea cables since the inception of the submarine
cable industry some seventy-five years ago. Although these sub-
stances are inadequate for modern telephone needs as regards their
electrical characteristics, their mechanical properties are peculiarly
adapted to submarine insulation. This is so much the case that gutta
percha can fairly be taken as a model which must be closely imitated
in respect to mechanical characteristics by any successful substitute.
This is fortunate since the use of any substitute which differs radically
from gutta percha would mean discarding large existing investments
in special technique, equipment and trained personnel, and would
involve serious risks as to the integrity of cables made with the new
material. It may be remarked in passing that no manufacturing
process requires a higher degree of insurance against occasional defects
than does the submarine insulation art, a fact that has engendered a
strong conservatism in the industry.

Because of its almost ideal mechanical properties, the requirements
for submarine cable insulation may conveniently be described by
reference to gutta percha. Gutta percha insulation, which often
includes more or less balata in its composition, is made of raw materials
carefully selected for quality, which are thoroughly washed and
extremely uniformly blended. The thermoplasticity of the material
is of great service in these operations and further permits it to be
readily extruded onto a conductor in multiple layers in a continuous
sheath with great exactness and freedom from mechanical defects.
After being forced around the conductor the material quickly sets to
a hard, tough covering when drawn through cold water. Its firmness
and toughness are essential to resist subsequent handling operations in
the factory, as well as those involved in laying, picking up and re-
pairing. The warm, soft material adheres readily to the conductor
and is well adapted to the making of joints in the insulation between
core lengths both in the factory and on the cable ship.

In addition to those excellent and unique mechanical properties,
gutta percha possesses electrical characteristics peculiarly adapted to
submarine cable construction. Its outstanding electrical merit con-
sists in the fact that its electrical characteristics are stable under sea
bottom conditions over a great many years.

Gutta percha is obtained from the latex of a large number of species
of trees growing wild in the forests of the Malay Peninsula and the
East Indian Islands. The products of the various species of trees are
by no means of equal value, varying as they do in the content of
hydrocarbon, resins, moisture and other substances. Since the ma-
terial is gathered and worked up upon the spot by primitive people a
great deal of carelessness as well as deliberate adulteration is practiced
and the material comes upon the market in a dirty condition and in a
bewildering variety of forms which almost prohibit effective inspection,
standardization and grading.

Balata comes from two species of trees of the same general botanical
family as gutta percha, but is native to the forest regions of upper
South America and is unknown in the gutta percha producing area
of the Far East. The latex of the balata tree is more fluid than that
of gutta percha, which permits the trees to be tapped and the fluid to
be collected at a central point in the forest, where the product from
various trees is mixed for recovery of the gum. Because of the small
number of species involved and the transportability of the fluid latex,
balata is produced in a much more limited number of grades and is
cleaner and more dependable as to uniformity of quality. Its essential

constituent is the same hydrocarbon which gutta percha contains.
In addition to the hydrocarbon, there is present in balata some 40
per cent of resins and amounts of dirt, moisture and other impurities
which usually total about 15 per cent. ‘The resins of balata are softer

H than those of gutta percha and make the product in its raw state a
little less desirable than the better grades of gutta percha from the
; mechanical standpoint. Balata, however, contains a smaller amount

of finely divided dirt or humus than gutta percha, which is reflected
in its superior electrical characteristics and lower water absorption.
The resins of both gums have been usually included with the hydro-

5 carbon in making submarine insulation. Sometimes, however, a
: portion of the resins are removed, partly to increase the toughness and
' partly to improve the electrical characteristics.

There are several methods which may be used for preparing gutta
hydrocarbon nearly free from resinous substances. One of these
methods involves dissolving the balata or gutta percha in warm
petroleum naphtha, filtering the solution from dirt and precipitating
the gutta hydrocarbon from solution by refrigeration, leaving most of
the resins in solution. A simpler and less expensive method, however,
is that of leaching out the resins by simply soaking the sheeted or
finely cut material in a suitable grade of petroleum naphtha at ordinary
temperature, followed by draining off the solution of resins and finally
evaporating the residual solvent from the extracted material.

The completely deresinated hydrocarbon from either source is not
suitable for use alone as submarine cable insulation because insuffi-
ciently plastic at safe working temperatures, as well as prohibitively
expensive. Otherwise the complete deresination of these products
would be highly advantageous as, for example, is indicated by the
superior electrical characteristics of deresinated balata shown in
Table 1. A substantial amount of experimentation upon the methods
of refining balata has been necessary to secure the excellent electrical
characteristics therein indicated but no revolutionary innovation has
been necessary.

TABLE I
EFFECT OF RESIN CONTENT ON THE ELECTRICAL CHARACTERISTICS OF BALATA
Electrical Characteristics 0° C., 1 Atm.,

In attempting to develop a new insulating material for deep sea
cables it seemed best to begin with gutta hydrocarbon as a basis,

since its mechanical properties are so unique, rather than to attempt
to synthesize a new chemical compound which would imitate it. In
order to overcome the excessive stiffness of the pure gutta hydrocarbon,
as well as its prohibitive cost, it was determined to attempt to blend
large quantities of rubber with it, since rubber is the nearest kindred
material and is commercially available at low cost. There resulted
thermoplastic products of fairly good mechanical characteristics which,
however, proved to be insufficiently stable electrically.

Meanwhile, a thorough study was being made of the electrical and
physical characteristics of rubber and particularly of the causes of its
electrical instability upon prolonged immersion in water. Our hope
that such a study would not only reveal the nature of the defects of
rubber but also suggest means for remedying them has been realized
to a gratifying degree.

Rubber, as is well known, is also derived from the latex of certain
trees, chiefly Hevea Brasiliensis. This tree has been cultivated in
large areas on the plantations in the Far East and the product is
obtainable commercially in excellently standardized grades. Its
principal constituent is a hydrocarbon scarcely distinguishable from
that of gutta percha by chemical means, but radically different from
it in physical properties, notably in that it has but a slight degree of
thermoplasticity and is far more distensible in the cold state. Aside
from the hydrocarbon, rubber also contains small amounts of resins,
proteins and other impurities, but the aggregate non-hydro-carbon
constituents in the better grades are usually less than 10 per cent in
contrast to 50 per cent or thereabouts for gutta percha and balata.

Rubber is used almost exclusively in industry in a vulcanized form,
that is, in combination with a small percentage of sulphur. In this
form rubber has also been used to a limited extent for submarine cable
insulation, but has long been recognized as lacking sufficient electrical
stability for deep sea cables designed to carry a heavy traffic. It is
still used to a considerable extent with a fair degree of success for
insulation on short cables where the electrical requirements are not
severe. In tropical waters it has the advantage over gutta percha of
greater resistance to teredo attack and to damage by high temperature.

Some years ago an extended study ! was made of the causes of the
electrical instability of vulcanized rubber, which led to the conclusion
that the water soluble impurities are largely responsible. These
impurities can be removed comparatively readily and satisfactorily
in the process of manufacture, and a submarine insulation of a fair
degree of stability is thereby attained.

Even so, vulcanized rubber is very inferior to gutta percha for
submarine insulation as the necessary manufacturing operations are
more difficult and likely to lead to defects. ‘The removal of mechanical
impurities is by no means simple because the raw stock is not plastic
enough for thorough straining. The lack of plasticity also interferes
with multiple covering of conductors, and the process of heating to
bring about vulcanization is liable to result in deformation of the
insulating layers. The joining and repairing of core lengths insulated
with rubber is also more of a problem than with gutta percha, which
can be so readily remolded in case imperfections appear in the course
of the process.

The methods of electrical stabilization of vulcanizable rubber
compositions are only partially effective in the absence of vulcanization
and it was therefore necessary to extend the study in an effort to
secure the desired electrical properties in rubber in the raw state.
It might be supposed that mere admixture of raw rubber with gutta
hydrocarbon would produce the necessary stability. This is true
only to a limited extent. When the proportions of rubber are high
enough to meet the mechanical and economic requirements, the
electrical stability is impaired.

EFFECT OF PROTEINS ON ELECTRICAL STABILITY OF CRUDE
RUBBER IMMERSED IN WATER

It has been previously shown that crude rubber contains consider-
able water soluble impurities and that their removal results in a
large reduction in water absorption.':?» Rubber so prepared absorbs
no more water than good cable gutta percha but in a raw state when
immersed in water, it fails sooner or later as an insulator, often sud-
denly and completely.

To determine the reason for this electrical ifstability of crude rubber
in water, samples of very pure rubber hydrocarbon completely freed
from proteins, resins and other impurities were prepared and tested.
It was found that this material not only absorbed very little water
but showed practically no change in electrical characteristics as a
result of prolonged immersion in water. The impurities natural to
rubber therefore seem to be responsible for its instability.

It has been known for many years that crude rubber contains
proteins, ordinary plantation rubber containing about 3 per cent.
Previous investigators have postulated and shown considerable
indirect evidence to the effect that the rubber globules in rubber latex
have an adsorbed film of protein around them and that this condition

also exists in crude rubber. It is also known that latex serum contains
a substantial quantity of protein in solution. The preparation of
crude rubber from latex by addition of acid or by processes of evap- :
oration of the water by heat undoubtedly results in the precipitation
of considerable quantities of this protein which becomes entrapped
between the globules as they coalesce. It is easy then to visualize
that in crude rubber there exists a continuous phase of protein or a
protein network which, acting like most protein matter, absorbs large
quantities of water, resulting in paths through which electrical con-
duction occurs.

The problem of developing a suitable commercial method for
preparing rubber free from nitrogenous matter offered many apparent
difficulties. The proteins are colloidal in nature and in the presence
of water form gelatinous masses rather than true solutions. On this
account they often cannot be removed by simple washing as can be done
in the case of gutta percha and balata. It has been known for some
time that proteins can be broken down to water soluble products by
boiling with dilute hydrochloric or sulphuric acids. This treatment
did not produce satisfactory results in the case of rubber. As a result
of many experiments involving a variety of methods, it was found that
heating rubber in an autoclave at an elevated temperature in the
presence of water alone fairly rapidly brought about the desired hy-
drolysis of the rubber proteins, converting them to water soluble
materials. Asa result of subsequent washing, it was found that the
nitrogenous bodies had been almost completely eliminated without
deleterious effect on the rubber hydrocarbon.

Rubber either in the form of sheets immersed in water or as an aque-
ous rubber dispersion such as latex can be employed in the process.
The treatment of latex, however, results in a more rapid hydrolysis
of the proteins. Considerable latitude exists in the choice of condi-
tions, but the following example will suffice to describe one method of
carrying out the process: ammonia preserved latex is diluted 1 to 5
with pure water. The latex is then heated in an autoclave for approx-
imately ten hours at 150° C. After cooling it is coagulated with
acetic acid and thoroughly washed. As a result of this treatment
the nitrogen content of the rubber is found to be less than 0.10
per cent, which is about one fourth that of ordinary plantation crude
rubber. Figures 1, 2, 3 and 4 illustrate the relative water absorption
and electrical stability of deproteinized rubber as compared with the
ordinary crude product. Vulcanized deproteinized rubber was also

= found to be somewhat superior to ordinary vulcanized crude rubber
of as regards its electrical stability in water.
i In addition to the superior electrical stability of deproteinized rubber,
m ; it was found to be more readily plasticized and mixed with gutta
ze 40
a
3.5
4
> 3.0 +
WwW
|
4 a
| |
z 20 amt
r < | |
| |
I
Le |
DEPROTEINIZED
05 + RuBBER =
= | |
| | it
) 2 3 4 5
t TIME IN WEEKS
{ Fig. 1—Effect of washing and removal of protein on the water absorption of crude
rubber when immersed in 3.5 per cent NaCl solution at room temperature.
|
$ 2 35 4 4 +
|
< / | / |
: > 25 / / t
|
‘ |
20 t 4
SMOKED | /
i SHEET | WASHED |
= |
| |
wl | | |
a DEPROTEINIZED |
RUBBER
2 3 4 5
TIME IN WEEKS
Fig. 2—Effect of washing and removal of protein on the dielectric constant of crude
rubber when immersed in 3.5 per cent NaCl solution at room temperature.

than is the case with crude rubber, thereby yielding a product with
better thermoplastic properties.

As previously stated, the principal con ituents of paragutta are
deproteinized rubber and purified gutta hydrocarbon. Specially
treated hydrocarbon or montan waxes may also be added as a third
constituent to modify mechanical properties and reduce cost. The
proportions of these constituents may be varied over a wide range to
achieve the desired characteristics, but in general rubber and gutta

Fig. 3—Effect of washing and removal of proteins on resistivity of crude rubber,
when immersed in 3.5 per cent NaCl solution at room temperature.
are used in about equal proportions and purified montan wax may be
added up to about 40 per cent. Superior electrical properties, how-
ever, result from the use of hydrocarbon waxes, which may be added
in amounts up to about 20 per cent. By the proper blending of these
materials, a thermoplastic insulation is obtained which closely ap-
proximates gutta percha in mechanical properties and is fully its equal
as to electrical stability in water. Its specific electrical characteristics

represent a substantial ir rovement over those of the classical insula-
tion compounds and its st is lower.

The final steps in pro ssing paragutta are very similar to those
used for gutta percha and involve blending and washing the depro-
teinized rubber and deresinated balata or gutta together, masticating
to remove excessive water and at the same time incorporating such
waxes as are found necessary. The material is then strained through
fine sieves under hydraulic pressure to remove adventitious impurities,
kneaded to remove air and finally placed on the covering machine rolls
to be forced around the conductor. The machinery in use for pro-

Fig. 4—Effect of washing and removal of proteins on conductivity of crude rubber
when immersed in 3.5 per cent NaCl solution at room temperature.

cessing gutta percha is suitable for handling paragutta in these
operations.

Tensile Properties: Although submarine insulation is not subjected
to tensile deformation in practice, tensile properties indicate to some
degree the relative mechanical suitability of a given material for the
purpose. Figure 5 shows the stress-strain characteristics of paragutta
and gutta percha submarine cable insulation. These results show

that paragutta has tensile properties equal to cable gutta percha
although its gutta content is substantially lower.

Compression Properties: The insulated submarine cable conductor
commonly known as the core is frequently subjected to uneven com-
pression stresses during manufacture, laying and repairing. The
insulation must therefore be capable of withstanding these stresses
without appreciable deformation. ‘To determine the relative merits
of paragutta and gutta percha in this respect their comparative stress-
strain characteristics under compression have been measured, using a
special compression machine,’ and are shown in Fig. 6. In this test

a steel rod 1.6 cm. in diameter was forced endwise into a sheet of the
material .375 cm. in thickness at a rate of about 4 cm. per minute
while simultaneously recording the deformation and load. These
results show that very little difference exists between these materials
in this test, and factory handling of cores confirms the general con-
clusion.

Flexibility: The flexibility of submarine cable insulation is important
because the core is subjected to considerable flexing during manu-

facture, laying and repairing and possibly at times during use, espe-
cially where tidal currents may cause movement in the cable. Para-
gutta and gutta percha cores have been subjected to slow and con-
tinuous flexing at 0° and 25° C. for long periods and it was found that
both materials will withstand millions of repeated flexures at small
amplitudes without failure. When the amplitude of flexure was
increased to strain the conductor slightly beyond its elastic limit,
the conductor always failed in advance of the insulation.

Plasticity Tests: Laboratory tests were made to determine the rela-
tive plasticity of paragutta and gutta percha, using both the Williams *

Fig. 6—Comparative compression properties of paragutta and gutta percha at 25

and the Marzetti® type of plastometers. These tests are valuable
guides but the final judgment of a material as regards thermoplasticity
was made by determining its workability on commercial gutta percha
insulating machines. Paragutta is somewhat more resistant to flow
than gutta percha at temperatures ranging from about 40° to 70° C.
When applied to the conductor, however, its greater resistance to flow
at elevated temperatures can be taken as an advantage as it lessens
the danger of faults occurring if the core should be accidentally exposed
to elevated temperatures or to conditions which might exist in con-
nection with cable used in the tropics.

Figure 7 shows the relative plasticities of cable gutta percha and
paragutta at several temperatures as determined by the Williams 4
method, which can be taken to indicate the relative plasticities of
these materials at working temperatures.

Brittle Temperature: It is extremely important that the temperature
at which submarine cable insulation becomes brittle should be far
below the range of sea bottom temperatures to be encountered in use.
This is one of the properties in which rubber and gutta percha greatly
excel any other available insulating material. Kohman and Peek °

TEMPERATURE -DEG. C.
Fig. 7—Effect of temperature on the plasticity of cable gutta percha and paragutta.

The amount of water absorbed by rubber and gutta percha when
immersed in water is the result of a complicated mechanism. ‘The
quantity and nature of water soluble or water absorbing impurities

TABLE II
BRITTLE TEMPERATURE OF PARAGUTTA AND OTHER INSULATING MATERIALS

Material (
Gutta Percha (Cable Insulation 23 to —36
Paragutta 45 to
Balata (Washed) 44 to —52
Balata (Washed and Deresinated 62 to —67
Crude Rubber 57 to —58
Vulcanized Rubber (Soft) —§3 tp — 58

in the rubber or gutta percha and the salt concentration of the water
in which the samples are immersed are controlling factors. The
enormous increase in the quantity of water absorbed by ordinary
rubber when immersed in distilled water as compared with its absorp-
tion in salt solutions has been explained on the basis of osmotic
theory.!. In accordance with this theory rubber acts as a semi-
permeable membrane. Water soluble crystalloids or hydrophillic
colloids (proteins) attract the water which enters the rubber by
diffusion. When immersed in distilled water these impurities tend
to reach infinite dilution with water, being opposed in this by the
resistance of the rubber itself to swelling. In salt solutions the
amount of water absorbed is finite and depends on the equalization
of osmotic pressures of the internal and external solutions. The
change in water absorption of pure rubber hydrocarbon with the salt
concentration of the external solution is small over the whole range,
which indicates that the water enters by a process of solution. This
has also been found to be the case for gutta hydrocarbon and is more
or less true for paragutta and gutta percha. The water absorption
in distilled water can therefore be taken as a measure of the freedom
from water soluble or water absorbing impurities. Figure 8 shows
the effect of NaCl concentration in the immersion solution on the
quantity of water absorbed by samples of rubber, paragutta and gutta
percha at room temperature. Samples of rubber containing water
soluble matter or proteins do not readily reach an equilibrium water
content in distilled water. Crude rubber has been found to absorb
more than 100 per cent water in distilled water at ordinary temperature
without reaching equilibrium.! Gutta percha, paragutta and pure
rubber hydrocarbon on the other hand reach a definite and lower
equilibrium water content in distilled water, which shows their greater
freedom from water soluble or water absorbing matter.

As the electrical stability of paragutta in sea water is of paramount
importance an exhaustive study has been made on a large number of
specimens as regards their changes in electrical values over long periods
of immersion in 3.5 per cent salt solution. Gutta percha insulation

CONCENTRATION NaCl IN IMMERSION SOLUTION-PER CENT
Fig. 8—Relation of water absorption to salt concentration in immersion solution,

The overall quantity of water absorbed, however, cannot be used
as a final criterion by which to judge insulation for it has been pre-
viously shown (Figs. 1 to 4) that washed crude rubber completely
fails as an insulator after absorbing less than one per cent water. The
mode of distribution of water absorbing impurities in an insulating
material has been found to be of utmost importance as regards the
magnitude of the effect of moisture in various insulating materials.
Examples where large effects on insulating properties are caused as a
result of moisture absorption by localized impurities are found in the
above case of proteins in crude rubber, water soluble salts associated

with fillers in vulcanized rubber ! and hygroscopic salts on the surfaces
of textile fibers.’

On the other hand, the electrical properties of paragutta or gutta
percha are not impaired when several times their equilibrium water
content is incorporated with them. Gutta percha, however, does
show an increase in capacitance of about 10 per cent as a result of
water absorbed by a completely dried specimen, but as it is always
the practice to apply it to the conductor in a wet condition this

Fig. 9—Changes in water content of 50 mil wet and dry paragutta and gutta percha

change is not of practical significance. The electrical properties of
paragutta on the other hand show practically no changes as a result
of moisture absorption by a dry sample. ‘These facts are taken to be
the best evidence of the electrical stability of paragutta in contact
with water.

Hundreds of specimens of paragutta and gutta percha have been
studied as regards changes taking place in electrical characteristics
after long periods of continuous immersion in 3.5 per cent salt solution.
These tests, some of which have been for periods of three to five years,
show that paragutta is fully equal to gutta percha as regards its

stability. When properly prepared both of these materials show
practically negligible changes in electrical properties as a result of
prolonged submergence in water. Sea bottom conditions are even
less likely to affect these materials than those existing in the laboratory.
This is because of the absence of light, limited oxygen supply and
low temperature, all of which reduce the tendency of materials such as
paragutta or gutta percha to oxidize or otherwise deteriorate. It
has also been shown? that the low temperature and high pressure
existing at sea bottom reduce the rate of water absorption but do not
materially affect the amount absorbed.

Electrical Characteristics: The electrical properties of paragutta
depend upon the particular composition chosen, the quality of the
raw materials and the care exercised in processing them. For long
telephone cable insulation, it is necessary to exercise the utmost care
to obtain a material having dielectric constant and specific conductance
values suthciently low to reduce to the minimum its effect on the
attenuation. On the other hand, for ordinary telegraph cables these
values are less critical and it may be advantageous to modify the
practice for purposes of economy. Representative values for the
electrical properties of a superior grade of paragutta and typical cable
gutta percha under sea bottom conditions are given in Table III. It
will be seen in this table that paragutta has a 20 per cent lower dielec-
tric constant and a specific conductance one-thirtieth that of ordinary
cable gutta percha under sea bottom conditions. The insulation
resistance and dielectric strength of the two materials are practically
the same.

COMPARATIVE ELECTRICAL PROPERTIES OF PARAGUTTA AND CABLE GUTTA PERCHA
AT SEA BottOoM CONDITIONS

Specific Inductive Effective A-C
Capacity 2° C., Conductivity 2° C.,
400 Atm., 2000 Cycles 400 Atm., 2000 Cycles
Unit = 10712 mho. cm.
Cable Gutta Percha. ..
3

The author wishes to acknowledge his indebtedness to Mr. R. R.
Williams for counsel and assistance during the prosecution of the
work and writing of the paper.

An Efficient Loud Speaker at the Higher Audible Frequencies. L. G.
Bostwick. This paper describes a loud speaker designed for use as an
adjunct to existing types for the purpose of extending the range of
efficient performance to 11,000 or 12,000 cycles. A moving coil piston
diaphragm structure is used in conjunction with a 2000-cycle cutoff
exponential horn having a mouth diameter of about 2 inches. Mo-
tional impedance measurements on this loud speaker indicate an aver-
age absolute efficiency of about 20 per cent within the frequency range
from 3000 to 11,000 cycles. The variation in response within this
band does not exceed 5db. By using a high-frequency loud speaker of
this type the efficiency and power capacity of the associated low-fre-
quency loud speaker can be improved and a uniform response-fre-
quency curve from 50 to 12,000 cycles can be obtained.

Results of Notse Surveys. Part I. Noise Out-of-Doors2 ROGERS
H. Gat. The purpose of a noise survey of a locality is to study the
space and time distribution of noise intensity, the frequency composi-
tion of the noise, the contributions of various noise sources, the relation
between the annoyance effect of the noise and its physical and auditory
characteristics, and the effectiveness of methods of noise reduction.
The extent to which each of these phases of the noise problem has been
investigated heretofore has depended upon the point of view of the
investigator and upon the apparatus employed. From one standpoint
or another, any audible sound may fall within the category of noise;
hence the variety of possible noise surveys is almost unlimited. Not
many such surveys have been carried out, however, partly because the
appropriate apparatus is of recent development; nor has any extensive
comparison been published between the results obtained in different
places and with different instruments. It has therefore seemed worth
while to assemble such previously published results as are available,
and certain new observations, in the present series of papers, of which
this paper deals with noise out-of-doors.

contact behavior of granular carbon of the type used in commercial
microphones.

Through a study of the temperature coefficients of resistance of such
contacts it is possible to conclude that the conducting portions of the
contact junctions are of the nature of carbon and that new contact
points are established or broken when the resistance is varied in a
reversible resistance force cycle.

The experiments show that for such reversible cycles the relation
between the resistance and force is of the approximate form
R= K(F)-". The exponent 7 varies considerably from cycle to cycle
but its average value depends on the force limits. The largest values
of n are obtained with the aggregates of granules under such conditions
of force limits that the elastic strains must be relatively large. A
maximum mean value substantially independent of the force limits
over a wide range closely approximates the value 7/9.

This value 7/9 is the maximum given by a theory of contact resistance
worked out by F. Gray, assuming that the contact is made between
two spheres of conducting material having surface roughness equivalent
to an assembly of minute spherical hills. On account of the elasticity
of the material both the microscopic area of contact between the spheres
and the microscopic areas of contact between the hills increase with
contact force. A strained aggregate of granules may therefore be made
to behave like an ideal single contact between spheres having a rough
surface.

For single contacts and for aggregates at small strains the value of 7
falls below the minimum value 1/3 which is accounted for by theory.
This is associated with internal contact forces, or cohesion, which render
the contacts relatively insensitive to changes in the applied force.
The existence of cohesion is readily demonstrated by the fact that
contacts always require a finite force to break them even when no
current has passed through the contact.

Metallography, at first thought, appears wholly unrelated to his-
tology or other branches of biology but the two branches of science do

have many points incommon. Both deal in the last analysis with the
structure of matter and, in each, the microscope is an indispensable
tool. Improvements in microscopic vision which enlarge the world of
vision in one branch of science inevitably have a reflection in the other.

It is not the purpose of this paper to enter into a discussion of struc-
tures of living cells as revealed by the ultra-violet microscope. More
particularly, the object is to present a tool for biological research; a
tool which enables us to photograph the structure at different planes or
levels within a single cell or group of cells; one which enables us to see
the living cell with a degree of precision and clarity not heretofore
possible” by any other known means and with a potential resolving
ability at least twice that of the best apochromatic system using visible
light.

Production of Plustic Molded Telephone Parts’ A.M. Lynn. The
Western Electric Company now manufactures for Bell System ap-
paratus a large number of different phenol-plastic, shellac, and hard-
rubber molded parts, the output of which varies from a few thousand
to several million per year. The majority of these molded parts are
produced in comparatively small quantities, but certain of them, such
as the phenol-plastic molded parts used in the hand-set type of tele-
phone, a new molded subscriber’s set housing, and the receiver shell,
cap, and mouthpiece used on the older type of desk-stand telephone,
are heavy-running parts. The tools and press equipment used in the
production of these parts are described in this paper.

Variation of the Inductance of Coils Due to the Magnetic Shielding
Effect of Eddy Currents in the Cores® Kk. L. Scott. An analysis is
made of the shielding effect of eddy currents on the flux in the interior
of cores of cylindrical or flat sheet material. It is shown that the
counter voltage of self inductance of an iron-cored coil is due only to
the component of flux in the core which is in phase with the flux at
the surface of the core. Expressions are obtained and curves plotted
showing the variations of inductance of a coil with frequency, or with
the conductivity and permeability of the core material. Sample
calculations and some experimental results are given. The results
show that the inductances at high frequencies are actually less than
the predicted values, which leads to the suspicion that some factor
other than eddy currents causes the flux in the interior of the cores to
decrease with increasing frequency.

Results of Noise Surveys. Part IT, Noise in Buildings? R. S.
TucKER. Noise experienced indoors is in one sense more important
than that experienced outdoors, for, with the growth of our industrial
civilization, increasing numbers of people are spending most of their
waking hours indoors. They are thus exposed to indoor noise for a
large part of the time, including the hours of work when noise has its
opportunity to impair their working efficiency.

Certain typical values for noise in various locations in buildings
have been published, and are summarized in this paper. Our knowl-
edge of indoor noise levels is far from complete, however. Further
information has been obtained in a survey of room noise in New York
City and the surrounding area which was made in 1929 by the National
Electric Light Association and the American Telephone and Tele-
graph Company in the course of the work of their Joint Subcommittee
on Development and Research. Some results of the New York City
measurements are given. About 70 test locations are included. It
will be realized that this is only a small sample of the total number of
places where indoor noise is experienced in New York City alone. The
conclusions given must therefore be regarded only as suggestive rather
than as holding true in any general sense.

CHARLES B. AIKEN, B.S., Tulane University, 1923; M.S. in Electrical
Communication Engineering, Harvard University, 1924; M.A. in
Physics, 1925. Bell Telephone Laboratories, 1928-. Mr. Aiken has
been engaged on work in connection with aircraft communication and
more recently with the design of broadcast radio receiver equipment.

F. E. Hawortn, A.B., University of Oregon, 1924; M.A., Columbia
University, 1929; Bell Telephone Laboratories, 1925-. Mr. Haworth’s
work has been in crystal analysis by means of X-rays, magnetic mate-
rials, and more recently in studies of dielectrics.

HERBERT E. Ives, B.S., University of Pennsylvania, 1905; Ph.D.,
Johns Hopkins, 1908; assistant and assistant physicist, Bureau of
Standards, 1908-09; physicist, Nela Research Laboratory, Cleveland,
1909-12; physicist, United Gas Improvement Company, Philadelphia,
1912-18; U.S. Army Air Service, 1918-19; research engineer, Western
Electric Company and Bell Telephone Laboratories, 1919 to date.
Dr. Ives’ work has had to do principally with the production, measure-
ment and utilization of light.

W. C. Jones, B.S. in E.E., Colorado College, 1913; Western Elec-
tric Company, 1913-25; Bell Telephone Laboratories, 1925-. As
Transmission Instruments Development Engineer, Mr. Jones has
specialized in the development and application of instruments for the
transmission of speech and music.

A. R. Kemp, B.S., California Institute of Technology, 1917, M.S.,
1918; Engineering Department, Western Electric Company, 1918-25;
Bell Telephone Laboratories, 1925-. Mr. Kemp has been engaged in
chemical research on rubber and allied materials used for submarine
and other types of insulation.

W. H. Martin, A.B., Johns Hopkins University, 1909; B.S., Mas-
sachusetts Institute of Technology, 1911; American Telephone and
Telegraph Company, Engineering Department, 1911-19; Department
of Development and Research, 1919-. As Local Transmission En-
gineer, Mr. Martin has been engaged in development work on the
transmission of telephone sets and local exchange circuits, transmission
quality and loading.

L. J. Sivran, A.B., Cornell University, 1916; Engineering Depart-
ment, Western Electric Company, 1917-19 and 1920-25; Bell
Telephone Laboratories, 1925-. Mr. Sivian’s work is in acoustics,
chiefly in connection with methods of electroacoustic measurements.

GEORGE C. SoutHwortu, B.S., Grove City College, 1914, M.S.,
1916; Ph.D., Yale, 1923; assistant physicist, Bureau of Standards,
1917-18; instructor, Yale University, 1918-23; Information De-
partment, American Telephone and Telegraph Company, 1923-24;
Department of Development and Research, 1924-. Mr. South-
worth’s work in the Bell System has been concerned chiefly with
the development of short-wave radiotelephony. He is the author of
several papers on radio-frequency phenomena.

The Bell System Technical Journal  1931-01: Vol 10 Iss 1