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Limit of recursive sequence a n+1=a n 1−{a n}a n+1=a n 1−{a n}
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Consider the following sequence: let a 0>0 a 0>0 be rational. Define
a n+1=a n 1−{a n},a n+1=a n 1−{a n},
where {a n}{a n} is the fractional part of a n a n (i.e. {a n}=a n−⌊a n⌋{a n}=a n−⌊a n⌋). Show that a n a n converges, and find its limit.
We can show it converges as follows: suppose a n=p n/q n=k n+r n/q n a n=p n/q n=k n+r n/q n, where p n=k n q n+r n p n=k n q n+r n, 0≤r n<q n 0≤r n<q n. Then
a n+1=p n/q n 1−r n/q n=p n q n−r n,a n+1=p n/q n 1−r n/q n=p n q n−r n,
so the denominator will keep decreasing until it is a divisor of p 0 p 0 (maybe 1). Also, note we may take p n=p 0 p n=p 0 for all n n.
Further, the limit will be ≤p 0 gcf(p 0,q 0)≤p 0 gcf(p 0,q 0), because if f∣p 0 f∣p 0 and f∣q n f∣q n, then f∣(p 0−k n q n)=r n f∣(p 0−k n q n)=r n, so f∣q n−r n=q n+1 f∣q n−r n=q n+1. But the limit may be strictly smaller; for instance, a 0=30/7 a 0=30/7 converges right away to 6.
Can we say anything else about the limit of a sequence starting with a 0 a 0? This was a problem on a qualifier, so I suspect there is more to the answer, but maybe not.
calculus
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edited Aug 22, 2013 at 17:59
Eric AuldEric Auld
asked Jul 23, 2013 at 1:21
Eric AuldEric Auld
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1 Let f(p,q)f(p,q) be the limit of the sequence when a 1=p/q a 1=p/q; then I believe that for fixed p p, the function g p(q)=f(p,q)g p(q)=f(p,q) is periodic with period p p, while for fixed q q, the function h q(p)=f(p,q)h q(p)=f(p,q) is periodic with period lcm[1,2,…,q][1,2,…,q].Greg Martin –Greg Martin 2013-07-23 08:25:43 +00:00 Commented Jul 23, 2013 at 8:25
Greg's belief is correct: for fixed p p, if q=p k+r q=p k+r with k≥0 k≥0 and 0<r<p 0<r<p then the a n a n sequence begins p/q,p/(q−p),p/(q−2 p),...,p/(q−k p)=p/r p/q,p/(q−p),p/(q−2 p),...,p/(q−k p)=p/r, so indeed the limit is determined by p p and r r. If q q is fixed, and p p and P P are congruent to one another mod lcm(1,2,...,q)(1,2,...,q), then p p and P P are congruent to one another modulo any integer between 1 1 and q q, so they're congruent mod q n q n for all n n. Thus the sequences of q n q n's and r n r n's for a 1=p/q a 1=p/q are identical to the corresponding sequences for a 1=P/q a 1=P/q.Michael Zieve –Michael Zieve 2013-08-20 18:55:41 +00:00 Commented Aug 20, 2013 at 18:55
Greg's functions g p g p seems to be surjective as a function on {1,…,p}{1,…,p} into the divisors of p p.Uwe Stroinski –Uwe Stroinski 2013-08-22 11:55:37 +00:00 Commented Aug 22, 2013 at 11:55
So where is the question here?Norbert –Norbert 2013-08-22 16:33:59 +00:00 Commented Aug 22, 2013 at 16:33
1 @Norbert. The second sentence from the end looks like a question to me (though a bit open-ended, I'll admit).Rick Decker –Rick Decker 2013-08-22 16:51:41 +00:00 Commented Aug 22, 2013 at 16:51
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Wanted to record some observations here...
If we consider the sequence
p=a 1 q+r 1 p=a 1 q+r 1
p=a 2(q−r 1)+r 2 p=a 2(q−r 1)+r 2
p=a 3(q−r 1−r 2)+r 3 p=a 3(q−r 1−r 2)+r 3
etc.
and try to solve for the remainders in the form r i=c i p−d i q r i=c i p−d i q, there is a nice recursive relation:
First, c 1=1 c 1=1 and d 1=a 1 d 1=a 1, and in general,
c j=1+a j(c 1+…+c j−1)c j=1+a j(c 1+…+c j−1) and d j=(1+a 1)(1+a 2)⋯(1+a j−1)a j d j=(1+a 1)(1+a 2)⋯(1+a j−1)a j
We can also write c j(1+a j)=c 1+…+c j c j(1+a j)=c 1+…+c j so that c j=1+a j a j−1(c j−1(1+a j−1)−1)c j=1+a j a j−1(c j−1(1+a j−1)−1). This can be expanded further to obtain the form
c j=1+a j[1+(1+a j−1)[1+(1+a j−2)[1+…[1+(1+a 2)]]…]c j=1+a j[1+(1+a j−1)[1+(1+a j−2)[1+…[1+(1+a 2)]]…], or even
c j=1+a j+a j(1+a j−1)+a j(1+a j−1)(1+a j−2)+…+a j(1+a j−1)(1+a j−2)⋯(1+a 2)c j=1+a j+a j(1+a j−1)+a j(1+a j−1)(1+a j−2)+…+a j(1+a j−1)(1+a j−2)⋯(1+a 2)
Note that the a i a i are strictly increasing in the sequence. Suppose the procedure terminates at the n n-th step (when r n=0 r n=0). The limit is then p/a n p/a n, and 0=r n=c n(p/q)−d n 0=r n=c n(p/q)−d n or that p/q=d n/c n p/q=d n/c n.
I still don't have a closed form expression, but for instance:
For rationals p/q p/q for which the sequence terminates in the first step, 0=r 1=c 1(p/q)−d 1=p/q−a 1 0=r 1=c 1(p/q)−d 1=p/q−a 1, so that q q is a divisor of p p, and the limit is p/q p/q.
For termination at the second step, 0=r 2=c 2(p/q)−d 2=(1+a 2)(p/q)−(1+a 1)a 2 0=r 2=c 2(p/q)−d 2=(1+a 2)(p/q)−(1+a 1)a 2, or that p q=(1+a 1)a 2 1+a 2 p q=(1+a 1)a 2 1+a 2. If there exists a 1<a 2 a 1<a 2 satisfying this equality, then the sequence terminates in the second step. One such criteria is if d d is a divisor of p p, q=d+1 q=d+1 and p/d−1<d p/d−1<d, then the sequence terminates in the second step to p/d=(1+a 1)p/d=(1+a 1).
For examples: 30/7=(1+4)6/(1+6)30/7=(1+4)6/(1+6). Also, 30/4=120/16=(1+7)15/(1+15)30/4=120/16=(1+7)15/(1+15).
Third step termination: p q=(1+a 1)(1+a 2)a 3 1+a 3(1+(1+a 2))p q=(1+a 1)(1+a 2)a 3 1+a 3(1+(1+a 2)), limit is (1+a 1)(1+a 2)(1+a 1)(1+a 2), and etc.
Also, it appears that if you plug in a n=d a n=d in any formula, you can generate for which p p the process converges to d d by substituting any a 1<a 2<…<a n−1<d a 1<a 2<…<a n−1<d. Maybe there's more special structure...
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edited Aug 26, 2013 at 19:02
answered Aug 24, 2013 at 22:19
EvanEvan
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Let n∈N n∈N, q i q i for 0≤i≤n 0≤i≤n be a non-decreasing sequence of positive integers and
a 0=(∑j=0 n 1 q 0⋯q j)−1 a i+1=a i 1−{a i}a 0=(∑j=0 n 1 q 0⋯q j)−1 a i+1=a i 1−{a i}
Then a i=q n a i=q n for every i≥n i≥n.
This follows by proving, by induction, that for every i≤n i≤n we have
a i=(∑j=i n 1 q i⋯q j)−1(1)(1)a i=(∑j=i n 1 q i⋯q j)−1
This is clearly true for i=0 i=0. Assuming (1)(1), then clearly a i≤q i a i≤q i. On the other hand, j≥i j≥i implies q j≥q i>1 q j≥q i>1 (the case q i=1 q i=1 is trivial), hence
1 a i≤∑j=i n 1 q j+1 i<1 q i−1 1 a i≤∑j=i n 1 q i j+1<1 q i−1
from which q i−1<a i≤q i q i−1<a i≤q i that's ⌈a i⌉=q i⌈a i⌉=q i. Consequently,
a i+1=a i 1−{a i}=a i q i−a i=(∑j=i+1 n 1 q i+1⋯q j)−1 a i+1=a i 1−{a i}=a i q i−a i=(∑j=i+1 n 1 q i+1⋯q j)−1
hence the assertion follows by induction on i i.
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edited Oct 31, 2019 at 21:57
answered Oct 31, 2019 at 11:49
Fabio LucchiniFabio Lucchini
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Your proof is correct if a0 is rational number. It will not converge for irrational numbers like e or pi. Assuming rationality, I don't think there is a closed form solution of converging number (until and unless you embed loop and conditional kind of behavior in closed form). There is equivalence class of converging point given q0 where solution will converge: the equivalence class are factors (I mean all factors not only prime factors) of p0 including itself. Therefore, if p0 is prime it will converge to itself.
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answered Aug 22, 2013 at 16:49
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It is stated at the beginning of the question that a 1 a 1 is rational (I guess this was changed to a 0 a 0 later).Jonas Meyer –Jonas Meyer 2013-08-22 17:44:45 +00:00 Commented Aug 22, 2013 at 17:44
@JonasMeyer Thanks for pointing out the mistake in the question...changed the a 1 a 1 to a 0 a 0.Eric Auld –Eric Auld 2013-08-22 18:00:17 +00:00 Commented Aug 22, 2013 at 18:00
how to show this is not convergent for irrational starting point? i have shown this seq is increasing so it must not be bounded above.CHOUDHARY bhim sen –CHOUDHARY bhim sen 2019-02-25 12:54:40 +00:00 Commented Feb 25, 2019 at 12:54
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2601 | https://pubmed.ncbi.nlm.nih.gov/7925671/ | Parvovirus B19 outbreak on an adult ward - PubMed
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. 1994 Oct;113(2):345-53.
doi: 10.1017/s0950268800051773.
Parvovirus B19 outbreak on an adult ward
C Seng1,P Watkins,D Morse,S P Barrett,M Zambon,N Andrews,M Atkins,S Hall,Y K Lau,B J Cohen
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1 PHLS Communicable Disease Surveillance Centre, London.
PMID: 7925671
PMCID: PMC2271527
DOI: 10.1017/s0950268800051773
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Parvovirus B19 outbreak on an adult ward
C Seng et al. Epidemiol Infect.1994 Oct.
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. 1994 Oct;113(2):345-53.
doi: 10.1017/s0950268800051773.
Authors
C Seng1,P Watkins,D Morse,S P Barrett,M Zambon,N Andrews,M Atkins,S Hall,Y K Lau,B J Cohen
Affiliation
1 PHLS Communicable Disease Surveillance Centre, London.
PMID: 7925671
PMCID: PMC2271527
DOI: 10.1017/s0950268800051773
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In November and December 1992, an outbreak of parvovirus B19 infection occurred among patients and staff on an adult mixed surgical ward at a large hospital in London. Three patients and 15 staff members were serologically confirmed as acute cases. The attack rate among susceptible members of staff was 47%. In those infected, arthralgia (80%) and rash (67%) were the most common symptoms. Of six susceptible in-patients on the ward, three became infected. One of the in-patients who had carcinoma of the mouth was viraemic for more than 10 days with marrow suppression resulting in the postponement of chemotherapy until intravenous immunoglobulin was given and he was no longer viraemic. Control measures taken included closure of the ward to new admissions, transfer of only immune staff to the ward, and restriction of the ward nursing staff to working only on that ward. Although no specific exposure was conclusively identified as a risk factor, there was a suggestion of an increased risk of acquiring parvovirus B19 infection among those staff who did not adopt strict hand washing procedures after each physical contact with a patient (RR = 2.33; P = 0.07). Knowledge of parvovirus B19 among interviewed health care workers was poor: only 42% reported knowing about parvovirus B19 and only 38% could name a patient category at risk of a severe outcome following infection. This is the first report of a nosocomial outbreak affecting an adult ward and of possible transmission of parvovirus B19 infection from staff to in-patients. Hospital control of infection teams should include parvovirus B19 in their outbreak containment plans.
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Structures and implications of the nuclease domain of human parvovirus B19 NS1 protein.Zhang Y, Shao Z, Gao Y, Fan B, Yang J, Chen X, Zhao X, Shao Q, Zhang W, Cao C, Liu H, Gan J.Zhang Y, et al.Comput Struct Biotechnol J. 2022 Aug 27;20:4645-4655. doi: 10.1016/j.csbj.2022.08.047. eCollection 2022.Comput Struct Biotechnol J. 2022.PMID: 36090819 Free PMC article.
First report on severe septic shock associated with human Parvovirus B19 infection after cardiac surgery.Xiang C, Wu X, Wei Y, Li T, Tang X, Wang Y, Zhang X, Huang X, Wang Y.Xiang C, et al.Front Cell Infect Microbiol. 2023 Apr 5;13:1064760. doi: 10.3389/fcimb.2023.1064760. eCollection 2023.Front Cell Infect Microbiol. 2023.PMID: 37091672 Free PMC article.
Use of ward closure to control outbreaks among hospitalized patients in acute care settings: a systematic review.Wong H, Eso K, Ip A, Jones J, Kwon Y, Powelson S, de Grood J, Geransar R, Santana M, Joffe AM, Taylor G, Missaghi B, Pearce C, Ghali WA, Conly J.Wong H, et al.Syst Rev. 2015 Nov 7;4:152. doi: 10.1186/s13643-015-0131-2.Syst Rev. 2015.PMID: 26546048 Free PMC article.
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2602 | https://www.convertunits.com/from/cubic+angstrom/to/meters%5E3 | Convert cubic angstrom to meters^3 - Conversion of Measurement Units
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2603 | https://math.stackexchange.com/questions/580545/suppose-a-nb-n-converges-does-a-nb-n-converges-also | calculus - Suppose $a_n+b_n$ converges. Does $a_nb_n$ converges also? - Mathematics Stack Exchange
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Suppose a n+b n a n+b n converges. Does a n∗b n a n∗b n converges also?
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a n,b n a n,b n - sequences
Suppose a n+b n a n+b n converges. Does a n b n a n b n converge also?
I tried thinking if I can learn something about a n a n and b n b n by the assumption a n b n a n b n converges.
I also tried to develop this equation |a n b n−L|<ϵ|a n b n−L|<ϵ assuming it is converging.
I didn't get any bright conclusions.
Will be glad help.
calculus
limits
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edited Nov 25, 2013 at 19:24
lily
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asked Nov 25, 2013 at 13:27
captain dragoncaptain dragon
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What does the first "sentence" mean? Certainly those are sequences, not sets.Marc van Leeuwen –Marc van Leeuwen 2013-11-25 13:51:41 +00:00 Commented Nov 25, 2013 at 13:51
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a n=n,b n=−n a n=n,b n=−n
a n+b n=0 a n+b n=0 but then...
a n b n=−n 2→−∞a n b n=−n 2→−∞
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edited Nov 25, 2013 at 13:39
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answered Nov 25, 2013 at 13:32
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1 When we are talking about convergence, I think it needs to converges to a real number. A sequences that tends to infinity does not converges Giiovanna –Giiovanna 2013-11-25 13:34:44 +00:00 Commented Nov 25, 2013 at 13:34
1 Yes, indeed. This only shows an example when the product does not converges. But it is really easy to show one that the product does converges. Just take 2 constant sequences. The sum is a constant and so does the product. Then, as we can see, the product can or not converge Giiovanna –Giiovanna 2013-11-25 13:39:13 +00:00 Commented Nov 25, 2013 at 13:39
3 Could you please let me know what are you expecting when you say "general way"user87543 –user87543 2013-11-25 14:04:31 +00:00 Commented Nov 25, 2013 at 14:04
3 @Giiovanna: The question is if a n+b n a n+b n, does a n b n a n b n also converge? Certainly, anyone can come up with cases where both converge. However, to negate a false implication, one needs to satisfy the hypotheses (in this case, a n+b n a n+b n converges), yet show that the conclusion is not necessarily true (in this case, a n b n a n b n does not converge). Finding a pair of sequences for which a n b n a n b n converges does not apply to the question.robjohn –robjohn♦ 2013-11-25 15:34:23 +00:00 Commented Nov 25, 2013 at 15:34
1 @Giiovanna If you were asked to prove that 2x is not equal to x^2 how would you do that? Then what would you say to the person that says well it's true for x=2!Cruncher –Cruncher 2013-11-25 18:01:03 +00:00 Commented Nov 25, 2013 at 18:01
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As shown in Praphulla Koushik's answer
a n=n,b n=−n a n=n,b n=−n
cancellation is a problem.
However, even if you restrict a n,b n≥0 a n,b n≥0 then the answer is no. For example, if
a n=2+(−1)n,b n=3−(−1)n a n=2+(−1)n,b n=3−(−1)n
then a n+b n=5 a n+b n=5 yet a n b n a n b n oscillates between 4 4 and 6 6.
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answered Nov 25, 2013 at 13:47
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2Limit of a sequence: a n≤b n≤c n a n≤b n≤c n
1Prove that 0≤a n,b n≤1 0≤a n,b n≤1 converge to 1 1 given lim n→∞a n b n=1 lim n→∞a n b n=1
1∑a n∑a n converges but a n=b n−c n a n=b n−c n for suitable (b n),(c n)(b n),(c n) is impossible
0lim n→∞a n b n=∞lim n→∞a n b n=∞, ∑a n∑a n converges, does ∑b n∑b n converge?
3(a n)∞n=1(a n)n=1∞&(b n)∞n=1(b n)n=1∞ are seq st (a n)∞n=1(a n)n=1∞&[(a n)∞n=1+(b n)∞n=1][(a n)n=1∞+(b n)n=1∞] con. Prove (b n)(b n) con
7Does there exists a sequence b n b n, s.t. lim b n=0 lim b n=0 and for every divergent series ∑a n∑a n, The series ∑a n b n∑a n b n also diverges?
0Suppose a n b n a n b n converges to a limit S. Show that ∑N n=0 a n∑N n=0 b n∑n=0 N a n∑n=0 N b n converges to that same limit S
2If two sequences {a n}{a n} and {b n}{b n} be such that a n>b n a n>b n then show that lim a n≥lim b n lim a n≥lim b n
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2604 | https://cs.brown.edu/courses/csci1951-w/lec/lec%203%20notes.pdf | CSCI 1951-W Sublinear Algorithms for Big Data Fall 2020 Lecture 3: Concentration Inequalities and Mean Estimation Lecturer: Jasper Lee Scribe: Ross Briden 1 Overview In this lecture, we’ll quickly recap how an optimization algorithm that fails with constant probability can be adapted into a high probability one. Then, we’ll introduce two concentra-tion inequalities – Hoeffding’s inequality and Bernstein’s inequality – for analyzing the sample complexity of the sample mean for estimating the mean of an underlying distri-bution. We also state (without proof) a sample complexity lower bound that demonstrates the sample mean of independent random variables, in non-asymptotic regimes, is a poor estimator for their true mean, and introduce a more robust estimator known as the Median of Means algorithm.
2 Recap from Last Class Suppose we have an optimization problem specified by an objective function f : S →R subject to a set of constraints {C1, . . . , Ck}, where S is any set and Ci ⊆S for all constraints.
Definition 3.1 We say that y ∈S is a feasible solution to our optimization problem if y ∈Tk i=1 Ci.
The objective of the optimization problem is to find a feasible solution x that either maxi-mize or minimize f(x). WLOG suppose we want to maximize f.
Now, suppose there exists an algorithm D that, with probability ≥2 3, returns an answer z so that f(z) ≥OPT−ϵ with a promise that z ∈Tk i=1 Ci, where OPT is the global maximum of f. Notice that the promise ensures that the algorithm returns a value that is feasible if it succeeds.
Now, we can design a new algorithm D′ that uses D as a subroutine to solve the opti-mization problem with high probability. In particular, given δ ∈(0, 1), D′ should run D for n = Θ(log( 1 δ)) iterations and return the largest feasible value returned by D. To see why D′ is correct, first notice that the promise ensures that every time D succeeds, the solution returned by D is feasible. Therefore, the maximum of feasible solutions is also feasible and will also be greater than OPT−ϵ by definition. Hence, D′ returns a valid solution if after n runs of D, at least one valid solution is returned by D. It then follows that the probability D′ fails is (1 −2 3)n = 1 3n . Therefore, 1 3n ≤δ ⇐ ⇒−n log(3) ≤log(δ) ⇐ ⇒n ≥log 1 δ / log(3) So, if D′ runs D for Θ( 1 δ) iterations, D′ will return a correct solution with probability at least 1 −δ, as desired.
1 This example, along with the examples we saw last lecture, demonstrates that many con-stant probability algorithms can be converted to high probability algorithms relatively eas-ily. So, before designing a high probability algorithm, you should always consider whether a constant probability algorithm would be sufficient for your problem. Moreover, we can generalize to the following this heuristic: Heuristic 3.2 When designing high probability algorithms, the sample / query complexity q(δ) of your algorithm should be at least as good as q(δ) = O(q( 1 3) log( 1 δ)). Otherwise, a constant probability algorithm combined with a boosting technique (e.g. finding the median or max of solutions returned by a constant probability algorithm) will improve the δ dependence to the multiplicative log( 1 δ) factor. Moreover, note that the log( 1 δ) term is not tight for all problems; we can sometimes do better than this!
3 Concentration Inequalities for Mean Estimation First, we’ll derive two concentration inequalities – Hoeffding’s inequality and Bernstein’s inequality – and use them to evaluate the sample complexity of the sample mean when estimating the true mean of a collection of i.i.d. random variables.
Remark 3.3 For the theorems below, we’ll be assuming that all random variables are one dimensional.
Lemma 3.4 (Hoeffding’s Lemma) Let X be a real-valued random variable so that P(X ∈[a, b]) = 1 for some a < b and E[X] = 0. Then, E[etX] ≤e t2(b−a)2 8 .
Remark 3.5 The proof of Hoeffding’s lemma is complex and involves some calculus, so it was not be covered in class. Nevertheless, the intuition is to notice x →ext is convex and thus Jensen’s inequality can be applied.
Theorem 3.6 (Hoeffding’s Inequality) Let {Xi}n i=1 be independent random variables so that P(Xi ∈[a, b]) = 1 for some a < b, and let ϵ > 0. Then: 1. P(Xn −E[Xn] ≥ϵ) ≤e −2nϵ2 (b−a)2 2. P(Xn −E[Xn] ≤−ϵ) ≤e −2nϵ2 (b−a)2 3. P(|Xn −E[Xn]| ≥ϵ) ≤2e −2nϵ2 (b−a)2 Proof. We will first show (1) and use it to easily prove (2) and (3). Now, P(Xm −E[Xn] ≥ϵ) ≤inf t>0 E[et(Xn−E[Xn])] etϵ = inf t>0 E[e t n (Pn i=1 Xi−E[Xi])] etϵ = inf t>0 Qn i=1 E[e t n (Xi−E[Xi])] etϵ 2 by Lemma (2.8) and since X1, . . . , Xn are independent. By applying Hoeffding’s lemma to this inequality, we have that inf t>0 Qn i=1 E[e t n (Xi−E[Xi])] etϵ ≤inf t>0 Qn i=1 e (t2/n2)(b−a)2 8 etϵ = inf t>0 e (nt2/n2)(b−a)2 8 etϵ = inf t>0 e (t2/n)(b−a)2 8 −tϵ Recall that ex is an increasing function, so the infimum of e (t2/n)(b−a)2 8 −tϵ will be determined by the minimum value of (t2/n)(b−a)2 8 −tϵ. Furthermore, this is a quadratic function, which implies that the minimum of (t2/n)(b−a)2 8 −tϵ will be −y 2x where x = (b−a)2 8n and y = ϵ. There-fore, the minimum occurs at t = 4nϵ (b−a)2 . Moreover, since t > 0, we can bound the infimum by t = 4nϵ (b−a)2 .
Now, it follows that inf t>0 e (t2/n)(b−a)2 8 −tϵ = e −2nϵ2 (b−a)2 Hence, inequality (1) follows. By negating Xn and translating, we get inequality (2).
Finally, by combining inequalities (1) and (2), we have that P(|Xn −E[Xn]| ≥ϵ) = P(Xn −E[Xn] ≥ϵ) + P(Xn −E[Xn] ≤−ϵ) ≤2e −2nϵ2 (b−a)2 which proves inequality (3).
Corollary 3.7 Suppose that {Xi}n i=1 are i.i.d. random variables so that P(Xi ∈[a, b]) = 1, with a < b, holds for all Xi. Then, given ϵ > 0, the sample complexity of the sample mean required to estimate E[X] to an additive error ϵ while only failing with probability at most δ is n = O( σ2 ϵ2 + M ϵ ) log( 1 δ).
Proof. Let ϵ > 0. Now, the probability of failure for this problem is P(|Xn −E[Xn]| ≥ϵ).
Furthermore, by Hoeffding’s inequality, we have that P(|Xn −E[Xn]| ≥ϵ) ≤2e −2nϵ2 (b−a)2 .
Hence, it follows that 2e −2nϵ2 (b−a)2 ≤δ ⇐ ⇒n ≥(b −a)2 2ϵ2 log 2 δ Thus, we should select n = O (b−a)2 ϵ2 log 1 δ to estimate the mean of {Xi}n i=1 to an additive error ϵ.
Is this a good sample complexity? In other words, can we exploit any more information about {Xi}n i=1 to reduce the sample complexity of this problem? Yes, in many cases we can further reduce the sample complexity. To see why this is intuitively true, suppose you have a collection of i.i.d. random variables X1, . . . , Xn that are drawn from a distribution that has high probability mass in the interval [0, 1] and zero mass outside of this interval, 3 Figure 1: How outliers can hurt sample complexity when using Hoeffding’s inequality. In this case, the interval [a, b] must be large enough to contain the points located in the green probability mass, making [a, b] quite large and thus increasing the sample complexity. In this case, M represents the distance from the mean of the distribution to the center of the outlier probability mass.
with the exception of one small interval [α, β] with non-zero probability mass and 1 ≪α.
Then, since P(Xi ∈[α, β]) > 0 for all Xi, b is forced to be greater than or equal to β (See Figure 1 for a visual example of this). However, if the mass in [α, β] is sufficiently small, then n samples will not see anything outside [0, 1], making Hoeffding’s inequality very loose.
The variance is a much more robust way to measure the ”width” of a distribution than the interval covering all its probability mass.
Hence, the robustness of variance motivates our next inequality: Bernstein’s Inequal-ity, which uses variance information to achieve a tighter bound.
Theorem 3.8 (Bernstein’s Inequality) Let {Xi}n i=1 be independent random variables so that P(|Xi−E[Xi]| ≤M) = 1 holds for all Xi for some M ≥0. Also, let σ2 = 1 n Pn i=1 Var[Xi].
Then, P(Xn −E[Xn]) ≤exp −nϵ2 2σ2 + 2Mϵ 3 Remark 3.9 The proof of this is quite complicated and was not covered in lecture.
Corollary 3.10 Suppose that {Xi}n i=1 are i.i.d. random variables so that P(|Xi −E[Xi]| ≤ M) = 1 holds for all Xi for some M ≥0. Then, given ϵ > 0, the sample complexity of the sample complexity required to estimate E[X] to an additive error ϵ while only failing with probability at most δ is n = O( σ2 ϵ2 + M ϵ ) log( 1 δ).
Proof. Fix ϵ > 0. Then, the probability of failure is P(Xn −E[Xn] ≥ϵ). By Bernstein’s inequality, we have that P(Xn −E[Xn] ≥ϵ) ≤exp −nϵ2 2σ2+ 2Mϵ 3 . Therefore, it follows that exp −nϵ2 2σ2 + 2Mϵ 3 ≤δ ⇐ ⇒ nϵ2 2σ2 + 2Mϵ 3 ≥log 1 δ ⇐ ⇒n ≥ 2σ2 ϵ2 + 2 3 M ϵ log 1 δ Therefore, n = O ( σ2 ϵ2 + M ϵ ) log( 1 δ) , as desired.
4 So, making essentially the same assumption that the random variables are bounded in an interval of width O(M) = O(b −a), we have two possible sample complexities for the sample mean to choose from: n = O (b−a)2 ϵ2 log 1 δ (derived from Hoeffding) and n = O ( σ2 ϵ2 + M ϵ ) log( 1 δ) (derived from Bernstein) when estimating the mean of independent random variables. It’s therefore natural to ask if and when the second sample complexity is better than the first.
The fundamental insight is that when the standard deviation σ of the random variables is significantly smaller than M, in other words σ ≪M, the sample complexity provided by Bernstein’s inequality is tighter since the bound provided by Hoeffding’s inequality grows quadratically with M. This is intuitive because factoring in information about variance will allow us to better approximate the underlying behavior of the random variables, requiring fewer samples to approximate their mean. In the regime where σ ≈M, the two sample complexities are asymptotically the same, so Bernstein’s inequality will provide minimal benefit over Hoeffding’s inequality. Nevertheless, Bernstein’s inequality provides a sample complexity at least as good as that given by Hoeffding’s inequality because σ2 ≤O(M2) always holds.
4 Comparison with CLT; Median of Means Method In statistics, the sample mean of a collection of independent random variables is often used to estimate their true mean.
Moreover, the central limit theorem states that the sample mean Xn of a collection of i.i.d. random variables X1, . . . , Xn exhibits Gaussian-like behavior 1 as n →∞. Yet, as we’ll see, in the non-asymptotic regime, when estimating E[Xn] with high probability, Xn does not behave necessarily like a Gaussian – no matter what concentration inequality you use to bound the sample mean. The upside, however, is that in the constant probability regime (i.e. when the probability of error is fixed), Xn does provide Gaussian-like guarantees on sample complexity! Using this, we will introduce the Median of Means algorithm which estimates E[Xn] with high probability while also giving Gaussian-like performance.
To start our analysis, recall that given i.i.d. Gaussian random variables X1, . . . , Xn ∼ N(µ, σ2), we have that P(|Xn −E[Xn]| ≥ϵ) ≈exp −nϵ2 2σ2 Therefore, exp −nϵ2 2σ2 ≤δ ⇐ ⇒n ≥2σ2 ϵ2 log 1 δ So, the sample complexity of the sample mean is n ≈2σ2 ϵ2 log 1 δ when X1, . . . , Xn are Gaussian. Given this, we can show that the sample complexity derived from Bernstein’s inequality is not Gaussian in all regimes: Proposition 3.11 n = O(( σ2 ϵ2 + M ϵ ) log( 1 δ)) is worse than n = O( σ2 ϵ2 log 1 δ ) and thus the sample complexity of the sample mean given by Bernstein’s inequality does not give Gaussian-behavior in all regimes.
Proof. There are three distinct cases: 1A similar statement holds for independent but not identical random variables, requiring additional constraints on the moments of these random variables. See Lyapunov’s CLT for more information.
5 1. Case 1: ϵ ≫σ2 M . Since σ2 ϵ2 has an inverse square dependence on ϵ, we have that M ϵ ≫σ2 ϵ2 . Therefore, O ( σ2 ϵ2 + M ϵ ) log( 1 δ) provides a worse sample complexity than O σ2 ϵ2 log 1 δ in this case.
Note that this is the case that concretely shows that Bernstein’s inequality does not give a Gaussian-like sample complexity for the sample mean.
2. Case 2: ϵ = Θ( σ2 M ).
Then, M = Θ( σ2 ϵ ).
So then follows that M ≈ σ2 ϵ .
So by substituting this value of M into O σ2 ϵ2 + M ϵ , we have that O σ2 ϵ2 + M ϵ ≈O σ2 ϵ2 Therefore, Bernstein’s inequality gives roughly a Gaussian sample complexity for the sample mean in this case.
3. Case 3: ϵ ≪σ2 M . Since σ2 ϵ2 has an inverse square dependence on ϵ, we have that M ϵ ≪σ2 ϵ2 . Therefore, since σ2 ϵ2 dominates M ϵ , Bernstein’s inequality gives the same sample complexity for the sample mean as the Gaussian case.
Since Bernstein’s inequality does not give Gaussian-like performance in some regimes, it’s pertinent to consider whether any concentration inequality can guarantee a Gaussian-like sample complexity for the sample mean.
Catoni 2012 provides a lower bound on the sample complexity for the sample mean, which answers this question negatively: Theorem 3.12 (Catoni 2012)2 Given a collection X1, . . . , Xn of independent random variables, the sample mean Xn needs Ω( σ2 ϵ2δ) samples, assuming that the second moment of Xn is finite.
Given this result, we know that there exists a bad distribution D so that the sample mean requires at least Ω( σ2 ϵ2δ) samples to estimate the mean of D to additive error ϵ, with proba-bility at least 1 −δ. Hence, in the high probability regime (when δ is not fixed), the sample complexity of the sample mean varies inverse linearly in δ, which blows up when δ is made small. Conversely, the Gaussian sample complexity for the sample mean grows at a much slower rate of log( 1 δ). Therefore, the sample mean is a poor estimator of the true mean of a collection of independent random variables because it does not have Gaussian-like perfor-mance, as claimed by the CLT, for some distributions.
However, by examining the lower bound Ω( σ2 ϵ2δ) closely, you’ll notice that if we fix δ as a constant, the lower bound has the same form as the sample complexity of the Gaussian case! Namely, the Gaussian sample complexity for the sample mean 2 σ2 ϵ2 log( 1 δ) differs from σ2 ϵ2δ only by a constant factor when δ is a fixed constant. So, in the constant probability regime, the sample mean could be a Gaussian-like estimator. In fact, we can easily demon-strate that the sample mean has Gaussian sample complexity in this regime. If we assume {X1, . . . , Xn} is a collection of i.i.d. random variables with variance σ2, then P(|Xn −E[Xn]| ≥ϵ) ≤σ2 ϵ2n 2 6 Figure 2: Concentration of the Sample Mean by Chebyshev’s inequality. Given this bound on Xn, it follows that the sample complexity is σ2 ϵ2n ≤δ ⇐ ⇒n ≥σ2 ϵ2δ So, if we fix δ as a constant, then the sample complexity of the sample mean becomes n = O( σ2 ϵ2 ), which is Gaussian. Therefore, in the constant probability regime, the sample mean is a good estimator of the true mean of X1, . . . , Xn.
The above observations can alternatively be summarized by Figure 2, which shows a “bump” denoting the distribution of the sample mean normalized to variance 1, namely √n σ Xn. We can divide the bump into two regimes: the “body” which is within a constant number of standard deviations from the mean, and the “tail” which is the rest of the distribution. Thus, by Chebyshev’s inequality, the body is always Gaussian-like, in a big-O sense. The tail, on the other hand, can be heavy (having a lot of mass, decaying very slowly) and badly-behaved, from Catoni’s lower bound.
Moreover, since the sample mean provides a good constant probability estimate for the true mean, we can combine it with the median trick from last lecture to design a high probability algorithm known as Median of Means: Algorithm 1: Median of Means input : X1, . . . , Xn samples where n = O( σ2 ϵ2 log( 1 δ)) output: A mean estimate µ so that |µ −E[Xn]| ≥ϵ with probability less than δ steps: 1. Divide samples in to m = Θ(log(1 δ)) groups 2. Compute sample mean Si for each group i, where each group is of size O( σ2 ϵ2 ) 3. Output median of {Si}m i=1 In the median of means algorithm, the sample mean of each group Si is within ϵ of the true mean with probability ≥2 3. This is done by selecting a group size of 3σ2 ϵ2 = O( σ2 ϵ2 ) in Step 2 of the algorithm. Thus, using the median technique from last lecture, the success probability of the entire algorithm is at least 1 −δ.
7 Remark 3.13 In conclusion, the median of means algorithm should be preferred over the sample mean for estimating the mean of a collection of independent random variables with high probability, as it provides Gaussian-like performance that is unattainable with the sample mean in general.
5 Conclusions • Constant probability algorithms are incredibly useful and can be used to build efficient high probability algorithms; moreover, when designing high probability algorithms, a constant probability counterpart should serve as a baseline for performance.
• While concentration inequalities are very useful, they are only an analysis technique and shouldn’t be blindly applied.
• Don’t forget about Chebyshev’s inequality, which can be powerful (and tight) when used correctly with other tools.
6 Additional Content Theorem 3.14 (McDiarmid’s Inequality) Let {Xi}n i=1 be independent random vari-ables, where each Xi : Ω− →Si for some set Si. Consider f : Qn i=1 Si − →R, and suppose that f is Ci-lipschitz in the i-th coordinate of the input for all i. Then, for any ϵ > 0, P(f(x1, . . . , xn) −E[f] ≥ϵ) ≤exp −2ϵ2 Pn i=1 C2 i Remark 3.15 By Ci-lipschitz, we mean that given a Ci ∈R≥0 and any two points x = (x1, . . . , xi, . . . , xn), x′ = (x1, . . . , x′ i, . . . , xn) ∈Qn i=1 Si that differ by only their i-th corrdinate, we have that |f(x) −f(x′)| ≤Ci Remark 3.16 Notice the connection between McDiarmid’s inequality and Hoeffding’s in-equality. Simply set f(x1, . . . , xn) = 1 n Pn i=1 xi and you’ll get back Hoeffding’s inequality.
8 |
2605 | https://eceseniordesign2022spring.ece.gatech.edu/sd22p35/trpbalkew.pdf | Directivity and Gain of Antennas and Applications of Parabolic Dish Antennas in Satellite Tracking and Communication I. Introduction The purpose of this document is to provide a basic technical review on the current state of antenna technology centered around satellite tracking applications. In particular, the directivity and gain of parabolic dish antennas are emphasized and analyzed. In the context of ground-based satellite tracking, these two parameters are critical in being able to predict various design requirements and specifications for a given tracking system, including but not limited to: input signal variability tolerances, tracking model error tolerance, system response time, and others. II. Fundamentals of Antenna Directivity and Gain The directivity of an antenna can be defined as its ability to radiate in a particular direction relative to an isotropic source, which is a theoretical antenna that emits electromagnetic radiation with the same intensity in all directions . The latter does not exist in practice but is widely used as the standard measuring device for quantifying the directivity of practical antennas. Antenna gain is directly related to directivity, in that the gain of an antenna is equal to its directivity up to a constant. This constant is the product of the dielectric and conduction losses associated with a given antenna. Furthermore, it is important to note that a physical antenna is incapable of converting 100% of the energy it receives from a power source into “information” (in the form of radiation) that can be accurately interpreted by a receiver. The quantity used to describe power losses of this kind is known as the reflection efficiency, which is used to compute what is known as realized gain . This in unison with the fact that gain is a function of received electromagnetic power (unlike directivity from a single source, which emphasizes the directionality of transmitted radiation) demonstrates that the gain parameter takes into account realistic non-idealities associated with satellite tracking and communication. III. Industry Usage and Applications to Project A variety of antennas are currently used in industry for satellite tracking. This document shall focus on a specific type known as parabolic dish antennas, since the primary system of interest later on will be a parabolic antenna. Parabolic antennas can be found in numerous aspects of day-to-day life, such as TV satellite dishes, cellphone communications, cellular data, and so on. At the precipice of these applications resides a fundamental minimum directivity and gain associated with these parabolic dish antennas. According to , high-performance parabolic antennas maintain a high efficiency due to the geometry of the device. In particular, the component of the antenna responsible for receiving and “guiding” the data to the desired destination (otherwise known as the reflector) is able to operate at higher efficiencies due to the symmetric parabolic profile of the antenna . Recent work done by Nagasaka, et al. have demonstrated gain values (aperture efficiency) as high as 34.0 dB at 12 GHz and 38 dB at 21 GHz. Also, work done by Jablon et al. demonstrated antenna gains as high as 40.5 dB at approximately 13 GHz using radar altimeter antennas, devices that are stated to be well-approximated, performance-wise, by wide dish parabolic antennas . This along with constructive wave interference that occurs prior to transmission (EM waves become in phase) leads to very high directivity and high gain associated with these antennas. In addition, as outlined in , parabolic antennas exist in numerous variations. One type, known as a Cassegrain antenna, is particularly efficient as a transmitting device due to the unique hyperboloid concave reflector of the structure. Lastly, it is important to note that both the directivity and the gain of a given parabolic antenna increases as the size (usually look at diameter) of the parabolic profile increases. This is primarily due to similar effects of the antenna’s physical structure that was outlined earlier. Due to the geometry-based increases in directivity and gain as well as the drop in radiated beamwidth, parabolic antennas are ideally designed for very high frequency applications in which high-frequency gain losses are prevalent. It is fundamental for antennas utilized in satellite tracking to possess and maintain high directivities and gains in order for a designed auto-tracking system to produce models capable of meeting design specifications. For example, formulating a system that incorporates both RF design and Machine Learning algorithms to keep a grounded satellite dish locked onto a low-Earth orbit satellite would first require an antenna with a high directivity and gain. This would allow for less non-idealities to factor into the RF design process, thereby decreasing the required complexity and robustness (with respect to handling input data) of the implemented ML algorithm. IV. Collaboration with VIASAT This project is to be done in collaboration with an industry leader in antenna design, analysis, and synthesis – VIASAT. As such, the important antenna characteristics and parameters outlined in this review shall serve as an initial foundation for discussions with VIASAT moving forward. V. Key Takeaways To conclude, the numerous advantages of parabolic dish antennas in relation to satellite tracking and communication and the work collaboration with VIASAT place this research effort at the forefront of improving performance and efficiency in this area across the board. The unique characteristics and advantages of parabolic dish antennas provide a research-driven advantage within our area of interest. VI. References “Antenna gain,” Antenna Gain - an overview | ScienceDirect Topics. [Online]. Available: [Accessed: 08-Oct-2021]. E. Notes, “Parabolic reflector antenna: Dish antenna,” Electronics Notes. [Online]. Available: [Accessed: 08-Oct-2021]. M. Nagasaka, S. Nakazawa and S. Tanaka, "Study on 12/21-GHz Dual-circularly Polarized Receiving Antenna for Satellite Broadcasting," 2018 IEEE International Symposium on Antennas and Propagation & USNC/URSI National Radio Science Meeting, 2018, pp. 339-340, doi: 10.1109/APUSNCURSINRSM.2018.8609229 C. Balanis, Antenna theory: Analysis and design. Hoboken, NJ: Wiley-Interscience, 2016. Allan R. Jablon and Robert K. Stillwell, “Spacecraft Reflector Antenna Development: Challenges and Novel Solutions.” journal article Antenna theory - parabolic reflector. [Online]. Available: [Accessed: 08-Oct-2021]. |
2606 | https://ozdic.com/collocation/negotiate | negotiate - OZDIC - English collocation examples, usage and definition
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negotiateverb
1 try to reach an agreement
ADV. carefullya carefully negotiated series of concessions| successfully | freely | individually, separatelyRents are individually negotiated between landlord and tenant.| jointly | directlynegotiating directly with the rebels| secretly | constantly, continuallyThe parameters of the job are being continually negotiated.
VERB + NEGOTIATE be able to | be prepared to, be willing toAre the employers really willing to negotiate?| attempt to, seek to, try to | manage to | help (to)
PREP. betweento negotiate between the two sides| forWe are negotiating for the release of the prisoners.| onThey have refused to negotiate on this issue.| on behalf ofnegotiating on behalf of Britain| withI managed to negotiate successfully with the authorities.
2 successfully get over/past sth
ADV. easily, safelyHe safely negotiated the slippery stepping stones.
VERB + NEGOTIATE be difficult toThe flight of steps was quite difficult to negotiate with a heavy suitcase.
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2607 | https://www.scribd.com/document/467618286/Ladder-problems-m2 | Ladder Problems m2 | PDF | Ladder | Center Of Mass
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Ladder Problems m2
Ladder problems use moments and force resolution to analyze situations involving ladders resting against walls and on surfaces. Examples provided involve finding tensions in supporting st…
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| Bailey & Scott's Diagnostic Microbiology Patricia M. Tille Elsevier Health Sciences, Feb 4, 2021 - Medical - 1184 pages Textbook and Academic Authors Association (TAA) Textbook Excellence Award Winner, 2024Selected for Doody's Core Titles® 2024 in Laboratory TechnologyPerfect your lab skills with the essential text for diagnostic microbiology! Bailey & Scott's Diagnostic Microbiology, 15th Edition Is known as the #1 bench reference for practicing microbiologists and as the preeminent text for students in clinical laboratory science programs. With hundreds of full-color illustrations and step-by-step methods for procedures, this text provides a solid, basic understanding of diagnostic microbiology and also covers more advanced techniques such as matrix-assisted laser desorption time-of-flight mass spectrometry. Written by noted CLS educator Dr. Patricia Tille, Diagnostic Microbiology has everything you need to get accurate lab test results in class and in clinical practice. - More than 800 high-quality, full-color illustrations help you visualize concepts. - Expanded sections on parasitology, mycology, and virology allow you to use just one book, eliminating the need to purchase other microbiology textbooks for these topics. - Hands-on procedures show exactly what takes place in the lab, including step-by-step methods, photos, and expected results. - Case studies allow you to apply your knowledge to diagnostic scenarios and to develop critical thinking skills. - Genera and Species boxes provide handy, at-a-glance summaries at the beginning of each organism chapter. - Learning objectives at the beginning of each chapter provide measurable outcomes to achieve by completing the chapter material. - A glossary defines terms at the back of the book and on the Evolve companion website. - New! Updated content includes infectious disease trends and new illustrations such as culture plate images of real specimens, complex gram stains, lactophenol cotton blue microscopy, and more. - NEW COVID-19 information has been added. - UPDATED topics include the Human Microbiome Project, expanded MALDI-TOF applications and molecular diagnostics in conjunction with traditional microbiology, additional streps, and significant news in mycology. - EXPANDED glossary defines terms on the Evolve companion website. Preview this book » |
Selected pages
Page 63
Title Page
Table of Contents
Index
References
Contents
| | |
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| I Basic Medical Microbiology | 1 |
| |
| II General Principles in Clinical Microbiology | 44 |
| |
| III Bacteriology | 208 |
| |
| IV Parasitology | 601 |
| |
| V Mycology | 791 |
| |
| | |
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| VI Virology | 884 |
| |
| VII Diagnosis by Organ System | 953 |
| |
| VIII Clinical Laboratory Management | 1085 |
| |
| Index | 1126 |
| |
| Copyright |
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Other editions - View all
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| Bailey & Scott's Diagnostic MicrobiologyPatricia TilleNo preview available - 2021 |
Common terms and phrases
acid-fast addition aerobic amplification anaerobic antibiotic antibody antigen antimicrobial antimicrobial agents assays associated aureus bacilli bacteremia bacteria beta-lactam biochemical blood culture broth Brucella Campylobacter catalase cause cephalosporins Chapter characteristics chocolate agar Clin Microbiol clinical microbiology clinical specimens CLSI cocci coli colonies containing detection diagnosis differentiate disk endocarditis Enterobacterales enterococci enzyme factors fermentation fluid fluorescent gene genera genus glucose Gram stain gram-negative gram-positive grow growth Haemophilus host human hybridization identification immune incubation infections inhibit inoculated isolated laboratory Legionella MacConkey agar MALDI-TOF medium membrane methods microbial microorganisms molecular morphology Mycobacterium Mycoplasma negative Neisseria normal microbiota nucleic acid organisms pathogens patients plate pneumoniae positive probe procedures produce protein Pseudomonas reaction resistance respiratory rods sample sequence serum sheep blood agar skin species Spectrum of Disease Staphylococcus sterile Streptococcus subsp swab Table target therapy tion tissue toxin tract tube tuberculosis vancomycin virulence
About the author (2021)
Department Vice Chair, Clinical and Health Information Sciences, Graduate Program Director/Professor, Medical Laboratory Science, University of Cincinnati, Cincinnati, Ohio; Chair, Microbiology Advisory Committee, International Federation of Biomedical Laboratory Science; Editor in Chief, International Journal of Biomedical Laboratory Science, International Federation of Biomedical Laboratory Science; President, American Society of Clinical Laboratory Science.
Bibliographic information
| | |
--- |
| Title | Bailey & Scott's Diagnostic Microbiology |
| Author | Patricia M. Tille |
| Edition | 15 |
| Publisher | Elsevier Health Sciences, 2021 |
| ISBN | 0323681069, 9780323681063 |
| Length | 1184 pages |
| Subjects | › Allied Health Services › Medical Technology Medical / Allied Health Services / Medical Technology |
| | |
| Export Citation | BiBTeX EndNote RefMan |
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2609 | https://nvlpubs.nist.gov/nistpubs/Legacy/TN/nbstechnicalnote667.pdf | • W ^ -fc NBS TECHNICAL NOTE 667 U.S. DEPARTMENT OF COMMERCE / National Bureau of Standards QC /OO .6/5753 />o. 6)67 F>t.
I / 97 5 Upper-Bound Errors in Far-Field Antenna Parameters Determined From Planar Near-Field Measurements PART 1: Analysis NATIONAL BUREAU OF STANDARDS The National Bureau of Standards 1 was established by an act of Congress March 3, 1901.
The Bureau's overall goal is to strengthen and advance the Nation's science and technology and facilitate their effective application for public benefit. To this end, the Bureau conducts research and provides: (1) a basis for the Nation's physical measurement system, (2) scientific and technological services for industry and government, (3) a technical basis for equity in trade, and (4) technical services to promote public safety. The Bureau consists of the Institute for Basic Standards, the Institute for Materials Research, the Institute for Applied Technology, the Institute for Computer Sciences and Technology, and the Office for Information Programs.
THE INSTITUTE FOR BASIC STANDARDS provides the central basis within the United States of a complete and consistent system of physical measurement; coordinates that system with measurement systems of other nations; and furnishes essential services leading to accurate and uniform physical measurements throughout the Nation's scientific community, industry, and commerce. The Institute consists of the Office of Measurement Services, the Office of Radiation Measurement and the following Center and divisions: Applied Mathematics — Electricity — Mechanics — Heat — Optical Physics — Center for Radiation Research: Nuclear Sciences; Applied Radiation — Laboratory Astrophysics 2 — Cryogenics" — Electromagnetics- — Time and Frequency 1 '.
THE INSTITUTE FOR MATERIALS RESEARCH conducts materials research leading to improved methods of measurement, standards, and data on the properties of well-characterized materials needed by industry, commerce, educational institutions, and Government; provides advisory and research services to other Government agencies; and develops, produces, and distributes standard reference materials.
The Institute consists of the Office of Standard Reference Materials, the Office of Air and Water Measurement, and the following divisions: Analytical Chemistry — Polymers — Metallurgy — Inorganic Materials — Reactor Radiation — Physical Chemistry.
THE INSTITUTE FOR APPLIED TECHNOLOGY provides technical services to promote the use of available technology and to facilitate technological innovation in industry and Government; cooperates with public and private organizations leading to the development of technological standards (including mandatory safety standards), codes and methods of test; and provides technical advice and services to Government agencies upon request. The Insti-tute consists of the following divisions and Centers: Standards Application and Analysis — Electronic Technology — Center for Consumer Product Technology: Product Systems Analysis; Product Engineering — Center for Building Technology: Structures, Materials, and Life Safety; Building Environment; Technical Evalua-tion and Application — Center for Fire Research: Fire Science; Fire Safety Engineering.
THE INSTITUTE FOR COMPUTER SCIENCES AND TECHNOLOGY conducts research and provides technical services designed to aid Government agencies in improving cost effec-tiveness in the conduct of their programs through the selection, acquisition, and effective utilization of automatic data processing equipment; and serves as the principal focus within the executive branch for the development of Federal standards for automatic data processing equipment, techniques, and computer languages.
The Institute consists of the following divisions: Computer Services — Systems and Software — Computer Systems Engineering — Informa-tion Technology.
THE OFFICE FOR INFORMATION PROGRAMS promotes optimum dissemination and accessibility of scientific information generated within NBS and other agencies of the Federal Government; promotes the development of the National Standard Reference Data System and a system of information analysis centers dealing with the broader aspects of the National Measurement System; provides appropriate services to ensure that the NBS staff has optimum accessibility to the scientific information of the world. The Office consists of the following organizational units: Office of Standard Reference Data — Office of Information Activities — Office of Technical Publications — Library — Office of International Relations — Office of International Standards.
1 Headquarters and Laboratories at Gaithersburg.
Nfaryland.
unless otherwise noted: mailing address Washington, DC.
20234.
- Located at Boulder, Colorado 80302.
Upper-Bound Errors in Far-Field Antenna Parameters Determined From Planar Near-Field Measurements PART 1: Analysis A.D.
Yaghjian Electromagnetics Division Institute for Basic Standards National Bureau of Standards Boulder, CO 80302 Sponsored by Air Force Avionics Laboratory Air Force Wright Aeronautical Laboratories Air Force Systems Command Wright-Patterson Air Force Base, Ohio 45433 ;\ OT BTANOAfiDf tTBRART DEC 3, 1 1975 Qc/co , US7S3 ftot 6 6 7 p-f I U.S. DEPARTMENT OF COMMERCE, Rogers C.
B.
Morton, Secretary James A. Baker, III, Under Secretary Dr. Betsy Ancker-Johnson, Assistant Secretary for Science and Technology NATIONAL BUREAU OF STANDARDS, Ernest Ambler, Acting Director Issued October 1975 NATIONAL BUREAU OF STANDARDS TECHNICAL NOTE 667 Nat.
Bur.
Stand.
(U.S.), Tech Note 667, 120 pages (Oct.
1975) CODEN: NBTNAE For sale by the Superintendent of Documents, U.S. Government Printing Office , Washington, DC. 20402 (Order by SD Catalog No.
C13. 46:667) $1.85 CONTENTS Page List of Figures and Tables iv I.
Introduction 1 CI.
Relationship of Errors in Gain Function, Sidelobe Level, Polarization Ratio, and Beamwidth to the Far-Field Error in Electric Field 4 II.
Error Analysis 8 A.
Finite Scan Errors 8 1.
Mathematical Formulation of the Problem 9 2.
The Fields of Electrically Large Aperture Antennas 13 3.
Evaluation of n(r) 18 B.
Position and Instrumentation Errors 30 1.
Position Errors 33 2.
Instrumentation Errors 48 C.
Multiple Reflections 58 IV.
Summary 69 Appendix A --Near-Fields of a Circular Antenna with Uniform Aperture Distribution 96 Appendix B --Position and Instrumentation Errors for the Null Depth of Difference Patterns 105 References 111 Acknowledgment 113 in LIST OF FIGURES AND TABLES Page Figure 1.
Main beam of a hypothetical test antenna 82 Figure 2.
Schematic of scanning geometry 82 Figure 3.
Schematic of aperture antenna 83 Figure 4.
Definition of a , 9 , D 83 to m ' m ' m Figure 5.
Schematic of aperture and scan areas 84 Figure 6.
Circular reflector antenna 84 Figure 7a.
Near-field centerline data (constrained lens) (z = 25 cm) 85 Figure 7b.
Near-field centerline data (reflector antenna) (z = 43.18 cm) 85 Figure 8a.
Change in gain vs.
decreasing scan length (constrained lens) 86 Figure 8b.
Change in gain vs.
decreasing scan length (reflector antenna) 86 Figure 9a.
Constrained lens sum port near-field log ampli-tude, f = 9.2 GHz, z = 25.0 cm, no radome 87 Figure 9b.
Near-field phase, constrained lens sum port, f = 9.2 GHz, z = 25 cm 87 Figure 9c.
Constrained lens antenna and probe 87 Figure 9d.
Near-Field Centerline Data, z = 25 cm 88 Figure 10.
Position errors in on-axis gain 89 Figure 11.
Comparison with Rodrigue et al .
for random phase errors 89 Figure 12.
Comparison with Newell for quadratic phase errors 90 Figure 13.
Amplitude errors in on-axis gain 90 Figure 14.
On-axis error in far-field from multiple reflections 91 Figure 15.
Far-field errors from multiple reflections (a/A = 12) 91 IV LIST OF FIGURES AND TABLES (continued) Page Figure Al .
Definition of t and a 102 Figure A2.
Dotted line shows region in which eq .
(A12a) holds 102 Figure A3.
Near-field amplitude of circular antenna 103 Figure A4 .
Near-field phase of circular antenna (a/A = 12)-104 Table 1.
Finite Scan Errors from Centerline Data of a Typical X-Band Antenna 92 Table 2.
XY-and Z-Position Errors for a Typical X-and K-Band Antenna 9 3 Table 3.
Instrumentation Errors in Measuring Amplitude 94 Table 4.
Total Far-Field Errors for a Typical X-and K-Band Antenna 9 5 v UPPER-BOUND ERRORS IN FAR- FIELD ANTENNA PARAMETERS DETERMINED FROM PLANAR NEAR- FIELD MEASUREMENTS Part I : Analysis A.D.
Yaghj ian ABSTRACT General expressions are derived for estimating the errors in the sum or difference far-field pattern of electrically large aperture antennas which are measured by the planar near-field scanning tech-nique.
Upper bounds are determined for the far-field errors pro-duced by 1) the nonzero fields outside the finite scan area, 2) the inaccuracies in the positioning of the probe, 3) the distortion and nonlinearit ies of the instrumentation which measures the amplitude and phase of the probe output, and 4) the multiple reflections.
Computational errors, uncertainties in the receiving characteristics of the probe, and errors involved with measuring the input power to the test antenna are briefly discussed.
Key words: Antennas; error analysis; far-field pattern; near-field measurements; planar scanning; planewave spectrum.
vr I .
Introduction The planar near-field scanning method has been used to measure with high accuracy the electromagnetic fields of microwave antennas.
High accuracy is possible primarily because so few restrictive approximations are involved in the formulation and application of the near-field techniques.
Moreover, the far-field errors associated with each approximation can be estimated because the approximations themselves can be expressed in convenient mathematical form.
This report essentially derives and evaluates some of these mathematical expressions under the given laboratory conditions to determine quan-titatively the accuracy with which the far-field of antennas can be measured by the planar near-field scanning method.
The error analysis was undertaken for two main reasons.
One, it was desired to find general upper-bound expressions for the limits of accuracy in computing far-fields from planar near-field measurements without resorting to direct far-field comparisons.
It has long been the feeling of those involved with the near-field measurement techniques at the NBS that these techniques were often more accurate than measurements taken on conventional "far- field" ranges with a standard antenna.
Thus comparisons with patterns measured on conventional far-field ranges would not give a re-liable evaluation of the near-field techniques which do not have to cope with proximity corrections, ground reflections, or the cali-bration of standard far-field antennas.
Two, the upper-bound ex-pressions could be used to stipulate design criteria for the con-struction of new near-field scanning facilities.
This meant that the upper-bounds for the accuracy in a given far-field parameter should be expressed in terms of measured near-field data and/or the computed far-field, the frequency and dimensions of the antenna-probe system, the variation in the positioning of the scanner, and the precision of the instrumentation which measures the probe output.
The design engineer could then compute from the upper-bound ex-pressions the near-field tolerances required to insure a given far-field accuracy for the range in size and frequency of the antennas he was considering.
There have been two major computer studies performed in the past to estimate the errors involved with planar near-field scanning measurements [11,16].
Rodrigue, Joy and Burns have introduced position and instrumentation errors into a hypothetical near-field distribution in order to compute their effects on the far-field.
Newell and Crawford [11,17] performed a similar analysis with mea-sured near- field data which included an estimate of errors involved with truncating the scan plane.
(Jensen has also made a numerical investigation into the accuracy of near-field measurement techniques but for nonplanar scanning only.) A major drawback of the computer studies, as well as direct far-field comparisons, is that they apply to particular antennas and their results do not necessarily represent upper-bounds to errors which will hold for large general classes of antennas.
It should be emphasized, however, that the computer studies are an extremely useful aid in giving direction to the general analysis and checking its results.
This report does not attempt to estimate the accuracy of the extrapolation technique for measuring the gain of antennas .
An error analysis involving the extrapolation technique has been performed recently by Kanda .
If we assume that the antennas under consideration are linear, of finite extent, and operating in a single mode at fixed frequency and amplitude, and that Maxwell's equations for free space describe the region in which the antenna is situated, the only approxima-tions involved in formulating and applying the planar scanning method are : 1) The fields outside the finite scan area are zero.
2) The scanner is aligned and positioned with infinite precision .
3) The instrumentation introduces no distortion and measures the amplitude and phase of the probe output with perfect accuracy .
4) Multiple reflections between the test antenna and "probe" antenna are zero .
5) Computation errors in "deconvolut ing" the measured near-field data to get the far-field are nil.
Errors caused by uncertainties in the receiving characteristics of the probe, and errors involved with measuring the input power to the test antenna are discussed in Section IV, which, incidently, contains a summary of the major results and conclusions of this report.
The far- field errors introduced by the computations (approxima-tion 5) are so much smaller than the combined errors in the far-field caused by a finite scan area, positioning, instrumentation, and multiple reflections (approximations 1-4) that they are of little consequence.
With the help of the sampling theorem, Fast Fourier Transform, and computers accurate to many places, the necessary deconvolution of the measured data can be performed with insignificant error.
The amplitude of the near-field multiple reflections (approxi-mation 4) can be estimated in practice by changing the distance be-tween the probe and test antennas.
Any periodic variations in the amplitude of the received signal repeating about every A/2 (A = wavelength) would be caused primarily by the multiple reflections.
In Section III.C the effects on the far-field of multiple reflections are discussed and estimated analytically.
Although they cannot be eliminated completely, multiple reflections can be reduced by using efficient absorber material, by decreasing the size of the probe, or by increasing the distance between the probe and test antenna.
Also, it is likely that the effect of multiple reflections on the far-field can be reduced by averaging the far-field patterns ob-tained by scanning on a number of near- field planes which are separated by a small fraction of a wavelength.
Far-field errors caused by the errors in the scanner position-ing and instrumentation (approximations 2 and 3) are determined from a common set of equations.
These equations are derived and eval-uated in Section III.B for both systematic and random near-field Without the advantage of the FFT, typically the computer would add 20,000 terms to calculate the variable which yields the far-field.
Even if we make the absurd hypothesis that every round-off error adds in the same direction, a computer of 10 -place accuracy adding 20,000 numbers would retain more than 5-place accuracy (.001%) for the sum.
Of course, in practice the FFT algorithm vastly reduces the number of necessary computations.
There is also the approximation associated with applying the samp-ling theorem, which assumes that the output of the probe is the Fourier transform of a band-limited function.
For a separation dis-tance between test antenna and probe of more than a few wavelengths, this can be shown to be an extremely good approximation which intro-duces negligible errors into the far-field.
errors.
"Position errors" may be reduced by scanning along both vertical and horizontal lines and appropriately averaging the two sets of measurements.
"Instrumentation errors" can be reduced if the distortion and nonlinearit ies of the receivers can be determined and included as part of the computer program that deconvolutes the near-field data.
The first portion of the error analysis (Section III. A) is devoted to determining the maximum far- field errors introduced by neglecting the fields outside the scan area.
As part of the analy-sis, asymptotic expressions are derived for the near-field in front of large aperture radiators .
All errors are assumed small enough so that the individual con-tribution to the far-field error from each source of near-field error can be estimated independently and then combined to give the total far-field error.
We shall find that all the individual errors in the far- field combine linearly except for the on-axis sum pattern position and instrumentation phase errors which combine quadratic-ally (see footnote 12).
1 1 .
Relationship of Errors in Gain Function, Sidelobe Level, Polarization Ratio, and Beamwidth to the Far-Field Error in Electric Field Let E (r ) denote the electric field (to within the limits of error) of an antenna radiating into free-space and ± AE(r) the limits of error involved with the measurement of E(r) (e time dependence has been suppressed).
The magnitude of the fractional error n(r) in the electric field amplitude can be defined for small errors as , n(r) = E±AE AE(r) |E(F) (1) (Henceforth, the < sign will be omitted when eq .
(1) or similar ex-pressions are used.) The Hermitian amplitude |E| = /E # E (the asterisk denotes the complex conjugate) is related to the power S radiated per unit area in the far-field or any other locally plane-wave field in free-space by S = /e /y | E | 2 From the values of n(r) as r approaches infinity and the approximate amplitude of the far-field |E(r)| , all other errors in far-field quantities can be found.
Four such quantities of interest are the gain func-tion, the sidelobe level, the polarization ratio, and the half-power beamwidth.
Of course, these four are not always the only far-field quantities of interest, but they will be used as typical examples to demonstrate how the error in any far- field quantity can be re-lated easily to n(r) and |E(r)| .
(In eqs .
(2)-(5) below, where the error in these four quantities is derived, the subscript M r+°°" is understood, but not shown explicitly.) The error n^ in the gain function G(r), which is proportional to the square of the far-field amplitude, may be written immediately as (for small |AE~|/|e|, so that |AE| 2 terms can be neglected) E ± AE El 2 1 I AE , G = ± 2 U±Ba.
g (2a) or in decibels = ±2n(r)G(r), dB _ -, n -.
n G = 10 log E ± AE 2 ^ = 20 log(l±n(r)) (2b) For small errors, n << 1 , and (2c) The sidelobe level, when meaningful, is usually stated in deci bels and is defined as the ratio of the maximum far-field intensity of the largest sidelobe to the maximum far-field intensity of the main beam For SL = 'side The error r\ in sidelobe level SL is s 'side ± AE side side E ± AE max max E | 2 max ' VIEmax and small r\ (r) , r, s = ±2[n(r s ) + n(r Q )]SL, (3a) with n(r ) and n(r ) denoting the values of n at the sidelobe maxi mum and main beam maximum respectively.
In most cases n(r ) will be much larger than n (r ) so that r\ = ±2ri(r )SL.
(3b) In decibels dB n = 20 log side AE s ide E ± AE max max log 'side max or for small n , and n (r ) >> n (r ), dB 7 n(r s ) (3c) i.e., as we might expect, simply the error in the gain of the side-lobe itself.
The polarization ratio (Pol) of the electric field is defined as the ratio of the minor to major axis of its polarization ellipse .
For a given amplitude of error field | AE | the maximum error in the polarization ratio at a point in the far-field occurs when the field AE has the same polarization ratio as E but with major and minor axes interchanged and the direction of rotation of the electric-field vector reversed. (Also the electric field vectors of AE and E must be colinear along the major-minor axes.) Under these conditions, the maximum error (APol) in polarization ratio is realized and can be expressed for small errors in the form (4a) APol = ±[1 + (Pol) l ] n(r) .
The upper-bound error in polarization is never greater than 2n (cir-cularly polarized) and for antennas with small cross polarization the upper-bound error is approximately n .
It is an interesting result that if the maximum error in polari-zation ratio were realized, the rotational shift in the polarization ellipse caused by the error field would be a minimum, i.e., zero.
Fortunately, as part of the derivation of eq.
(4a) the maximum possible rotational shift \b in the polarization ellipse for a r r max r r given | A E | can also be determined.
Specifically, 4>max 1 + (Pol) 2 ' (PoI) 2 J radians (4b) For a far-field of approximately circular polarization (Pol -1) the shift in polarization ellipse introduced by the error field may This result, the proof of which is straightforward but rather lengthy, was obtained by Flemming Jensen and brought to my attention by Flemming Larsen both of the Technical University of Denmark, Lyngby, Denmark.
become large --as one might expect.
Finally, it is mentioned that footnote 11 should be consulted when applying eqs .
(4) to instrumen-tation amplitude errors.
The half-power beamwidth is defined for a plane containing the line along the maximum intensity of the beam.
It is simply "the angle between the two directions in which the radiation intensity is one half the maximum value of the beam" .
If we know E(r) and AE(r) in the far- field we can determine the maximum errors in measur-ing the beamwidth.
In figure 1 and |E ± AE 2 E AE (neglecting | AE | 2 terms, as usual) are plotted for the main beam of an arbitrary test antenna.
The beamwidth is 0,+6 2 .
From figure 1, we see that the error in 9-.
is A0 1 = ± 2|AE(9 1 )| |E(6 1 ) AE(6 1 )| |E(8 1 ) tan a E(9 1 ) d|E(9 1 ) ±n(9 1 )|E(9 1 ) dlECe^l radians . ~dQ d9 A corresponding expression holds for the error in A9~.
The total fractional error n R in beamwidth may be written (5a) (5b) or, for symmetrical beams (9=9,= 9~), simply n B = ±A9 9 .
±ri(9) E(9) fi d|E(9)| d9 For large circular or square apertures of uniform amplitude and phase distribution, |F(0)|/ e^-L equal 1, and n™becomes simply, phase distribution, |E(0)|/ 9 L-t4 j [ can be shown to approximately ! B :n(8) (5c) As expected, eqs.
(2)- (5) verify that a knowledge of |E(r)| and n(r) as r -» °° (as mentioned above, the "r > °°" is suppressed in eqs.
(2) -(5)) is all that is required to determine the errors in the gain function, sidelobe level, polarization ratio, and beamwidth.
If desired, errors in other far-field parameters can equally well be expressed in terms of |E(r)| and n (r) .
The far-field |F| may be computed from the measured near-field data or estimated analyti-cally.
Thus, the problem of finding the far-field errors reduces to the problem of evaluating n (r ) of eq.
(1) in the far-field.
Here we are neglecting the error caused by the change in the_maximum value of the beam.
If this error is significant compared to AE(9,) it should be added to it .
III.
Error Analysis A.
Finite Scan Errors The purpose of the present section is to estimate the maximum errors in the far-field introduced by scanning in the near-field over a plane of finite area.
"Finite" is the key word.
In prin-ciple, the planar scan method requires that the output of the probe be recorded over an infinite plane in front of the test antenna.
In practice, of course, only a finite area of the plane is scanned and the fields outside that finite area are set equal to zero.
The scan area is usually, but not necessarily, rectangular with the boundaries commonly chosen where the output of the probe antenna is down 30 or 40 dB or more from its maximum.
For many microwave an-tennas such a scan area turns out to be about two or three times the aperture area of the antenna.
The present analysis will be re-stricted to electrically large (average width/wavelength > 10 will do) aperture-type antennas, usually but not necessarily operating at microwave frequencies.
The phase will be assumed fairly uniform across the aperture with the amplitude of the field either uniform or reaching a maximum near the center of the aperture and convexly 2 tapered toward the edge to reduce the sidelobe radiation .
Near-field, centerline, amplitude and phase data for the sum pattern of a typical antenna measured at NBS can be seen in figure 9d .
Re-flectors, large horns, and broadside arrays are probably the most common examples of the test antennas under consideration.
The re-sults of this finite scan part of the error analysis apply to antennas with their boresight direction steered at an arbitrary angle with respect to the scan plane.
Initially we are assuming the antenna is operating in a sum mode or pattern.
For finite scan errors it will be shown that difference patterns need not be considered separately since they are formed by the superposition of two sum patterns with wavefronts slightly skewed to create a "null" in the boresight direction.
For position and instrumentation errors, however, Section III.B along with Appendix B shows that sum and difference patterns must be con-sidered separately, at least when determining far-field errors near the boresight direction.
The approach which is used to estimate the finite scan errors involves finding an upper-bound to the appropriate integral of the fields outside the finite scan area.
Physical optics and the geo-metrical theory of diffraction are used to show that the fields out-side the scan area are determined chiefly by edge diffracted fields of the antenna.
For each antenna these edge diffracted fields are different.
However, they all can be expressed in a general form (eq.
(19)) which allows upper-bound expressions (eqs.
(32) and (36)) to be found by evaluating the integral outside the scan area in terms of the probe output on the edge of the scan area.
Outside the "solid angle" formed by the edge of the aperture and edge of the scan area, the evaluation of the integral can be done by the method as stationary phase to show that in this region the far-fields com-puted from the near-field data cannot be relied upon with any con-fidence.
Well within the solid angle the integral evaluation can be performed through integration by parts to yield an upper-bound expression for finite scan errors from both centerline data scans (eq.
(32)) and full scans (eq.
(36) or (32)).
1 .
Mathematical Formulation of the Problem Suppose we want to determine the radiation pattern of a given antenna of aperture area A bounded by the curve C, as in figure 2.
The task is accomplished experimentally by recording the output of an arbitrary but known probe antenna (for two orientations, in general) as the probe scans in front of the radiating test antenna on a plane of area A' bounded by C 1 .
The z-axis will always be chosen perpendicular to the scan plane with origin in the antenna aperture.
The scan plane, however, may not always be chosen parallel to the aperture plane.
For beams steered off -axis, larger scan areas may be required if the scan plane does not lie perpendicular (approx-imately) to the boresight direction.
After taking a double Fourier transform of the probe output in each orientation, the radiating characteristics (S, (K) ) of the test antenna are found simply by solving simultaneously the two resulting linear equations.
In particular, if the probe were a perfect dipole, the output of the probe would be proportional to the electric field at the dipole in the direction of the dipole moment.
The expression for S, (K) (defined with respect to the reference plane z = 0, and the transverse part of the propagation vector denoted by K) then becomes S l0 (K) = —7— e iYd / E (P,d)e" iK ' P dP, (6) o f - ioot j , .
(e time dependence) where E (P,d) is the electric field, i.e., the output of the dipole probe in two mutually perpendicular orientations, transverse to the z-axis at the point (P,d) in the scan plane A'.
(The equation of the A' plane is z = d.) The amplitude of the input mode to the test antenna is designated by a , and the variable y is defined by 1, 7 TT Y = (k 2 -K 2 )'2 (k = 00/c = —r) , where the radical is chosen to keep y positive real or imaginary.
' For the sake of simplifying the theory we will assume a perfect electric dipole as the probe.
The dipole gives information about the electric field only at a point.
All physically realizable probes respond to a weighted average of the field near the probe.
Thus it is expected that the errors in the computed far-field introduced by omitting a part of the infinite scan plane are as great or greater for a perfect dipole than for any other probe antenna, and that the following conclusions and resulting upper bound expressions (32) and (36) hold for arbitrary probes.
Also, at this point in the error 3aThe sampling theorem shows that to obtain the far-field pattern the double integral in eq .
(6) can essentially be replaced by a double summation over points in A' separated by about A/2 or less.
Thus, in practice, data need be taken only at a finite number of discrete points for eq .
(6) to be evaluated.
However, for most of the error analysis_we prefer (somewhat arbitrarily) the integral representation of S.
to the summation.
r lo Strictly speaking, the Fourier transform in eq.
(6) and eqs .
(7), (10), and (11) below may not converge^ to a unique value as the scan area approaches infinity because E+-(P,d) has a 1/P dependence in a lossless medium as P->°°, and thus gives rise to a rapidly oscillating part in the transform as P-».
Usually this oscillatory term can be ignored with impunity because it vanishes upon integration when taking the inverse transform.
However, it does determine the limit-ing value of the finite scan errors (see footnote 9).
10 analysis any uncertainty in the receiving characteristics (S') of the measuring probe are ignored.
For eq.
(6) to represent S, (K) exactly, the probe scan and subsequent integration would have to be performed over the infinite plane.
Thus the error (AS, ) produced in S, by a scan of finite r lo^ r lo J area may be written formally as AS-= —^— e" iYd / E (P,d)e~ iK ' P dP, (7) 10 4^ 2 a A' t o c where A 1 is the infinite area outside or complementary to A' .
The far-fields can be found from S, (K) by taking the inverse double Fourier transform of eq .
(6) and evaluating the resulting expression by the method of stationary phase for double integrals.
So doing yields ikr E.(r) = -27ia ik cosB S, (kR/r) (8) ^ v j lo v J r X" -> co and from eq.
(8) ikr AE^(r) = -2iTa ik cosGAS, (kR/r) .
(9) t K J o lo v r r -> oo (cos9 = z/r, z>0) Substitution of S, and AS, from eqs.
(61 and (7) into eqs.
(8) and lO lO ivy v.
j -i v j (9), respectively, produces expressions for the far -electric-field and its error in terms of the near-field data: E (7) = - ik C056 e ik(r - d C0S ^/ E t (P,d)e" lFR ' P dP (10) Z 27rr A' r X" --°o AEJ7) = -i\^-e ik(r " d COs9) / E t (P,d)e" lFR ' P dP.
(11) c X -> oo (For a dipole probe, E (P,d) and the output of the probe are identi-cal.) Equations (10) and (11) express mathematically the well-known 11 "Fourier optics" result that the far-field amplitude is proportional to the spatial Fourier transform of the near-field times cos0.
Division of eq .
(10) by eq.
(11) results in the fractional error r\ in the transverse part of the far-electric -field n t (r) AE t (r) E t (r)| -i-R-P / E.(P,d)e r dP A' z -i-R-P | E (P,d)e r dP A' Z (12a) or simply cos6 n t (r) = -i- R-P { E. (P,d)e r dP A' z __c Ar |1\ (r) | (12b) Of course, the fractional error r\ in the total far-electric field is not necessarily n f except on the z-axis.
However we can find n in terms of n f by a simple argument.
In the far-field |E (r) differs from |E(r)| at most by a factor cosG, i.e.
|E(r) |cos6 _< |E (r) | < |E(r) Similarly AElcosG < AE t | < AE (12c) (12d) Consequently ri is greater than n t by at most a factor Q , and from eqs .
(12) we can express the maximum possible n in the far-field as n(r) = AE / E.(P,d) A' Z -i- R-P e r dP Xr|E(r)| (13) •]"->• 00 For the denominator of eq .
(13) we can use the far-field estimated analytically or computed from the measured near-field data.
Thus, the problem of finding n reduces to that of estimating E (P,d) on A^, i.e., on the area outside the finite scan plane.
12 2 .
The Fields of Electrically Large Aperture Antennas If the behavior of the electric field E (P,d) were known in the area A, the integral of eq.
(13), r I = / E.(P,d")e dP, (14) A t L Ac could be evaluated and the far-field errors could be found immediately from eqs .
(l)-(5).
Even asymptotic methods cannot be applied to eq .
(14) until the behavior of E (P,d) is determined.
Fortunately, the electric field outside the aperture of electrically large aperture antennas can be determined analytically from the electric field dis-tribution across the aperture.
In fact, it will be shown shortly that just the electric field at the edge of the aperture distribution is required.
As a first step in finding the electric field E (P,d) to use in eq.
(14), consider the aperture antenna drawn schematically in figure 3.
(The boundary or rim of the antenna is assumed to lie in a plane with e here chosen perpendicular to the plane.
These restrictions are relaxed later.) It is well-known that the electric field everywhere to the right of the infinite plane A^ can be expressed in terms of an integral of the electric field over A^, EOO = - ^ / [e z xE t (R')] x VG(F,R») dR' , (15) 00 where lklr-R' I G = SL-!
' |r-R»| and E. (R' ) is the transverse electric field on the plane A^ emanat-ing from sources to the left of A^ (e.g., electric fields from a feed located to the right of A^ would not be included in E.(R'), except, of course, indirectly as reflected fields from the antenna).
For an aperture antenna, whether it be a reflector, large horn, or broadside array, the components of the electric field E. (R T ) are 13 4 slowly varying in phase and amplitude across the aperture, except possibly right near the edge.
The amplitude often tapers toward the edge of the aperture and drops abruptly (in a distance less than a wavelength) to zero or near zero beyond the edge.
Thus the limits of integration in eq.
(15) reduces to the aperture area A or at most a couple of wavelengths beyond A.
Also, since eq.
(14) requires only the transverse component of E (r) , we can ignore the z-components of eq.
(15) and write E t (r) = ' 17 |y / E t (R')G(F,R') d R' .
(16) For electrically large apertures, eq .
(16) can be evaluated asymp-totically.
Before doing this a couple of remarks are in order.
The first has to do with neglecting the fields outside the aperture area A when in reality there may be scattering from the edge of the antenna aperture.
In those cases it may seem unreasonable to neglect, initially, the fields beyond the aperture area A as part of the procedure to calculate the fields beyond the scan area A' .
The "canonical" problems that can be solved exactly, such as scatter-ing by an infinite wedge or elliptical disk, indicate, however, that at short wavelengths the scattered fields are caused predominantly by high intensity fields within a wavelength or so of the edge, and indeed the fields more than this distance beyond the edge may be neglected .
Of course, this assumption has been confirmed by experiment and constitutes the basic postulate of the geometrical theory of diffraction .
In terms of the radiation from large aperture antennas, it means simply that the fields everywhere to the right of the aperture plane depend solely upon the fields within and near the edge of the aperture, even when appreciable scattering from the edges is present.
The second remark concerns antennas that have part of their feeding mechanism mounted to the right of the aperture plane--as in the case of reflector antennas.
As mentioned above, the direct radiation from the feed must not be included as part of E.(R') for eq .
(16) to be correct.
However, scattering from feed mounts 4For electrically steered arrays the planes of "uniform" phase may be skewed with respect to the aperture plane.
This situation is considered at the end of the section.
14 (including the feed itself) of the fields reflected by the antenna are not taken into account by eq .
(16).
Although the mount scattered fields can be ignored if they are small compared to the fields scattered from the aperture edge, there is no reason to believe this will always be the case even in the region A'.
Fortunately, the results of the geometrical theory of diffraction (GTD) can also be used to determine the behavior of the fields diffracted from the mounting in front of the aperture as well as from any sharp edges at the boundary of the aperture .
Moreover, we shall find that the final expression (eqs.
(32) and (36)) for the far-field error does not require an explicit estimation of either the mount or edge diffracted fields.
Of course, for many antennas such as horns and arrays there are no obstructions in front of the aperture.
For the moment we shall ignore the problems of the exact nature of the edge diffraction and diffraction from feed mounts and return to evaluate eq.
(16) for a smoothly tapered amplitude within A and zero outside A.
For apertures which are many wavelengths across, the double integral (16) can be expanded in an asymptotic series.
The first three terms in the series for integrals like eq.
(16) have been derived by Van Kampen .
Keller, Lewis and Seckler apply Van Kampen' s results to eq.
(16) specifically.
The final expression has been confirmed by the present author using an approach different from Van Kampen' s.
In the shadow region, i.e., the entire half space z > excluding the cylindrical volume formed by the projection of the aperture area A along the z-axis, the electric field in eq.
(16) is approximated by / E\ (F) = I -t v J L 2tt m am a +D sin6 mm W H cos9 m E.
m tm e sin0m m 4 (17) The variables in eq.
(17) are defined with the help of figure 4.
The distance from the point r to the edge of an aperture with a smooth boundary has at least one relative maximum and one relative minimum.
The subscript m simply refers to the mth relative extremum point on the edge of the aperture.
D is the distance from r to the ° r m mth relative extremum, a is the radius of curvature of the edge (in 7 m b K 15 the plane of the aperture) at the mth relative extremum, 9 is the _ m angle between D and the z-direction, and E is the value of the transverse electric field at the edge of the aperture excluding the fields diffracted from the edge.
The radius of curvature a is taken b m positive if the distance D is a relative minimum and negative if a am 1 1/2 m relative maximum.
The radical a +D sin0 mm m is taken positive if real and negative if imaginary.
Of course, all subscripted quanti-ties are, in general, functions of position r.
Actually, in order for expression (17) to remain valid, must be greater than a few A/£ (I e /A) , and D no closer than a couple of wavelengths X from the edge.
This latter restriction says simply that eq.
(17) does not describe the reactive fields.
The former restriction can be understood physically by dividing the aperture into Fresnel zones for the direction .
For greater ave mm 6 than a few X/l there are enough Fresnel zones (even for the far-field) to assure that the main contribution to the fields in the shadow region are from the fields near the edge of the aperture rather than from the fields well within the aperture.
The expression (17) does not necessarily account correctly for scattering from edges that may be present on the boundary or rim of the antenna.
The necessary modification of eq .
(17), for example, when scattering from sharp conducting edges occurs can be extracted from Kellers GTD .
Kouyoumj ian [9a] has written Keller's results in a form similar to eq.
(17).
The GTD expression differs from eq.
(17) only in that the factor cos0 E.
is replaced by v J J m tm r J l' 1 g (0 ) + e}-g. (0 ), (18) tm 5 II v nr tm 6_L V nr ' K J —II —J-where El and E^ are the transverse components of the incident tm tm r electric field parallel and perpendicular to the edge of the aper-ture.
The factors g and g can be found from either reference [6a] or [9a] , but it is not necessary to know them explicitly for our purposes.
Moreover, scattering from other than sharp conducting edges can also be handled by changing appropriately the factors & and gj_ in eq.
(18) .
5Note that in a similar manner it can be argued that the far-fields for less than a couple X/£max (&max = maximum width of aperture) are determined mainly by the near- fields well within the boundary of the aperture.
16 Thus it is seen from eqs .
(17) and (18) that regardless of the nature of the scattering from the edges, the fields in the shadow zone appear to emanate from points along the edge of the aperture.
For our purposes it proves convenient to write simply the one equa-tion valid for the scattered field in the shadow zone ikDm E t (r) = I F (r)e m .
(19) m F is defined by comparing eq.
(19) with eqs.
(17) and (18).
The essential property of F (r ) , which allows the asymptotic evaluation ikD — of eq.
(14), is that unlike e m it varies slowly with r.
In fact, for the error analysis it turns out that this is the only property of F (r) that is required.
Never is it necessary to evaluate F.
(r) explicitly, although for reasons of general interest it has been evaluated in Appendix A for the circular aperture of uniform dis-tribution.
Results from Appendix A are also used in Section III.B.l.
Before carrying out the integration of eq .
(14) it should be mentioned that eqs.
(17) and (18) are not valid for large D if the radius of curvature a approaches infinity.
For example, the ex-pressions would be modified if the aperture were rectangular.
How- _ ikDm ever, the modifications occur in F (r) but not e .
Thus, eq.
(19) remains valid for all shaped apertures even when part of the edge is a straight line or has infinite radius of curvature.
Also if the edge of the aperture has points where the radius of curvature is much smaller than a wavelength (e.g., the corners of a rectangular aperture) , these corners and tips contribute to the field.
For large apertures at least it can be shown [5-9] that their contribution is usually much smaller (higher order in X/ I ) than the edge fields of eq.
(19), and thus can usually be neglected for the purpose of evaluating eq.
(14).
However, even if they can-not be neglected, the fields from corners and tips can be expressed in the same form as eq.
(19).
Electronically steered aperture antennas (broadside phased arrays) are also covered by eq.
(19) with the appropriate modifica-tions of F (see, for example, reference which deals with an arbitrary phase of the field across the aperture) .
When the axis of the main beam is steered away from the perpendicular to the aper-17 ture, the region of validity of eq.
(19) changes to the entire half space in the direction of the new axis, excluding the projection of the aperture along that axis, i.e., eq .
(19) is still valid in the shadow zone regardless of what direction the beam is steered.
Eq.
(19) also applies to antennas operating in a difference pattern where the on-axis field drops to a sharp minimum.
As in the case of the sum pattern, the fields in the shadow zone are determined by whatever value of electric field impinges upon the edge of the aperture.
And again the variation in electric field separates into ikDm _ a rapidly oscillating part e and a slowly varying part F .
Eq .
(17) (or (17) modified by (18)) represents the first term in an asymptotic expansion of the electric field.
The higher order terms are assumed so much smaller than the first that they are neg-lected.
However, if the electric field at the edge of the aperture becomes too small, the second term in the asymptotic expansion must be included.
Even then, for reasonably smooth aperture distribu-tions, this second term also has the form of eq .
(19) but with F depending on the derivative of the electric field at the edge rather than the field itself.
This "slope diffraction coefficient" has been derived for the GTD by Hwang and Kouyoumj ian [9b].
In brief, eq .
(19) describes the electric field in the shadow zone of nearly all large aperture antennas including electronically steered arrays, antennas excited in a difference pattern, and an-tennas with aperture distributions that taper to zero at the edge.
3 .
Evaluation of n(r) The evaluation of the integral in eq.
(14) and subsequently n(r) in eq.
(13) is accomplished by first substituting the electric field from eq.
(19) into eq.
(14), 2tt °° ik[D -p sinBcos (4>-<> )] ; (V d) e m P tm L ' J r -' 'P I = I J "J F t fP,d)e m P ' J p dp deb.
(20) m The vectors P and r have been written in cylindrical (p,(b ) and spherical coordinates (r,0,-<f) )] occur at 3D ^ = p sine sin (cf) -cf)) (21a) ^p p 3D 3p^ = sine cos(cj)-cj) ) .
(21b) 3D Consider the derivative ^ , which can be interpreted physically by referring to figure 5.
Let the vector P be the perpendicular pro-jection of the line D such that J m Then p = /p 2 -Ji 2 .
m r m 3D 3D 3p p-£ m m ± in it m _ m _m ' m 3p 3P 3Pm 3P Pm C0SYm , m x x m where y is the angle between P and D .
Ordinarily, the scan areas 'm & mm J ' are appreciably larger than the aperture area so that 9£m Pm ~ i and (21b) may be written cosy = sine cos(cf)-(() ).
(22) 3Dm Similarly, it can be argued that -^— is much less than p for scan 3c|)p areas appreciably larger than the aperture area.
Thus for 6 not too small eq.
(21a) implies the critical point must be near $=<{>.
Now cosy has a minimum value greater than zero because y never reaches 'm b m 90°.
The minimum value of cosy occurs at the maximum value of 'm 19 ym = Ym v or on tne boundary of the scan area near for 9 < 90 -v fd>1 .
For > 90 -y critical points exist at max no cf) -<J) P 90 m m and eq.
(20) can be evaluated immediately by the method of station-ary phase for double integrals .
Such a procedure yields after some rearrangement F.
AD ik(d cos6+x sin9) f-v tm m v m J I = I -r^r^ e m cos6 (23) Actually, eq.
(23) was derived using the approximation D -/(p±x ) +d -d X and — ^—— << 1, where x and d are defined in figure 5.
The above p dcp m to approximations simplify the mathematics but do not alter by a great amount the amplitude of the final expression (nor the following conclusions) .
The implications of eq .
(23) prove to be quite significant be-cause it shows that for 9 > 90 -v () "the magnitude of the inte-max v J b gral I is of the same order as the amplitude of the electric field itself with each term multiplied by A D /cos9.
By referring back to eqs .
(12) and (13) we see that this result implies the following: In the re gion outside of the envel ope , which is formed by the ra ys running from the edge of the aperture throu gh the bounda ry of the scan area , the fractional error n(r) is on the or der of unity.
Thus , outside the envelope the fax fields COmputed from a planar :5can in the near- field cannot be relied upon with any con fidence .
Moreover, diffraction from feed mounts, if present, does not affect the above conclusion, because the radiation scattered by the feed mounts grazes the boundary of the scan area at a wider angle than the radiation from the edge of the aperture.
The conclusion says, essentially, that the planar scan tech-nique does not give information about the fields outside the solid 20 angle formed by the edge of the aperture antenna and the boundary of the scan area.
(We use the term "solid angle" loosely since it will not be a solid angle in the strict mathematical sense unless the extension of its sides meet at a single point.) As an example consider a circular reflector antenna of radius "a" scanned in the near-field on a larger concentric circular area of radius a/2.
Suppose the scan area were a distance d = a in front of the aper-ture.
Then, as shown in figure 6, the above results reveal imme-diately that the data from the near-field scan would not contain reliable information about the fields outside the angle 9 = 22.5°.
b m Newell and Crawford reached the same conclusion from ex-perimental data taken on scan planes at different distances in front of the same microwave antenna.
It appears from the above analysis that their conclusion is a general result which holds for all elec-trically large aperture radiators.
It should be emphasized that the above results were derived for the sum and difference pattern of electrically large aperture an-tennas and for a scan area that extends well beyond the main near-field beam region.
Also the main near-field beam has been assumed to be characterized by planes of fairly uniform phase.
The above conclusion would not necessarily apply, for instance, to broadbeam horns with dimensions on the order of a wavelength or less, to beams steered nearly to the edge of the scan area and scan areas just covering the main near-field beam, to apertures on a finite ground plane, to defocussed antennas, or to electrically large aperture antennas with a diverging or converging lens placed within or directly in front of the aperture.
Fortunately, special situations and classes of antennas such as these can often be analyzed sep-arately within the framework of the preceding analysis and results.
The special cases mentioned above are discussed in the following paragraph.
Specifically, an analysis similar to the preceding shows that the encircled conclusion applies to the latter three classes of antennas (the defocussed antennas and antennas with a ground plane or lens) , provided the scan area extends well beyond the edge of the antenna, the ground plane or lens, and provided the edge of the ground 21 plane or lens is used as the base perimeter for the solid angle when the edge of the ground plane or lens extends appreciably beyond the edge of the aperture and significant scattering occurs at these edges.
Also, it can be shown that the encircled con-clusion applies to broadbeam horns if the center of the horn is taken as the base of the solid angle instead of the perimeter of the horn.
(For scan areas much larger than the aperture of the horn, there is little difference in size between these two solid angles.) Similarly, for antennas with their mainbeam steered close to the edge of the scan area it may be more accurate to choose a point nearer the center of the aperture rather than the edge to determine the side of the solid angle near the direction of the main beam.
Again it makes little difference for large scan areas.
In general, when the scan area is close to the boundary of the main beam, the base of the solid angle outside of which the computed far-field pattern is unreliable tends to shift from the perimeter of the aperture (ground plane or lens) toward the center.
Often the in-crease in solid angle is slight, however.
In brief, the above encircled conclusion (in its stated or slightly modified forms just explained) applies to a very large variety of antennas, including electrically large aperture antennas operating in a sum or difference pattern (with or without beam steer-ing, defocussing, a finite ground plane, or modifying lens) and broadbeam horns.
Next we want to evaluate I of eq.
(20) for points within the solid angle formed by the edges of the antenna and scan area.
Within this solid angle the integrand contains no critical points of the first kind.
Consequently, the p integration can be done by parts to yield ik[D'-p'sin9 cos (-(J> ) ] , 2tt F (p' ,<j) )p' e m p t-wtIJ tm p <v (24) m o [cosy'-sinG cos(d)-cb )] F 3D' where again cosy' has replaced -^—^ , and the primes refer to points on the boundary of the scan area.
It should be noted that eq.
(24) represents the first term in an asymptotic expansion of eq.
(20), 22 (24) remains valid for y 1 right up to 90°.
and when cosy 1 gets too small eq.
(24) no longer represents a good approximation to eq.
(20).
We can get an idea of the largest per-missible y' by realizing that the result (24), if valid, must be much smaller in amplitude than the amplitude of the integrand of eq.
(20) multiplied by the change in distance p as D changes from D' to D' + A/2.
A little mathematics shows that this condition is always m y~Y 7 satisfied if cosy' is greater than about V-p-, where d' is the per-pendicular distance from the edge of the aperture to the scan plane.
However, when used to find an upper bound expression for |l| (see eq.
(28) below) , eq Eq .
(24) has been derived under the condition that the fields emanate or at least appear to emanate from points on the edge of the antenna aperture.
If the fields are also scattered from the feed mounts, eq.
(24) must include these fields as well associated with these mount scattered fields will always be equal to or greater than that of the edge diffracted fields--since the fields scattered by feed mounts make larger angles with the plane of the scan area at its boundary than the edge diffracted fields.
Thus, the integration by parts remains valid (as an upper bound ex-pression within the solid angle formed by the edges of the aperture and scan area), and eq .
(24) holds even when there exists appreciable radiation scattered from feed mounts in front of the antenna.
For angles near the z-axis (sin0 << cosy ) eq .
(24) becomes The cosy 1 'm I = \ 2tt 7T1 J F + e tm ikD' m cosy' 'm J p' e ikp' sin9 cos ((f) -(j) ) d (25) Upon taking the amplitude of eq.
(25) we find , 2tt ih !
2tt F.
e tm ikD' m cosym p' d(f) .
P (26) For scan areas whose boundary is well outside the edge of the aper-ture, cosy' -1.
For scan areas with boundaries somewhat close to ' 'm the edge of the aperture, the term (in the summation over m) which corresponds to the minimum D' (maximum y') predominates (assuming f m v.
i m ; f to 23 fairly uniform illumination around the edge of the aperture) , so that I m F + e tm ikD' m cosy' 'm cosy m 'max ^ Ft ikD' m m In either case, we see from eq .
(19) that iyp',y tm ikD' m cosy' m cosymax where F (p',4> ) is the electric field at the boundary of the scan area Eq .
(26) can now be written it-i .
A 2tt cosymax 2tt J o E t (p', )|p' d(j) , (27) or A Lmax 2 cosy max (28) max with |E c denoting the maximum amplitude of the transverse elec-trie field found on the boundary C of the scan area, and L is the maximum width of the scan area.
If the output of the measuring probe at the limits of the scan area is down at least X dB from its maximum output, then eq.
(28) may be rewritten A Lmax 10 X_ 20 2 cosy 'to' (29) max where E represents the highest amplitude of the transverse electric tO y g field found on the scan area.
' Equation (29) combines with eqs .
(13) and (14) to yield an upper bound expression for the error n in 'For an arbitrary probe (i.e., not necessarily a dipole probe) E-to represents the highest output amplitude of the probe on the scan area and X the largest output amplitude of the probe at the edge of the scan area measured in dB down.
^The maximum electric field on the scan area in the near-field (z << A/A) of an electrically large aperture is very nearly equal to the maximum electric field on the aperture itself.
24 the far-field produced by neglecting the fields outside the finite scan area: __X , T max ., n 2 „-f=\ nCF)<-A-L 10 _gW 9 (30a) E t 2 cosy If ^— da I 1 max ' i E .
' A to o where g(r) is the ratio of the amplitude of the maximum far-electric field to the far-electric-field in the given direction r.
In other words, it is the inverse of the normalized far-field pattern.
Use has been made of eq.
(10) which shows that for sum patterns Aril" I -1/ E\ da I , (30b) o where |E I is the amplitude of the maximum far-field, and the o r -"00 integration is performed in the near-field over the area A , that part of the beam which has nearly uniform phase.
Since we are in the near- field, A - A cos0 , where A is the aperture area.
(The factor cos0 accounts for the reduction in effective aperture area o r for beams which are steered off-axis electronically through an angle .
) 6 o Because |E. I/E < 1, where E is the highest amplitude of elec-1 t ' o — ' o 6 r trie field on the scan area, and E.
-E cos0 , we can write too o = a cos0 /A = a/A, (31) J E^- da A c to o where the factor a is greater than its minimum value of 1.0 (for apertures of uniform amplitude and phase) but less than 5 for most tapered aperture distributions found in practice.
For example, the tapered distributions found in Table IX of have a maximum a of 4.0.
Newell has found that the factor we have called a has not been greater than about 2 A/A for any microwave antenna he has measured.
Since the effective area A (see for a definition of effective area) is less than the aperture area A and greater than . 5A for most aperture antennas , the experience of Newell 25 also indicates that a is less than 4 or 5 for nearly all electrically large aperture antennas found in practice.
Equation (31) combines with eq.
(30a) to give the final expres-sion for n(r) nO) x , T max t A 2 r— -, a A L 10 g(r) 2 A cosymax (32) max max where A = area of the antenna aperture.
A = wavelength.
= maximum width of the scan area.
= maximum acute angle between the plane of the scan area and any line connecting the edges of the aperture and scan area.
= the largest amplitude of the probe output at the edge of the scan area, measured in dB down from the maximum amplitude of probe output in the scan plane .
= a "taper" factor--equal to a minimum of 1.0 (for apertures of uniform amplitude and phase) and less than 5 for most tapered distributions found in practice.
(See eq.
(31) for the pre-cise definition of a.) g(r) = ratio of the amplitude of the maximum far-electric-field to the far-electric-field at the given direction r, i.e., the inverse of the normalized far-field pattern.
(g(r) = 1 for the center of the main beam, or beams if a difference pattern.) (If desired the errors in the gain function sidelobe level, polariza-tion ratio, and beamwidth may be calculated from eqs .
(2) -(5) once n and the far- field pattern is known .
) Equation (32) has been derived for antennas operating in a sum pattern.
But since a difference pattern can be divided into two, approximately equal, sum patterns with wavefronts slightly skewed, eq.
(32) holds for difference patterns as well.
(For a difference pattern one should still use the taper factor a of the constituent sum patterns .
) In summary, eq.
(32) applies to either sum or difference pat-terns of all electrically large aperture antennas (including antennas 26 with their boresight direction steered away from the axis perpendicu-lar to the scan area) within a solid angle (sine << cosy ) about the axis perpendicular to the scan area.
Again, it has been assumed that the scan area extends well beyond a main near-field beam which is characterized by planes of fairly uniform phase.
It can be shown that eq.
(32) and eq.
(36) below apply even to defocussed antennas, apertures in a finite ground plane, and to antennas with a diverging or converging lens, provided the angle y is chosen in accord with max the second paragraph preceding that of eq .
(24).
The condition (sine << cosy ) can be made more specific by returning to eqs.
v max ^ J t>n (24) and (25).
For all practical purposes, eq .
(25) follows from eq.
(24) if (33) sine < ~- cosy = T sinf 2 ' max 2 max (6 = 90 -y ) v max 'max For example, if y were 45°, condition (33) becomes 6 < 20°, which is a large enough angle to include many side lobes of most microwave antennas (assuming their boresight direction at = 0) .
Roughly speaking, eq.
(32) represents a valid upper bound within the region < i 2 max As an upper bound, eq.
(32) remains valid for y right up to max 90°.
However, from the discussion immediately following eq.
(24), it is unlikely that eq.
(32) would remain small enough to be very useful when (34) where d refers to the minimum perpendicular distance from the mm r r edge of the aperture to the scan plane.
The error n given by eq .
(32) can be compared with the results of the empirical error analysis performed by Newell and Crawford They took "centerline" data on a near-field scan plane 25 cm in front of a circular, fixed-beam "constrained lens" array, which was 80 cm in diameter (see figure 9).
The centerline was 213 cm long.
Assuming a rectangularly separable field pattern, they first used the 213 cm centerline data to compute the far-field pattern.
Successively, more and more of the 213 cm distance was deleted and 27 the corresponding far-field pattern computed.
In that way they could get an idea of the errors involved in scanning on a near-field plane of finite area.
The envelope of their near-field centerline amplitude is repro-duced in figure 7a.
The on-axis gain change computed by Newell and Crawford by deleting distances from the scan line is reproduced by the dashed line in figure 8a.
The solid lines in figure 8a represent the maximum envelope of on-axis gain change calculated from eqs .
(32) and (2c) of the present report.
The values of X and Lmax (see eq.
(32)) were taken from figure 7a.
The value of a was estimated from eq.
(31) and figure 7a to be about 3.
The remaining parameters needed to calculate n from eq.
(32) are contained in reference : -if d Y „ = tan t r max T max — a U z J J A = 3.26 cm A = ira 2 a = 40 cm d = 25 cm g(r) 1.
Figure 8a confirms the result that the fractional error r\ of eq.
(32) represents a reasonable upper bound.
In the region max L /2a > 1.7 (y < 42°) the upper-bound error from eq .
(32) is max no more than double the error estimated by computer "deconvolut ion" of the near-field data.
The upper bound error grows inordinately mo y large, however, for L ' /2a much less than about 1.2 (y > 75°), max as eq.
(34) predicts.
Figure 8b shows the same comparison as that made in figure 8a but for a 46 cm (18 in) reflector antenna operating at 60 GHz (A = .5 cm).
(The envelope of the amplitude for a centerline scan of this antenna is shown in figure 7b.
The value of a is about 2.) The dashed line in figure 8b represents the on-axis gain change which Newell computed by deleting distances from the centerline scan length taken 43.18 cm in front of the aperture.
The agreement be-tween his computations and the upper bound solid curve calculated from eq.
(32) is even closer than for that of the constrained lens 28 H antenna (figure 8a).
The upper bound error is no more than double ) which is very the computed error for L /2a > 1.05 (y ov < « max close to the value of y < 84° predicted for the range of useful-1 max r & ness by eq.
(34) .
It appears from this somewhat limited experimental evidence with centerline data that the simple formula (32) provide s a useful upper bound error at least for scan areas with cosy > 'max It should be noted, however, that centerline data do not accountror changes in phase of the field around the perimeter of the scan area, and thus would predict larger finite scan errors in most cases than the com-plete 2-dimensional scan data.
An upper bound expression of smaller magnitude generally than eq.
(32) that takes the phase changes into account can be derived by returning to eq.
(24).
Under the condition of eq.
(33), eq.
(24) becomes — 2tt cosymax 2tt / E t (p',-<}> ) p 'p' d<|> (35) Again E (p',(J> ) refers to the transverse electric field, i.e., output of the dipole probe, at the boundary of the scan area.
Substitution of eq.
(35) into eq.
(13) yields the fractional error n (r) in the far-field for 1 max (90 Y ) 'max ; ' nO) ; 2tt -ikp'sinQ cos(-cf) ) J E t (p',cf> )e P p» d 27rr I E (r) I cosy i ^ j i r->oo 'max (36) The far-field pattern r|E(r)| in eq.
(36) can be approximated AE to r ^°° analytically by — or found by deconvoluting the measured near-aAg(r) _ field data.
The field E t (p',(j> ) at the boundary of the scan area z p can be taken from the measured near-field data.
Thus, in practice both the numerator and denominator of eq.
(36) can be determined straightforwardly.
Although eq.
(36) involves more computations than eq.
(32) even for the on-axis value for which 6=0, it could be com-puted by a simple routine added to the program which deconvolutes the near-field data, since for arbitrary probes E (p',cj) ) can be replaced by the output of the probe (for two orientations, in 29 general) en the perimeter of the scan area.
In cases where the antenna pattern is assumed separable in xy coordinates and only centerline data is taken, eq.
(36) cannot be applied but eq .
(32) still can. 9 In Table 1 are listed the errors calculated from eq.
(32) for some of the far-field parameters of a typical X-band antenna operat-ing in a sum and difference pattern.
The finite scan errors are proportional to wavelength, so changing the wavelength while holding the other antenna dimension the same merely changes the values in Table 2 proportionately.
Of course, such an isolated change is rather unrealistic.
B .
Position and Instrumentation Errors To determine the radiating fields of an unknown antenna by scanning on a near-field plane with a given probe, the output b ' (P) of the probe must be recorded throughout the scan area A' (see foot-note 3a).
In principle, the data points should lie in a plane and the position of the probe should be recorded exactly as it scans from point to point.
And, ideally, the instrumentation used to measure the phase and amplitude of b' should do so with perfect accuracy .
Obviously, in practice, neither the position of the probe nor the phase and amplitude of the probe output b' can be measured exactly.
Regardless of how small the uncertainties in the measure-ment of b'(P) they will introduce errors into the calculated far-field.
It is the purpose of this section to derive general expres-sions which estimate the magnitude of the errors in the far-field produced by the inaccuracies in measuring the position and output of the probe in the near-field.
Specifically, we want to evaluate AE in the far-field so that the fractional error n(r) of eq.
(1) can be determined.
(In Section y It is interesting to note that ri in both eqs .
(32) and (36) does not approach zero but simply an insignificantly small number as the scan boundary approaches infinity.
This limiting value of r\ represents the contribution of the oscillatory part of the Fourier transform men-tioned in footnote 3b.
Consequently, once the edges of the scan area reach the region where the fields behave as 1/p', there may be no advantage to scanning on a larger area at least if only the pattern near the boresight direction is required.
It is shown in reference that eqs.
(32) and (36) slightly modified apply to broadbeam horn antennas as well as electrically large aperture antennas.
For these broadbeam antennas the limiting value of n can be significant.
ilk II the errors in gain function, sidelobe level, polarization rati, and beamwidth are derived in terms of n(r) and the far-field pat-tern.) In order to simplify the theoretical analysis, the errors will be evaluated as if the probe were a perfect electric dipole.
The justification for choosing a perfect dipole in the analysis is similar to that stated in Section III. A.
The dipole measures the electric field components at a point.
All physically realizable probes respond to a weighted average of the fields near the probe.
Thus any small error in position would be expected to change the output of a perfect dipole by as much or more than any other probe.
As for instrumentation errors, they remain essentially independent of the particular measurement probe.
Also, as in the previous sec-tions, any uncertainty in probe receiving characteristics (S^-, ) will be ignored for this part of the error analysis.
Under the above conditions the far-field error in AE may be found with the help of eq.
(10): A E .(F) = - ik C0S6 e ik(r " d C0SG) / AE t (P,d)e~ lFR ' P dP.
(37) Z 2ttt A' r Y -> oo AE (P,d) is the difference between the actual electric field at the point (P,d) and the measured output of the hypothetical dipole probe (for two orientations in general) at the point (P,d).
For errors near rn O -y" the z-axis (0 < 2A/L , see footnote 5) eq .
(37) becomes iic iicrr <n --"i^R-P AE(r) = -i£-e 1Kir " aj J AE (P,d) e r dP, (38) 2i\r A r - °° o where A designates that part of the scan area over which the major variations in phase are relatively small.
(For the position and instrumentation error analysis the scan plane is assumed to lie nearly parallel to the near-field planes of "uniform" phase, i.e., perpendicular to the center of the main beam of sum patterns or to the null axis of difference patterns.) For scan planes in the very near-field, A is approximately equal to A cosG , i.e., the projected area of the antenna aperture.
(Only electrically large aperture 31 antennas are being considered.) 6 is the angle between the perpen-O /\ dicular to the aperture and the perpendicular (e ) to the scan plane.
0=0 for beams which are not steered off-axis electronically.
The scan area outside A can be neglected for this part of the o error analysis because the rapidly changing phase in the region out-side A contributes little when integrated to find the far-field and the errors in the far-field near the z-axis.
It follows from foot-note 5, or more rigorously from an asymptotic analysis like the one performed in Appendix A for a circular aperture of uniform distribu-tion, that eq.
(38) can be used as an upper bound expression for AE in the region given approximately by < 2A/L where L is the maximum breadth of the partial scan area A .
This region is large enough to include the first sidelobe maximum of many electrically large aperture radiators.
For example, a circular aperture of uniform distribution and radius a -L /2 has its first sidelobe maximum at o an angle -1.7X/L radians.
Physically this 2A/Lmax condition says that for most electrically large aperture antennas, the part of the near-field (A ) over which the phase is fairly uniform strongly influences the far-field within the angle 2A/Lmax .
Beyond this angle the edge diffracted fields dominate the far-fields to a greater and greater extent until in the far sidelobes the fields are determined essentially by the edge dif-fracted fields alone.
To find either the position or instrumentation errors, the integral -i^R-P T = J AE.(P,d) e r dP (39) A z o in eq.
(38) must be evaluated.
Of course, the integration can be performed only after AE (P,d) is found.
In Section 1 below, AE t , T , and thus n (r ) are evaluated for position errors, and in Section 2 for instrumentation errors.
The approach that is taken is quite straightforward.
For posi-tion errors AE is expanded in a Taylor series about (P,d) assuming the deviation in the position of the scanner is small compared to a wavelength.
The Taylor series and the integral (39) into which it 32 is substituted divides naturally into a longitudinal or z-position part and a transverse or xy-position part.
The upper-bound for each integral and thus for the z-and xy-position errors are then determined as a function of the measured near-field data and the computed far-field pattern.
For instrumentation errors, AFT.
is expressed in terms of the amplitude and phase errors introduced by the nonlinearities in the receivers which measure these quantities as the probe traverses the scan area.
Mathematically, the integrals involving the receiver phase errors are handled in the same way as z-position errors, and thus the final upper-bound expressions have the same form.
The integrals involving the amplitude errors have their upper-bound determined by characterizing the receiver nonlinearity in measuring amplitude in dB of error per dB down from the maximum amplitude of the probe output on the scan area.
1.
Position Errors Consider the scanner which moves the probe throughout the near-field scan area.
Typically the scanner covers the area by travers-ing a grid or raster of lines while the probe output is recorded at given points along each line.
Ideally, all the scan lines lie per-fectly straight and parallel in a single plane, and the position of the data points along each line is recorded exactly.
In reality, of course, none of these idealizations hold, basically because the scan lines will never be perfectly straight and the data must always be recorded over an interval rather than at a point.
Regardless of the reason for the position errors, they all effect the near-field data simply by positioning the probe at points (P+AP, d+Az) rather than (P,d) .
In other words, the difference AE (P,d) in eq.
(39) can be written for position errors as AE t (P,d) = E t (P+AP,d+Az) -E t (P,d), (40) where, as usual, E (P~,z) is the electric field at the point (P,z) in the near-field.
In general, AP~ and Az, which will be referred to as displacement errors, are functions of the transverse position P.
33 Before continuing with the analysis it should be pointed out that displacement errors caused by a small initial translation of the entire scanner with respect to the test antenna will not cause a change in the computed far-field amplitude, as eqs .
(6) and (8) indicate.
Also, an initial rotation of the entire scanner through a small angle will have no effect on the far-field pattern other than to rotate the entire pattern through that same angle.
Since the displacement errors must be much smaller in magnitude than a wavelength, the right hand side of eq .
(40) can be expanded in a three dimensional Taylor series.
By letting Ar = AP+Aze and keeping only the first two terms in the series, eq.
(40) becomes AE t (P,d) = Ar-VE t (P,d) + j Ar • [Ar «VVE (P,d) ] (41) where VVE^ = (WE )e + (WE )e t v x^ x v y y and VE\ = (VE )e + (VE )e .
t v x J x y y Substitution of eq.
(41) into eq.
(39) yields a useful expression for I , 1 -i-R-P = / Ar- [VE. +yAr-WE.
] e r dP.
A z L z o (42) Again it is emphasized that Ar is, in general, a function of P, the transverse coordinates over which eq .
(42) is integrated.
Also, as we shall see shortly, it is necessary to retain the second term in the integrand of eq.
(42) when on-axis errors for sum patterns are considered.
As a check on eq.
(42) let R = (on-axis) and Ar be constant over A so that Ar can be taken outside the integral.
The o terms containing the transverse part of the gradient operator con-vert to line integrals around the boundary of A .
These line inte-grals must equal zero because in effect eq.
(42) assumes negligible fields outside A .
Only the "z" derivatives, which can also be taken outside the integral, are left, and eq.
(42) may be written 34 ^B Az |-+ Az 2 |-r 8z 3z / E dP.
A z o With the aid of eq.
(10) , eq.
(43) converts to (43) I = -(Az+ikAz 2 )2^re lk ( r ~ d ) E^ (r) o J t K J X" -> oo (44) Equations (38), (39), (42) and (44) combine to show that E(r)+AE(r) T-X30 1 + C^Az) 8 E(r) f -»-00 i.e., the error in the far field amplitude calculated from eq.
(42) I A -p I is of higher order than (-U-<-) 2 , i.e., it is negligible when Ar is constant --a result which must hold if eq.
(42) is a valid expres-sion for I , because, as mentioned above, a translation of the entire scanner has a negligible effect on the far-field pattern.
Equation (42) also checks in a similar way for R f 0, but the proof is more involved.
In order to evaluate T of eq.
(42) exactly, both Ar and ET (F) in the near- field would have to be known.
Fortunately, an upper bound approximation can be found for T without detailed information about Ar or E (r) .
Consider one component, say E , of the integrand of eq .
(42).
For an electrically large aperture antenna, we can express E within .A.
the area A as o E (P,z) (1+AA (P,z))E (P) e i(kz + cJ)ox +Act> x (P,z)) (45) where d> is a real arbitrary phase constant, and AA , Ad> and E r ox J r x' r x ox are real functions of the indicated near-field coordinates.
The functions AA (P,z) and A<J) (P) typically oscillate over a distance equal to or greater than a wavelength, each with a magnitude usually much less than one.
E (P) is the smoothly tapered amplitude func-tion.
In other words, eq.
(45) simply states that the near-field across the area A of an electrically large aperture antenna has a phase equal essentially to kz and a smoothly tapered amplitude except for small variations which oscillate over distances equal to or 35 greater than a wavelength.
(Recall that for the position and instru-mentation error analysis we are assuming that the perpendicular to the scan area is approximately aligned with the boresight direction of the antenna.) In Appendix A eq.
(45) is verified for a circular aperture of radius "a" and uniform aperture distribution (E = E ) .
In that 1 ^—r— ox case AA and A<|> are on the order of — /A/a or less.
Rusch and Potter report similar results for the circular aperture .
Equa-tion (45) has been verified experimentally by the substantial near-field scan and extrapolation data taken at the NBS .
The measured amplitude and phase on a near- field plane of a typical microwave antenna are plotted in figures 9a,b,d.
This particular scan was taken 25 cm in front of the circular "constrained lens array" (see figure 9c) of radius 40 cm operating at 9.2 GHz (A = 3.26 cm).
This antenna is the same one described in the paragraph following eq.
(34) and whose centerline amplitude envelope is plotted in figure 7a.
When eq.
(45) is substituted into the integrand of eq .
(42) and all terms higher than either second order in |Ar|/A or first order in |Ar|//A~ are discarded, the following expression for the x component of I remains I -/ ox [Ar-V|E x 2ttAz X ox + i '2ttAz' A I j .
i.
-i-R-P r where E i4>.
dP, (46) x E e x or from eq .
(45) |E I = fl + AA )E 1 X ' v x ; ox d> = kd + + A .
yx Y ox r x It has been assumed that the transverse (P) and longitudinal (z) displacement errors are of the same order of magnitude.
Eq.
(46) can i i(kd + (j)ox ) be simplified further by writing e as e (cosAcfj^+i sinA<|> x ) , to put I in the form r ox 36 i(kd+ + i sin A<|> )e r dP .
(47) (6 = 2-ttAz/A) First consider on-axis errors for antennas operating in a sum mode, that is ^--<< 1 or 6 less than about A/ (10 Lmax ) .
Then the exponential in eq.
(47) can be approximated by unity.
The arbitrary phase constant i> can be chosen so that Ad> varies about zero r ox Tx throughout the area A .
Furthermore, since the variations in Ad> te o T x are small, oscillate many times across the area A , and remain com-pletely independent of the variations in the displacement errors Ar of the scanner, to a high probability, c|> can be chosen such that the integration in eq.
(47) which is multiplied by sinAcj) can be neglected compared to the maximum possible value of the cosAcf> inte-gration.
In addition, since E cosAc}> remains positive throughout A for a sum pattern, the reference plane (z=d) for 6 can be chosen to make J 6E cosAcJ) dP = 0.
(48) ; ox Yx • J o Thus, under the above conditions the iS term of eq .
(47) is elimi-nated entirely and eq.
(47) simplifies to i(kd + ) n I = e ox j ( Ap . v E _ i 6 2 E ^ cos A(J) dp> ox i v t ox 2 ox ; x o (A = A +e |-) • t Z dZ J (49) E in the first term of eq .
(49) has replaced |E I =(1+AA ) E of ox n v J r ' x 1 v x^ ox eq.
(47) because the many oscillations of AA would also (to a high probability) eliminate upon integration all but higher order con-tributions from the V(AA ) term.
v x' It is interesting that eq.
(48) expresses the same condition chosen by Ruze in his well-known work on "antenna tolerance theory," and under the above assumptions the z-position errors always lower the on-axis gain value of the sum pattern.
Of course, in his work 6 represented a small "arbitrary 37 phase error or aberration" (what we call A ) rather than the dis-placement errors of a near- field scanner.
It would be desirable that the reference planes of Ax and Ay (AP = Axe +Aye ) be chosen such that / AP«V E cos A<|> dP" became zero also.
Unfortunately, such a A Z 0X X 9E choice is impossible (in general) because neither derivative, ox 8E 9x or ox , stay the same sign throughout A .
From eqs .
(38) and (39) it is seen that the fractional error n of eq.
(1) can be written as n(r) Ar |E(r) ox + |I | 2 1 oy X->oo (50) The amplitude of I is found from eq .
(49) to be |I | = |/ (AP-V.E - \ 6 2 E )cos Ac}) dP 1 OX ' A t OX 2 OX ; X o < AP X6 / |V^E I dP + \ 6 2 max i ' t ox ' 2 max i ox A / E_ dP, Ao (51) where AP and max 2tt max are the maximum magnitudes of the transverse (P) and longitudinal (z) displacement errors, respectively, intro-duced by the near- field scanner.
The last integral in eq.
(51) can be related to the maximum far-field |E I , i.e.
from eq .
(10) or (30b) xo r--°o 7 / E dP = ArlE I ^ ox ' xo ' r^-°° o (52) The remaining integral, J A V^E dP, t ox ' ' (53) is a bit more troublesome.
However, it can also be estimated by expressing the integral in polar coordinates (p,) as follows: / |V.E I dP = f A t °X A o o 3E ox 3P -, 8E 1 ox ^ 2 P 94) pdpdcj) (54) 38 If we assume that the amplitude of the aperture distribution tapers with much greater slope in the radial direction (p) than in the azi-muthal direction () , the second term under the radical sign of eq .
(54) can be neglected, leaving 9E _ f ox Ao 3P pdpdcj) .
(55a) 9E The negative sign is present in eq .
(55a) because ox 3p is negative for amplitude distributions which taper toward the boundary of A .
(We are assuming p = at the maximum of E .) Integration of eq.
O -A.
(55a) by parts with respect to p yields 8E •J A ox 3p where E and L xo o o max pdpdcj) = J E dpdc}> A ox o tt_ r T max 2-C 1j xo o (55b) denote the maximum amplitude of E and the maxi-ox mum breadth of the partial scan area A , respectively.
With the aid of eq.
(55b), eq .
(54) becomes / |V F | dP < £ Lmax E { ' t ox 1 — 2 o xo o (56) Since the area A is in the very near- field of the aperture antenna, the maximum amplitude E on A is approximately equal to the maxi-mum amplitude on the aperture itself.
maximum far-field by (see eqs .
(30b) and (31)) Thus E is related to the xo aAr Exo ' r-xo A and eq.
(56) becomes (57) J |V^E I dP < { ' t ox 1 — o , T max i ~ i TTaArL E o ' xo ' r-»°° 2 A (58) Substitution of eqs.
(52) and (58) into eq.
(51) yields r . „ . max TraAP n L I I < ArlE I OX ' — ' xo ' ]f->OQ TTaAP 1/ max o 2 A + i 6 2 2 max (59) 39 A corresponding inequality holds for ll I with IE re-r ?
n 7 ' oy ' ' yo ' r->°° placing |E | ^oo .
At first sight, one may raise the objection that the reference plane for <5 cannot necessarily be chosen so that both eq.
(48) and the corresponding equation with E are satisfied simultaneously.
It should be noted, however, that the two equations need not be satisfied simultaneously.
It is only necessary that one reference plane can be found that allows eq .
(48) to hold, and that a second reference plane can be found that allows the corresponding equation with E to hold.
If the two reference planes are a slight distance apart, the relative phase of the x and y components of the far-electric-field will be in slight error but not the Hermitian amplitude of the far-electric-field, i.e., the far-field pattern will remain unchanged even though the far-field polarization ratio will be shifted slightly.
Eq.
(59) and the corresponding equation for |l | combine with eq.
(50) to give an expression for the maximum position error n in the far-field: n(r) An T max TraAP L -.
max o 1 x-2 + -7T 2 A 2 maX A J 10Lmax o (sum patterns) (60a) Equation (60a) was derived assuming the antenna was operating in a sum pattern and does not apply to difference patterns since eq.
(48) may not be satisfiable for difference patterns.
However, in Appendix B it is shown that an error expression similar to eq .
(60a) may be derived near the boresight direction (null axis) of difference patterns as well.
Specifically, n(r) An T max TraAP L max o 2 A + 4AF6 max g(r), 6 < J 10Lmax where AF and g(r) are defined below after eq.
(61) Finally for the angular region > A/10 Lmax o (difference pattern) (60b) but still less than 2A/L (see eqs .
(37) and (38)), we can derive an upper bound expression similar to eq.
(60a).
Specifically, the magnitude of I Qx in eq.
(47) may be written 40 J<| |Ar-V|E || dP + \ j |6E | dP.
A X A 0X o o ox The term second order in <5 has been ignored since now the first order term in 6 is the major contributor to the z-displacement error.
When we carry the analysis through in a way similar to that described between eqs .
(51) to (59), the following expression for the frac-tional error for both sum and difference patterns results : n(r) < An T max ^ TTO.AP L max o o g(r) 2 A » max 1QLmax 2A max o (60c) Although eq.
(60c) was derived for 9 < 2A/L , it remains valid as an upper-bound expression all the way to = tt/2.
This result follows from the fact, which will not be proven here, that the con-tribution from the integration in eq.
(37) outside the area A is negligible compared to the upper-bound contribution obtained from inside A and expressed by eq.
(60c).
(It is interesting to note that the maximum possible errors in even the far sidelobes are determined by the displacement errors across A , whereas the field itself in the far sidelobe region is determined by the near-field outside A .) o ' Thus, if we combine eqs.
(60a), (60b), and (60c); insert the approximations, L A 2ttAPmax max o Imax A (£max ) 2 into eqs.
(60); and let max A far-field hemisphere emerges an upper-bound expression for r\ valid over the entire n(r) < A + n max max l z 2£ 6 2 max g(r) 2 ^z =( 8AF 5 max max (sum patterns) (difference patterns) (sum and difference patterns) 10£max 10£max <e< (61) 41 where X = wavelength „max = maximum width of the antenna aperture A™a Y = 2lTAP rr1 o V / A » where AP is the maximum amplitude of the trans-IlLdA IlLdA IIldA verse displacement errors within the partial scan area A .
o (A is that part of the scan area over which the phase is fairly uniform.
For near-field scans parallel to the aper-ture A -A, the aperture area.) 6 m o V = Z^^z /\ , where Az m „ v is the maximum amplitude of the IIldA IlLdA ITldX longitudinal displacement errors within the partial scan area A .
o AF = fractional difference between the amplitude of the two main far-field lobes of the difference pattern (see Appendix B) a = a "taper" factor --equal to a minimum of 1.0 (for apertures of uniform amplitude and phase) and less than 5 for most tapered distributions found in practice.
(See eq .
(31) for the precise definition of a; for a difference pattern one should still use the taper factor of the constituent sum patterns .
) g(r) = ratio of the amplitude of the maximum far-electric-field to the far- electric- field at the given direction r, i.e., the inverse of the normalized far-field pattern.
(g(r) = 1 for the center of the main beam, or beams if a difference pattern.) (If desired, the errors in the gain function, sidelobe level, polari-zation ratio, and beamwidth may be calculated from eqs .
(2) -(5) once n and the far-field pattern are known.) The expression (61) represents an upper bound to the far-field error n caused by inaccuracies in the position of the near-field scanner.
It applies to both sum and difference patterns in the entire forward hemisphere of all electrically large aperture antennas.
Equation (61) was derived under the assumption that the scan plane is parallel or nearly parallel to the near-field planes of "uniform" phase, i.e., the plane perpendicular to the electrical boresight direction.
In addition, eq.
(61) holds for arbitrary (random as well as systematic) errors in the positioning of the scanner, since only the 42 maximum magnitude of displacement errors A and 6 are required to r max max l to evaluate eq.
(61).
However, it proves useful to derive an ex-pression similar to eq.
(61) which separates the effects of the systematic and random errors in position.
To do this and also clarify what is meant by systematic and random errors, consider the motion of the scanner as it takes mea-surements along lines in the near-field plane.
As the scanner moves along each line it will deviate from a perfectly straight -line by a gently varying curve that will contain at most a few oscillations from one end of the scan line to the other.
These curves, which may also change gently from scan line to scan line, represent the sys-tematic errors in position of the scanner.
For example, a slight warp or deformation of the scanner frame would create a systematic error .
Superimposed upon the systematic deviations from the straight-line would be position errors which changed randomly (within limits) from measurement point to measurement point.
These random errors in position have zero or nearly zero mean and could result, for example, from vibrations of the entire scanner or from a slight play in the drive mechanisms.
The maximum magnitude of the random errors are often smaller than that of the systematic errors.
However, it is possible that the scanner is aligned so precisely, that essentially all but the random errors are eliminated.
Return to eq.
(47) and separate AP and 6 into systematic and random displacement errors, i.e., AP = AP S + AP rn (63a) 5 = 5 S + 6 rn .
( 63b ) —rn All the integrals in eq.
(47) which contain linear terms in AP or tji rn rn <5 can be dropped.
Since AP "" and 6 change randomly from measure-ment point to measurement point (over distances less than A/2) , the integrals actually summations --see footnote 3a) containing these linear terms will be extremely small compared to the largest possible errors produced by the remaining integrals (summations).
Thus, sub-stituting eqs .
(63) into eq.
(47) and proceeding as we did before with eq.
(47), yields the desired expression for n separated into 43 systematic and random errors, n(r) aX . s 21max max s ,rn g(r) 2 (6 ) 2 +(6 in ) 2 (sum patterns) max' v max' r ' i S ' rn =< 8AF 6 : max max (difference patterns) (sum and difference patterns) 6 < X 1(Umax A — <e< 10JImax (64) All symbols have the same definitions found after eq.
(61).
The superscripts "s" and "rn" refer to "systematic" and "random" errors respectively Note from eq.
(64) that to the given order of approximation the longitudinal random errors (6 ) , but not the transverse random errors (A ), cause an error in the on-axis far-field of sum pat-r n terns (provided, of course, that A is of the same order of magni-rn tude or less than 6 ).
In addition, eqs .
(64) and (61) above show that the maximum possible transverse (xy) position errors do not depend upon wavelength for a given g(r) since A behaves as 1/X.
max Also note that in expressions (64) and (61) above, the z-position ma v error is not continuous across the angle 6 = X/lOi .
There lies no contradiction in this fact since the expressions remain valid as inequalities.
The jump between the two regions merely indicates that the upper-bound was determined by a different method in each region.
Obviously expressions (61) and (64) do not represent a rn o -y-least upper- bound throughout the region 9 > X/KU .
This region will be discussed further at the end of the section.
One can get an idea of the magnitude of the position errors by plotting the on-axis gain from eqs.
(64) and (2c) for the fixed-beam constrained lens array described in the paragraph following eq.
(34) and shown in figure 9.
In that case, A/£max = .04 g(r) = 1, 44 and with a equal to 3, eq.
(64) combines with eq.
(2c) to give s nf < ± 8.7 A" + J-(^ S ) max 2 K max ; 2^ max ; (65) I£ we assume 6 written rn max = and let 6=6' o max = A" max eq.
(65) may be dB < ±i 7 6 o (.03 + f « ). (66) dB particular antenna, 6 must be less than about r ' o The solid curve in figure 10 depicts the maximum value of r\ r in u eq.
(66) versus 6 , i.e.
when the random errors are negligible.
The o dB s s dashed line represents the maximum error r\ n when the A = 6 = r G max max rn and 6=6 , i.e., when the systematic errors are negligible, o max' ' J & & To insure that the on-axis gain is less than .01 dB for this 027.
That is, each component of systematic displacement error of the scanner must be less than .027 A/2tt, or less than about .004A.
For this constrained lens antenna A = 3.26 cm, so the errors in each component of dis-placement must be less than about .14 mm for better than .01 dB accuracy in the on-axis far-field.
For random errors only (the dashed line) the z -displacement errors must be less than about .26 mm for .01 dB accuracy in the on-axis far-field for this particular wavelength.
As a matter of fact, when the systematic errors are negligible, eq.
(64) shows that dB the "random" error in gain x\ n for the sum-pattern main-beam of an & G,rn r arbitrary (electrically large aperture) antenna can be written simply as dB < + 8^ f6 rn . 2 G,rn — 2 ^ max-' (6 rn = 2TiAz rn /A) • max max } (67) dB rn rn For x\ n < ± .01, 6 must be less than about .05 radians or Az G,rn — ' max max less than .008 of a wavelength (3 degrees).
That is, the random errors in z-position should be no greater than about ± .01A to insure a .01 dB accuracy in the gain of the main beam.
Equation (67) re-veals that the random error in the on-axis far-field gain of sum patterns increases as the square of the random error in the Actually, as mentioned above, the z-position or phase errors always lower (under the stated assumptions) the on-axis gain of sum patterns, but the + sign will be included throughout to keep the same form to all the error expressions.
45 z-position of the near-field scanner.
For example, a position error of ± .025A -10° (@ 3.125 x .008A) leads to a maximum r^ B of about 2 G,rn ± . 1 dB (@ (3.125) x .01) for the center of the main beam.
To date, little experimental data are available with which to compare the results of eqs .
(61), (64) or (67).
However, a compari-son can be made between eq .
(67) and the computations performed by Rodgrigue, Joy, and Burns .
They introduced errors into a hypothetical near-field distribution in order to compute the effects of the errors on the far-field.
The results of their computations for the effect of random phase errors (or equivalently z-position errors) on on-axis gain are plotted in figure 3-14 or A-21 of , and are reproduced here by the dashed line in figure 11 below.
The solid line is n^ plotted from eq .
(67).
One can see from figure G,rn r n v J & 11 that the two curves are in close agreement.
The solid line lies slightly above the computed dashed line as indeed it should if eq (67) represents an upper bound.
Figure 12, which will be explained in greater detail below under instrumentation errors, shows a comparison between the effects on the on-axis gain of a sum pattern from systematic (quadratic in this case) phase errors introduced into actual near-field data by Newell [17,21] and the corresponding upper-bound results calculated from eq.
(64).
Again agreement is close, with the upper-bound curve lying just above the actual curve for small deviations in phase Next, let's calculate the effect of systematic z-displacement errors on the depth of the null for an antenna operaing in a dif-ference pattern.
From eq .
(64) tv B < ± 35 AF 6 s g(r) , 'G,s — max & • J ' where g ( r ) is no longer equal to unity because r is in the direction of the boresight null.
Typically, AF is about .01 and the depth of the null is 25 dB down from the main beams of the difference pat-tern.
Thus g(7) ~-10 5/4 -18, and n„ < ±b . 3 o l G,s — max 46 rn for the null depth.
For example, if Az = .01 A.
(which would cor-max respond to an accuracy of about .27 dB for the main beams) eq.
(68) insures that the null depth would be accurate to within .40 dB .
This surprisingly high insensitivity of the null depth to z-displacement errors (or, equivalently , phase errors) has been ob-served by Newell upon introducing phase errors into the measured near-field data of a number of antennas operating in the difference mode.
To understand the reasons for this high accuracy in null depth, one must refer to the derivation in Appendix B.
There appears to be no simple way to explain this result heuristically .
It is also found in Appendix B that the maximum shift 6 , . r .
ri^ shift in the direction of the far-field null of a difference pattern caused by z-position errors is given by the simple expression which is not a function of wavelength: 4Az'max 2X shift Imax tt£max max radians .
(68) In general, the shift in the null caused by all other sources of error are negligible compared to this shift caused by the z-position error or, equivalently, the phase errors.
Table 2 lists the upper-bound position errors in a number of far-field parameters for a typical X-band and K-band antenna.
The values in the table were calculated from expressions (61) and (68) .
As eqs .
(61) and (64) show, the maximum possible transverse (xy) position errors do not depend upon frequency for a given g(r).
Table 2 also shows quite dramatically that the z-position or phase errors everywhere except in the boresight direction can be extremely large compared to all other representative sources of error --com-pare Tables 1-4.
(This is also true of the shift in the difference pattern null.) Especially note that the error in a -25 dB sidelobe can be several dB for phase errors (2ttAz /A) of just a few degrees max Whether or not these maximum possible far-field errors are actually experienced in practice depend strongly upon the shape of the near-field z-position or phase error throughout the scan area.
For example, we shall find in the next section that receiver phase dis-tortion usually has a functional dependence which introduces 47 negligible errors into the sidelobe fields.
It is important to know exactly what effect various distributions of near-field phase errors have on the far-field in order to avoid experimentally, if possible, the distributions which produce large far-field errors in the directions of interest.
Such a detailed study of the depen-dence of the far-field errors on the functional form of the near-field z-position or phase errors will not be included as part of this report but will be contained in a forthcoming report by Newell 2.
Instrumentation Errors The amplitude and phase of the probe output are measured at discrete points as the probe moves back and forth across the near-field scan area.
The receivers are capable of sampling and recording the amplitude and phase to within a certain accuracy only.
The errors in the near-field data, caused by the inaccuracies in the receivers or instrumentation used to measure the probe output, pro-duce errors in the computed far-field.
This section estimates the far-field errors under given limits of accuracy of the instrumenta-tion which measures the amplitude and phase of the probe output.
It is emphasized that the errors produced by the imperfect positioning of the scanner were determined in the previous section and are not considered as part of the instrumentation error analysis of this section.
Also the instrumentation errors associated with convert-ing analogue to digital information is assumed negligible.
Under the conditions explained in Section III.B, the far-field errors can be found in the region e < x/(icumax ) by evaluating the integral (39) I = / AE.(P,d) dP.
° A z o Here, AE (P,d) represents the difference between the measured (E t ) and "actual" (E ) output of the probe at the point (P,d), i.e.
AE (P,d) = E™eas (P,d) -E t (P,d).
(69) 48 If we look at just the x -component first, the integral I and eq.
(69) combine to give I ox = / (E^eaS -Ex ) dP.
(70) o By writing and Ex = |E |-e X (71a) i(cf> +ACJ) 1 ) Emeas = M E i + AA ± ]e x x (71b) X L ' X ' X J ' K J where AA and Acf> are the errors in amplitude and phase respectively -A.
-A.
introduced by the measuring instrumentation, eq.
(70) becomes I« v = / [Cl'E l+AA^e X -|E |]e x dP.
(72) ox i L v ' x ' x' ' X ' J • J o I I In general, both AA and Ac)) are functions of the transverse coordi-.A.
-A.
-p ' nates ¥.
For small errors A<J) << 1, so that yC x « 1 + iAcf) 1 -x— Y x 2 and I ox = / [AA + i(|E x | + AA^)A^ - -^(A^Me X dP.
(73) o All terms higher than second order have been neglected in eq .
(73).
icj> x If a sum pattern is assumed and the exponential e is written as in eq.
(47) , i x i(kd+<|> ) e = e (cos A + i sinA(j) ), -A.
A with the arbitrary phase constant chosen as in eq.
(47), then to OX a high probability, the integration in eq.
(73) which is multiplied by the oscillating quantity sinA can be neglected compared to the -A.
49 maximum possible value of the cosA x = 0, (74) o which again expresses essentially the same condition as that chosen by Ruze in his work on antenna tolerance theory.
Under the above conditions, the imaginary term within the brackets of eq.
(73) is eliminated and the magnitude of I reduces to to ox T l E I T |I 0X I = 1/ [AAx - ^^(A4> x ) 2 ]cosAcJ) x dP| (75) 1{ Kl d¥+ iKmax^ 2 I l E xl d¥ -o ' o Acf> ov is the maximum value (in absolute value) of the instrumentation xmax -. n phase error within the partial scan area A .
Since it can be shown that (see eq .
(10), (30b) or (52)) / |E I dP = ArlE I (76) o where |E I is the magnitude of the maximum x-component of far-xo r^"°° field, I I I may be rewritten ' ' ox ' ; Because the errors in measuring phase are usually greatest at points in the scan area where the amplitude is least, it is desirable to choose the partial scan area A as small as possible when estimating Acj) 1 .
The far-field within <2A/£max is hardly effected by the T xmax ' ' J near-field outside that part of the scan area where the amplitude of electric field is equal to the edge taper down from the maximum amplitude.
Thus for the sake of designating maximum errors in phase measurement on a near-field scan plane, the area A need be chosen no larger than this effective "edge taper" scan area.
50 I I < J I AA 1 t dP + tCAcf) 1 ) 2 Ar|E | ox ' — i ' x 1 2^ Y xmax ; ' xo o p->oo (77) It is usually possible to express the errors in amplitude in dB per dB change of amplitude.
That is, the receivers are assumed to read the correct (zero error) amplitude of the probe output at its maximum value point on the area A .
At all other points the ampli-tude lies a certain number of dB down from its maximum value.
Typically the dB errors in measuring the amplitude are linearly related to this number of dB down from the maximum amplitude, and thus the errors can be expressed in dB per dB down.
Even if the actual amplitude error curve is not linear, for the sake of the upper bound expression (77), it can be replaced by a straight line (linear curve) which is equal to or greater than the actual error curve .
Specifically, if N, R designates the amplitude error in the number of dB per dB down, and A JT) the amplitude in dB down from the j Old maximum amplitude, AA can be expressed as (for small errors) AA I "x x NdB W 8 - 7 (78) By definition AdB 20 log xo xo IE X (79) where E denotes the maximum of E on the scan area xo ' x ' tion of eq.
(79) into eq.
(78) and the result into eq.
Substitu-(77) yields ox < N I dB / (E -IE A xo ) dP + =-(Acj) ) 2 Ar|E I J 2 V Y xmax ; ' xo ' r-^°° (80a) With the help of eqs .
(57) and (76), eq .
(80a) becomes 'aA I < ox ' — NdB A -1 + ^(Acf) 1 ) 2 ^ Txmax ; Ar E xo ' r--°° (80b) By combining eq.
(80b) and the corresponding equation for |l | with o eq.
(50) , assuming E _.-£, and approximating —r- by 1, the upper-bound expression for the fractional far-field error n caused by the 51 instrumentation is obtained 11 n(r) < [Nj B (a-l) ^(A<ax ) 2 ] 9 <^max (sum patterns) .
(81) Equation (81), which is analagous to the position error equation max (60a), holds only for sum patterns within an angle A/(10£ ) of the boresight direction.
For difference patterns as well, and for > A/(10£ ), the following upper bound expression analagous to the position error equation (61) applies: n(r) < [2N^ B (3 : ) + n] ^^ (A(J> ) where dB 8AF A(J) A<(> max r = < max (a-D 2/g(r) (a-l)/2 (sum patterns) (difference patterns) (sum and difference patterns) (sum patterns) e (difference patterns) A 10£max 10£ A max <e< 10£max A (sum and difference patterns) aALmax 101 10A max <e< 2A max 3A <e< 1 V.
Imax (82) = wavelength.
= the maximum instrumentation errors involved in measuring the amplitude of the probe output --N,„ is expressed in error per dB amplitude down from the maximum amplitude on the scan area.
(For the present purposes, the amplitude error is designated as zero at the maximum amplitude.) Hwhen the maximum value of the probe output on the scan area for the x-orientation is very different from that of the y-orientation (E / E ) , it can be shown that eq .
(81) remains valid as an upper yo xo n v bound for the errors in magnitude of the far-field.
But for errors T in polarization ratio an extra term, N , R | x | /8 . 7, must be added to eq.
(81), where xp is the difference between the maximum probe out-puts measured in dB for the two orientations.
52 Ad) = the maximum instrumentation errors (expressed in radians) Ymax r J involved in measuring the phase of the probe output on the effective scan area A (see footnotes 10 and 13) .
o J -fractional difference between the amplitude of the two main far-field lobes of the difference pattern (see Appendix B) .
'taper" factor --equal to a minimum of 1.0 (for apertures AF a = a g(r) max max of uniform amplitude and phase) and less than 5 for most tapered distributions found in practice.
(See eq .
(31) for the precise definition of a; for a difference pattern one should still use the taper factor of the constituent sum patterns .
) = ratio of the amplitude of the maximum far-electric-field to the far- electric-field at the given direction r, i.e., the inverse of the normalized far-field pattern.
(g(r) = 1 for the center of the main beam, or beams if a difference pattern.) = maximum width of the antenna aperture.
= maximum width of scan area.
= area of antenna aperture.
The derivation of r] is identical to that done for z-position errors z I X 2 X in Section III.B.l.
The derivation of 3 for <0< is io£max £max ax accomplished by the same procedure used above for 9 < X/(10£m ).
In the far sidelobe region, 9 > 10A/&max the far-field errors be-come approximately equal to the corresponding errors in the near-field amplitude, and after using eqs.
(30b), (31), and (79) 3 in this region can be written as shown in eq.
(82) .
Between 9 equal to may rn o "Y" 2X/£ and lOX/il the value of 3 can be estimated by connecting rnn v rn o "V" a straight line from its value at 2A/£ to its value at 10A/JI 3 near the boresight direction < X/(10£ ) of difference patterns is derived in Appendix B.
(Note that the error factor 3 is generally much smaller near the boresight direction or null axis of difference patterns than near the center of the main beam of sum patterns.
This result occurs, as Appendix B shows, because the instrumentation dis-torts the amplitude on the "positive" and "negative" sides of a dif-ference pattern by approximately the same amount.) Appendix B also shows that we can write an upper-bound expression for the null shift of difference patterns caused by the instrumentation errors: 53 2Ad) X Ymax , .
u . £ .
< radians.
shift — n max (83) Equation (83) is identical to eq .
(68) with the instrumentation phase error replacing the z-position error.
Equation (82) represents an upper bound to the far-field errors produced by the instrumentation which measures the amplitude and phase of the probe output.
It applies to either sum or difference patterns in the forward hemisphere of all electrically large aper-ture antennas.
As with eq.
(61), eq .
(82) was derived for scan planes which are parallel or nearly parallel to the plane perpendicular to the electrical boresight direction.
(If desired, the errors in the gain function, sidelobe level, polarization ratio, and beamwidth may be calculated from eqs .
(2) -(5) once n and the far-field pattern are known .
) A comparison of eq.
(82) with eq .
(61) shows that the instrumen-tation phase error Ad) in eq .
(82) has taken the place of 6 eq in max l A max (61).
This result acts as a check on eqs.
(61) and (82) because 6 simply represents a phase error caused by a z-displacement max -i -j -p error in the position of the scanner.
Also, Ad) can be separated r max r into a random and systematic part to get a result analagous to eq (64).
However, the random phase errors introduced by the instrumen-tation are usually much smaller than the systematic phase errors, and thus can usually be neglected.
Figure 11 reveals that the phase (Acf> ) part of eq.
(82) for max the on-axis gain of a sum pattern is in good agreement (as an upper bound) with the computer error analysis performed by Rodrigue, Joy and Burns on a hypothetical near-field distribution with random phase errors.
Newell [17,21] has introduced phase errors which are quadratic with respect to the xy coordinates, into the actual near-field data of the 60 GHz, 46 cm (18 inch) reflector antenna whose 12 Note that the "phase error terms" in eqs.
(61) and (82) for sum patterns and < A/(10£max ) depend on the square of 8 and A6 JIlci.A.
IIld-A.
respectively.
Thus, these two squared terms should not be added directly when estimating the total error in the far-field.
Instead, 6 should be added to Acf) 1 before the square is taken.
max max 54 near-field amplitude envelope is shown in figure 7b.
The resulting JD errors in r\ r for on-axis gain computed by Newell are shown with the dashed line in figure 12.
The solid line represents the maximum phase error plotted from eqs .
(82) and (2c).
Again it appears that the expression (82) represents a useful upper bound estimate for the instrumentation phase errors in on-axis gain, especially when the measured phase is accurate to within a few degrees across the effective scan area A (as is usually the case in practice).
The accuracy with which the receiver can measure the phase of the probe output is related to the amplitude of the probe output.
Specifically, the smaller the amplitude the larger the phase errors usually become (see footnote 10).
For example, a typical receiver used at the NBS near-field range measures phase to within ± .001 radians (± .05°) at the maximum amplitude on the near-field scan plane, and ± .01 radians (.5°) at an amplitude 20 dB down from the maximum.
Thus, for an edge taper 20 dB down, AcJ) =.01, and eq .
(82) max shows that the error in the main beam of a sum pattern caused by the errors in measuring the phase of the probe output are negligible (r^ = Z^L (-01) 2 < ± .001 dB) .
The same is true for the null depth and shift of difference patterns.
Of course, the errors in sidelobe level could be affected to a greater extent by the phase errors, depending on the shape and distribution of the phase errors across the scan area.
However, since receiver phase errors usually increase monotonically with decreasing amplitude, it can be shown as a consequence that the upper-bound off-axis or sidelobe phase errors (A<j) . /2 g(r)) given in eq.
(82) represents a much larger error in far-field than would usually occur in practice.
Thus , in general , receiver phase errors have a relatively small effect over the entire far- field of sum or difference patterns .
In fact, compared to the maximum possible effect that typical z-position errors can have on the off-axis far- fields, instrumentation phase errors can be ignored completely in the off-axis region.
Even for aperture antennas with edge tapers greater than 20 dB down, it is unlikely that the near-fields outside these -20 dB points (or even the -15 dB points) have a significant effect on the maximum pos-sible far- field errors.
In other words, regardless of how large the edge taper, the effective scan area A need not extend beyond about the -15 or -20 dB points.
55 For receivers which measure phase with high accuracy, the phase part of the instrumentation error can be neglected and only the amplitude error remains in eq.
(82), i.e.
n(r) < N^Cg 1 ) g(F).
Amplitude errors for < 2\/ I do not depend directly on frequency or the size of the antenna, only on the taper factor a of the near-field beam, the receiver inaccuracy N, R , and the inverse of the normalized far-field pattern g(r).
(For null depth of a difference pattern the amplitude errors are extremely small and do not even in-volve g(r).
These results are proven in Appendix B.) Also, in the far sidelobe region, 6 > 10A/& , the instrumentation amplitude errors are relatively small, usually less than a few tenths of a dB for N, R less than a few thousands of a dB per dB .
The far-field error in the on-axis (g(r) = 1) gain for sum patterns is found from the above equation and (2c) to be n£ B < ± 8.7 N B (a-l).
(84) Note that for a=l (uniform amplitude distribution) the far-field error, caused by the instrumentation errors in measuring near-field amplitude, equals zero--as it should since the receivers measure essentially at a constant amplitude across the effective scan area.
The maximum error in on-axis gain (eq.
(84)), which is linear with respect to N, R , is plotted with the solid lines in figure 13 for different values of a.
Rodrigue et al .
have also computed linear amplitude errors for their hypothetical near-field distribu-tion, which has an a exactly equal to 3.0.
Their results (see figures 3-5 or A-5 of ) are reproduced by the dotted line in figure 13.
The errors computed by Rodrigue et al .
and the maximum errors predicted by the analytically derived expression (84) are in good agreement.
The solid line for a=3 lies above the dotted line --as it must if eq.
(84) represents a valid upper bound expression for the errors To insure an on-axis gain error less than ± .01 dB , the error in measuring the near-field amplitude should be kept less than about ± .001 dB per dB down.
Unfortunately, the accuracy of most receiver 56 systems is of the order of ± .01 dB per dB down (rather than ± .001) which, according to figure 13, can lead to errors in the on-axis gain of about ± .1 dB .
This value of far-field error can be larger than the errors from all other sources combined.
Thus, if high accuracy is desired, special effort should be devoted to designing a receiver system which can measure the amplitude of the probe output to better than ± .001 dB per dB down.
Alternatively, the amplitude calibration curve for the receivers could be determined to within ± .001 dB per dB down and the errors in amplitude compensated for by including the calibration curve as part of the computer program that deconvolutes the near-field data.
This latter correction procedure has been adopted by A.C.
Newell et al.
at the National Bureau of Standards.
The difference between the on-axis gain computed by Newell [17,21] with and without the ampli tude calibration curve is shown by the dashed line in figure 13 for the 46 cm (18 in) reflector antenna operating at 60 GHz.
(Actually, Newell found a .112 dB on-axis gain difference for an amplitude cali-bration curve that deviated by at most .02 dB per dB down over the effective scan area.
The dashed line in figure 13 assumes linearity and simply connects the origin to the point .112 dB at .02 dB per dB .
) The value of a for this antenna was estimated at 2 from the measured near- field data shown in figure 7b.
Again it is seen from figure 13 that the computed errors for this particular antenna correspond quite well with the maximum possible errors predicted by the general expression (84) for a = 2.
In brief, the computations of both Rodrigue et al .
and Newell indicate that eq.
(82) yields reasonable values for the maximum far-field errors expected from instrumentation errors in measuring the amplitude and phase of the probe output.
Table 3 lists some representative far-field amplitude errors calculated from eq.
(82) for an antenna with taper factor a equal to 3, and a receiver nonlinearity in measuring amplitude (N, R ) equal to .002 dB per dB .
Note the extremely small effect that amplitude errors have on the null depth of difference patterns.
As Appendix B shows, this small effect on the null depth is due to the fact that the receiver which measures the amplitude of the probe output distorts the opposite sides of the near-field difference pattern by approxi-mately the same amount.
57 C.
Multiple Reflections Consider a probe which scans on a plane in the near-field of a radiating test antenna.
The radiation that the probe receives can be described by an infinite series of rapidly decreasing terms, with the first term equal to the unperturbed field of the test antenna.
This unperturbed field scatters from the probe, reflects from the test antenna and other nearby objects, and returns to the probe to give the second term in the series.
The return radiation again scatters from the probe, reflects, and returns to the probe to yield the third term.
The process repeats ad infinitum.
In order to determine the far-field of the test antenna by "deconvoluting" the measured near-field data (without knowing the detailed scattering properties of the probe or test antenna) , the multiple reflections must be neglected.
That is, the second and higher order terms in the infinite series just described are assumed negligible when applying the planar near-field scanning techniques Multiple reflections can be reduced by decreasing the size of the probe antenna, by increasing the distance between the probe and test antenna, and by appropriately covering the scanning range with efficient absorber material.
These measures will not, however, eliminate the multiple reflections entirely.
It is the purpose of this section to estimate the effects of the multiple reflections on the far-field which is computed from the near-field scan data under the assumption of zero multiple reflections As we shall see shortly, it is a fairly simple matter to esti-mate maximum and minimum values expected for the far-field errors produced by the multiple reflections.
However, it is impossible to derive an accurate estimate of the far-field errors analytically without knowing the phase of the multiply reflected fields throughout the scan area.
Thus, the only reliable way to get an accurate esti-mate of the far-field errors caused by multiple reflections is through measurement.
Specifically, a number of near-field scans could be 14in principle, the effect of multiple reflections could be eliminated at microwave frequencies by the use of gated sinewaves instead of CW.
Unfortunately, the speed of electromagnetic propagation (c) is so great that the necessary gating times are too short for the present-day elec-tronics to handle.
It is possible, however, in the analagous measure-ment of electroacoustic transducers to eliminate the problem of multiple reflections by the use of gated sinewaves because the speed of sound is much smaller than that of light .
58 taken on parallel planes separated by about 1/4 wavelength and the far-field computed from each scan.
Any differences observed in the far-fields computed from the separate near-field scans would indicate the extent of the effect of multiple reflections (assuming the scan area is large enough so that changes in finite scan errors are neg-ligible) .
In this way the far-field errors can be determined straightforwardly and accurately.
Of course, the main disadvantage of this "straightforward method" of determining errors lies in the time and effort it takes to record data and compute far-fields from several near-field scan planes for every antenna that has to be mea-sured.
Because of this disadvantage it may prove worthwhile, par-ticularly when the multiple reflections are very small, to derive the following very approximate, yet general, upper and lower bound expressions for the far-field errors caused by the multiple reflec-tions.
In addition to the upper and lower limits of errors, the far-field errors will be derived for multiply reflected fields which satisfy a certain class of hypothetical near-field distributions.
Consider a probe antenna scanning on a plane in the near-field of an electrically large, aperture antenna.
Assume once again, for the sake of simplifying the mathematics, that the probe behaves as an electric dipole, i.e.
its output in one orientation is propor-tional to E and in a second orientation proportional to E .
Then x_ y the error (AE) in the computed far-electric- field can be expressed with the aid of eqs .
(37) and (12d) , AE < "A e ik(r-dcos6) j Agnr (F^ e " 1 rR " P dF< (85a) ;f->-co rn "p The superscript "mr" on AE denotes that part of the transverse near-electric-field caused by multiple reflections, and A' refers to the scan area.
The determination of (AE) requires the evaluation of the integral -i-R-P I = / AE™r (P,d)e r dP, (85b) A' Z 59 or if the x- component is concentrated on first I = / AE I ' 1I (P,d)e r dP.
(85c) x The amplitude and phase of AEmr can be written explicitly as • ^nir ,pinr . .mr r x AE = AA e x x to recast eq.
(85c) in the form i(cJ)mr -^R.P) I = / AA™r e x r dP.
(86) x A , x The maximum value of the integral in eq .
(86) occurs when <pmT = —R»P, i.e.
I < / AA™r dP.
(87) x A , x If polarization is not changed drastically upon reflection from the mr probe and test antenna, AA will be roughly proportional to the magnitude of the x-component of electric field at the probe.
That is, Mmr a £ mr |E , } X X ' X ' ' v mr where e is an average proportionality constant between the ampli-tude of the multiply reflected x-component of electric field and the amplitude of the total x-component of electric field as the probe traverses the scan area.
Substitution of eqs .
(88) and (52) into eq.
(87) yields I < emr A r IE I .
(89a) x — x ' xo ' r->°° v In the same manner, the expression for I is found to be I < e mr A r IE I .
(89b) y — y l yo ' r->°° An upper bound expression for the fractional error n emerges when t eqs.
(89) and (85) are combined with eq .
(1): n(F) < e mr g(F), (90) 60 mr where e is the average proportionality constant between the ampli-tude of the multiply reflected electric field (probe output) and the amplitude of the total electric field (probe output) as the probe traverses the scan area.
The value of e can be estimated experi-mentally by changing the distance between the probe and test antenna by a few wavelengths at various locations within the scan area.
Periodic variations in the amplitude of the probe output which re-peated about every A/2 would be caused primarily by the multiple re-flections.
As usual, g(r) is the ratio of the amplitude of the maxi-mum far-electric-field to the far-electric -field at the given direction r.
Equation (90) represents an upper bound expression for the errors in the far-field caused by multiple reflections.
It applies to the entire far-field of electrically large aperture antennas.
However, for nearly all antenna-probe interactions it will give much too large an estimate of the far-field errors because it was derived under the unrealistic assumption that the multiply reflected fields possessed just the right phase across the scan area to maximize the far-field errors.
In reality, as the probe traverses the scan area it will usually experience large variations in the phase of the multiply reflected fields throughout the scan area that will greatly reduce their effect on the far field.
We emphasize the word "usually" because there exist some antennas (like the constrained lens, fixed-beam array show.n in figure 9) which present a rather flat reflective surface to the radiation scattered from the probe, and thus a rather constant effective path length and phase for the multiply reflected fields as the probe scans on a plane parallel to the aperture.
In that case the maximum error could be experienced at the center of the main beam.
The multiple reflections would have a minimum effect on the far-field when their phase varied radically over the scan area, or rn y V more precisely, when cj) - — R«P in eq.
(86) has no critical points and varies rapidly with position P on the scan area.
Then eq.
(86) written in polar coordinates (p, cj) ) can be integrated by parts with respect to p to give X x s 271/ X 3f/dp' P e P P' dV (91) 2tt AA™r (p',cj) ) ikf(p\ct ) o 61 where p' refers to the distance from the origin within the scan area to the point on the boundary of the scan area at the angle $ .
The function f(p', ), which is assumed to possess no stationary points p with respect to p, is defined by kf = mr x - y R ' p -(92) As mentioned above, the smallest values of I occur, in general, when the phase function f varies rapidly with p and cf> .
Experience at the NBS indicates that it is unlikely that the average phase variations in multiple reflections which occur in practice ever exceed 360° per wavelength across the scan area.
Thus it is also unlikely that | I | will be smaller than its value when f is chosen as a function which changes an average of about 360° per wavelength of motion in any direction across the scan area.
For example, one such function is f = p(l + cos 2 9 ) .
(93) Substitution of the function (93) into eq .
(91) shows that the magni-tude of I may be expressed as x 7 ^ >-£ 2, AA^p', ) / -x— p' e ikp' (l+cos 2 9^) d(J) l+cos^e p (94) Equation (94) is written as an inequality to emphasize that for nearly all antennas |l | would be larger than the right side of eq (94).
In order to get an idea of the value of eq .
(94), assume that << p' so that the points of stationary phase of p ' (l + cos 2 ^ ) d(J) p with respect to /A L ve AAave (95a) with L equal to the average of the width of the scan area, and AAave equal to the average amplitude of the x-component of the multi-ply reflected electric field at the four points <> = 0, tt/2, tt and 3tt/2 62 on the boundary o£ the scan area.
If we approximate AA by e |E I , where |E | refers to the average amplitude of the x-XX X component of electric field on the boundary of the scan area, eq.
(95a) becomes x A /, , ave mrip i ave > t~ vX L e El — 2tt x ' x ' (95b) Since II = /|I : + | I | 2 , eq.
(95b) combines with the corresponding x y equation for | I | to give y I > -^ /A L 1 — 2tt ave A yC (96) The average amplitude |E | of the transverse electric field may be expressed in terms of the maximum transverse electric field (E^ ) on r to the scan plane, -Xave E t | ave = 10 20 'to (97) .ave with X denoting the number of dB down from that maximum (see footnote 7) .
Equations (30b) and (31) can be utilized in conjunction with eqs .
(97) and (96) to give the final expression for the minimum value of Hi , Xave I I > /A L ave 10 20 mr e a A 2 r I E I (98) From eqs.
(98), (85) and (1) we find the minimum value expected for the fractional error n in the far-field caused by multiple reflections Xave n(r) > f A 1 ave 3/2 10 20 mr ,— .
a £ SO) (99) (The aperture area A has been taken as . 5(L ) 2 .) The inequality (99) represents a lower bound expression for n in the sense that the actual far-field errors caused by multiple reflections would, to a high probability, be greater than but could lie reasonably close to 63 the value of the right side of eq.
(99).
Of course, there remains the possibility that for some points in the far-field the effects of the multiple reflections will cancel to such a degree that n would actually be less than the right side of eq .
(99).
These exceptional points must be ignored if eq .
(99) is accepted as a valid lower bound Between eqs .
(90) and (99) we have approximate upper and lower limits to the value of n, Xave A 1 ^L ave 3/2 a 10 20 mr ,—.
.
,—, .
mr e g(r) n(r) e g(r) (100) where mr X a ave ave = wavelength.
= the average ratio of the amplitude of the multiply reflected probe output to the amplitude of the total probe output as the probe traverses the scan area.
(Its value can be estimated experimentally by changing the distance between the probe and test antenna at various locations within the scan area, and calculating one-half the fractional peak to peak height of the variations in amplitude that repeat about every X/ .
) = average width of the scan area.
= the average amplitude of the probe output at the boundary of the scan area measured in dB down from the maximum amplitude of probe output in the scan plane.
= a "taper" factor --equal to a minimum of 1.0 (for apertures of uniform amplitude and phase) and less than 5 for most tapered distributions found in practice (see eq.
(31) for the precise definition of a; for a difference pattern one should still use the taper factor of the constituent sum patterns) g(r) = ratio of the amplitude of the maximum far-electric-field to the far-electric-field at the given direction r, i.e., the inverse of the normalized far-field pattern.
(g(r) = 1 for the center of the main beam, or beams if a difference pattern.) 64 (If desired, upper and lower limits for the errors in the gain func-tion, sidelobe level, polarization ratio, and beamwidth may be found by combining eq.
(100) with eqs .
(2) -(5) and the far-field pattern.) The value of the factor „ave x 1 3/2 a 10 20 T avej TT for a typical microwave antenna and scan area is on the order of .001; in which case eq.
(100) becomes .001 emr g(r) < n(r) < emr g(r).
(101) It is clear from eq.
(101) that the errors in the far-field produced by multiply reflected fields of a given relative amplitude, i.e.
a given e , can span an extremely wide range of values depend-ing on the variation in phase of the multiply reflected fields across the scan area.
(For example, if the multiple reflections are down 40 dB (e = .01), eqs.
(101) and (2c) show that the multiple re-flection error in the on-axis gain of the main beam lies between about ± .0001 and ± .1 dB .
) Essentially, the right side of eq.
(101) gives the far-field errors when the effective phase of the multiply reflected fields is uniform, and the left side when the phase varies an average of 360° every wavelength across the scan area.
Because of the extremely large range in the possible value of n, it appears unlikely that a precise value of the far-field errors produced by the multiple reflections can be obtained by any method other than direct measurement.
As was mentioned at the beginning of this sec-tion, data could be taken on a number of parallel scan planes separa-ted by about A./4, and the far-fields computed for each plane.
Any differences noted in the computed far-fields for the separate scan planes would be caused primarily by the multiple reflections.
In addition, it appears likely that the effect of multiple reflections could be reduced appreciably by averaging the far-fields obtained from a number of different scan planes separated by a small fraction of a wavelength over a distance of one wavelength.
Finally, we shall evaluate the far-field errors for multiply reflected fields which are described by a class of hypothetical 65 near-field distributions.
In particular, assume that the fields are mr linearly polarized and that cb in eq .
(86) has the functional dependence , cb = 2tt 1 X 2 J (102) where A, and A~ are arbitrary real constants.
Also, assume that the test antenna has a circular aperture of radius "a" and uniform ampli-mr tude distribution.
Then for scans in the very near-field, AA in mr x eq.
(86) can be approximated by a constant (AE ) inside the aperture area and zero outside.
Under these conditions, the integral (85b) may be written in polar coordinates as ,mr ztt a 1 I^ 3 [3—cos 8 +-r—sincb -sin0cos(cb -cb)] A L A I = AE" 11 / / e J J o o p A pdpdcb (103) The ) = B cos(cb -b) A, p A ?
p p p (104) where and B = /(A/A^sinecoscb) 2 + (A/ \^- sinBsincb) (105a) b = tan t— -sinG sincb A -, sine cosi (105b) Using the integral representation of the Bessel function, 2tt J (z) = J^ / e iz cosip 2tt dijj eq.
(103) reduces to ,mr I = 2ttAE ]111 / J O J O A o 2TrpB p dp (106) 66 and eq.
(106) becomes I = AaAEmr 1 2-rraB A (107) By combining eqs .
(107), (85), and (1), the fractional error n in the far-field takes the form aAE mr n(r) = J 1 27raB A Br |E(r) (108) ;£->-oo For a circular aperture with uniform amplitude distribution E , e< (10) can be integrated to yield the far-field, E(r) I = E -(cos 2 9 + sin 2 cos 2 (j)) J 2Tra A sin6 cose (109) The definition of n(r) loses significance when its value becomes larger than 1, i.e.
when the far-field amplitude |E(r)| becomes less than the amplitude of the far-field error.
To avoid this 9-TT Q situation, J-, (—r— sine) in eq .
(109) can be approximated by its envelope , ffiz -|J]_(z) I ~ J x (z) = ' 7TZ \ z 2 ) z < 2 z > 2 (110) Substitution of eq.
(110) into eq .
(109) and the result into eq.
(108) transforms the expression for n into mr .
„ e sine n(r) = (2^aB^ 1 X 1 B(cos 2 e + sin 2 e cos 2 4)) J 2-rra sin6 (111) r mr A -cmr /X2 ^ (e = AE /E ) o ' o J If A, and X ?
(see eq .
(102)) are chosen such that A, = A ?
= A , and (J) is set equal to zero, B can be written from eq .
(105a) as A ^ o sine + sin 67 and r\ becomes mr -a e smQ n(6) = j 2Tra 4-o t— sine o +sin^0 X X X X -sinf +sin 2 6 J^ 2fTa X (112) sin9 We can compare this result with the previous maximum and "minimum" values estimated for r\ in eq .
(100) when g(r) = a = 1, and X =0.
For X = °° (uniform phase for the multiply reflected fields) eq.
(112) should equal the right side of eq .
(100).
For X = X eq.
(112) should be of the same order of magnitude as the left side of eq .
(100).
Indeed, when X eq.
(112) becomes emr (the right side of eq.
(100)), and when A = X it reduces to approxi' mately .06 (A/a) 3/2 (100) since A = na 2 was taken as .5(L ) in eq which is nearly equal to the left side of eq.
(100).
This agreement between eqs .
(112) and (110) supports the validity of both expressions Equation (112) reveals that the on-axis far-field error n(o) can be expressed in the especially simple form, n(o) = emr J X (113) Figure 14 shows the on-axis error n(o) plotted against the variable /2i\a/X .
Note that for X < 2a, n(o) is given approximately by .06 3/2 cos 8.9 X mr mr and never gets larger than about .1 z" x .
That is, if the phase of the multiply reflected fields changes 360° or more across the diameter of the aperture area, the error in the on-axis far-electric- field is mr mr less than one-tenth e , where e is the ratio of the amplitude of the multiply reflected electric field (probe output) to the total electric field (probe output) as the probe traverses the effective scan area.
Only when the phase of the multiply reflected fields is uniform (X = °°) across the scan area does the on-axis far-field o , mr error equal e In figure 15, n(6) is plotted for different values of a/X Q from eq.
(112) for a/A = 12.
When X is greater than a couple of aperture AC diameters, the envelope of the errors from multiple reflections are mr on the order of e throughout the far-field.
As X becomes less & o than a couple of diameters, the near-axis errors grow much smaller mr than e but the envelope of errors in the far sidelobes remains at mr about e .
Finally, when X gets as small as the free-space wave-mr length A, the far-field errors are much smaller than e all the way to 6 = 30° in the far-field.
IV.
Summary The far-field characteristics of a radiating test antenna can be determined throughout the forward hemisphere by scanning on a near-field plane of the test antenna with a probe antenna of arbi-trary but known receiving characteristics.
The amplitude and phase of the probe output are recorded on the near-field scan plane and the far-field pattern is computed by "deconvolving" the near-field data.
The accuracy of the computed far-field depends upon the size of the scan area, the accuracy with which the scanner positions the probe, the accuracy with which the instrumentation measures the amplitude and phase of the probe output, the extent of the multiple reflections, the computation errors involved with deconvoluting the near-field data, and, of course, the accuracy with which the probe is calibrated and the input power to the test antenna is measured.
Essentially, this report derives upper bound expressions for the errors in the far-field pattern produced by these sources of error in the near-field measurements.
The upper-bound expressions are written in a form that can be used to stipulate design criteria for the construction of near-field scanning facilities.
In particular, the limits of accuracy in a given far-field parameter are expressed in terms of the measured near-field data and/or the computed far-field, the frequency and dimensions of the antenna-probe system, the systematic and random variation in the positioning of the scanner, and the precision of the instrumentation which measures the probe output.
In order to simplify the mathematics, the probe was usually assumed to be an electric dipole, although the resulting upper bound expressions hold for arbitrary probes.
69 The analysis and resulting upper-bound expressions are not re-stricted to a particular antenna as previous computer studies [11,16] and direct far-field comparisons have been, but apply generally to electrically large aperture antennas which can be operating in either a sum or difference pattern.
The results for position and instrumen-tation errors apply only to nonscanning antennas in the sense that the analysis for these two sources of error assumed that the scan plane lay perpendicular or nearly perpendicular to the boresight direction of the antenna pattern.
Position and instrumentation errors for beams which are steered away from the perpendicular to the scan plane will be included in a subsequent report by Newell Except for the position and instrumentation errors, however, the re-sults of the present report, and in particular the error expressions pertaining to the truncation of the scan plane, apply to arbitrarily steered antennas.
Broadbeam antennas where the wavelength is on the same order of magnitude as the dimensions of the aperture are not examined in this report.
But it is shown in a report by Crawford et al .
that many of the conclusions and upper bound expressions derived here apply directly or in slightly modified form to broadbeam antennas It is emphasized that the upper-bound expressions derived in this report determine the limits of accuracy of the far-field com-puted from the planar near-field scanning technique without resorting to comparisons with direct far-field measurements-.
This is probably the foremost purpose of the report along with the report acting as an aid in deciding design criteria and tolerances for the construc-tion of new near-field scanning facilities.
It has been the feeling of those involved with near-field measurement techniques at the NBS that often the near-field techniques determine the far-field more accurately than conventional far-field measurements with a standard antenna.
Thus comparison with measurements made on conventional 'far- field" ranges would not be a reliable method, even if it were feasible, for estimating the accuracy of the near-field techniques, which do not have the problems of proximity corrections, ground re-flections, or the need of a standard far-field antenna.
A brief summary of the major conclusions and results of the report follows: 70 Errors in computation can be ignored immediately.
A simple exercise in Section I showed that their effect on the far-field is extremely small compared to the effects of the other sources of error.
Far-field errors from the approximation involved in applying the sampling theorem are also negligible (see footnote 1) .
In Section II it was demonstrated that errors in various far-field parameters could be expressed conveniently in terms of the fractional far-electric-field error (n = I AE I / I E I ) and the approxi-vi I I'll p-^-oo ' r tr mate far-field pattern.
In particular, the errors in gain function, sidelobe level, polarization ratio, and beamwidth were expressed in terms of n (see eqs.
(l)-(5)).
In Section III. A the far-field errors associated with neglecting the near-fields outside the finite scan area were investigated.
First it was shown analytically that no reliable information about the far-field pattern could be obtained by the planar near-field scan method outside the "solid angle" formed by the edge of the antenna aperture and the boundary of the scan area.
Well within this solid angle (0 < ~-, see eq.
(33) and following), reasonable upper l m cix bound expressions for the finite scan errors were found that could be applied to center-line data (eq.
(32)) as well as full-scan data (eq.
(36) or (32)).
As part of the finite scan analysis, asymptotic expres sions for the near-fields in front of a circular antenna of uniform aperture distribution were derived in Appendix A and plotted in figures A3 and A4 .
A comparison was made between the empirical analysis of finite scan errors performed by Newell and Crawford with centerline data and the maximum errors calculated from the upper bound expression (32).
Agreement is quite reasonable, as figures 8a and 8b indicate.
Table 1 shows the finite scan error for various far-field parameters of a typical x - band antenna.
The finite scan errors are proportional to wavelength, so changing the wavelength while holding the other antenna dimensions the same merely changes the values in Table 1 proportionately.
Of course, such an isolated change is rather unrealistic.
Deviations in the position of the probe from its assumed posi-tion in the scan area will produce errors in the near-field data which show up as errors in the computed far-field pattern.
Section 71 III.B.l derives an upper bound expression throughout the forward hemisphere for the far-field errors produced by both systematic and random errors in the positioning of the probe (see eqs .
(61), (64), (68) and figures 10, 11, and 12).
(Of course, a uniform displacement of the scanner does not alter the far-field pattern, and a uniform rotation of the scanner simply rotates the entire far-field pattern through the same angle.) The contribution from the inaccuracies in the position of the scanner divides naturally into a transverse or xy (parallel to the scan area) and longitudinal or z (perpendicular to the scan area) part.
For sum patterns, the on-axis far-field errors were found pro-portional to the xy-displacement errors of the scanner but propor-tional to the square of the z-displacement errors (normalized to wavelength) .
This latter result is analogous to that obtained by Ruze in his classical work on "antenna tolerance theory" for a small arbitrary phase error or aberration in the surface of an an-tenna.
This result does not imply that the z-displacement errors generally have a much smaller effect on the main beam than the xy errors, because the multiplying factor is, in general, much larger for the z term.
It does imply, however, that random and systematic displacement errors in the longitudinal or z direction weigh equally in their contribution to errors near the center of the main beam of sum far-field patterns.
Unlike the z errors, random errors in the transverse or xy-displacement of the scanner have a negligible effect throughout the far-field compared to systematic transverse errors of the same order of magnitude, and the xy-position errors do not depend on wavelength.
The xy-position errors also differ from the z-position errors in that the same xy error expression applies throughout the far-field hemisphere whereas the z errors are given by one expression close to the main beam, i.e., near the boresight direction, and another expression for the errors off -axis.
Specifically, the maxi-mum possible off-axis (or sidelobe) z-position errors depend linearly upon the systematic z-displacement errors of the scanner and are much larger than the on-axis errors, which are proportional to the square of the z-displacement errors.
In fact, as Tables 2 and 4 indicate, these off-axis errors caused by displacements in the 72 z-position of the scanner can be much larger than the total off-axis errors from all other sources combined.
The present report does not examine in detail the relationship between the distribution of z-displacement errors and the off -axis or sidelobe errors, but such an analysis will be included as part of the subsequent report by Newell .
For difference patterns, the effect of the near-field displace-ment errors on the far-field were given by the same upper-bound expressions derived for sum patterns, except for the effect of z-displacement errors on the null depth.
This error in null depth for difference patterns was shown in Appendix B to be independent of the value of the null depth itself and proportional to the z -displacement errors, but, in general, proportional by a very small proportionality constant.
In fact, for most antennas and reasonably accurate scan-ning systems, the z-displacement errors do not affect the null depth by more than a few tenths of a dB .
This rather surprising result, which is rather difficult to explain without going through the mathe-matics of Appendix B, has been confirmed by an empirical error analysis performed by Newell on the measured near-field data of a number of antennas operating in the difference mode.
Although the z-displacement errors do not have a strong influence on the depth of the null of difference patterns, eq .
(68), which was also derived in Appendix B, shows that they do have a strong influence on the direc-tion of this null.
In fact, the effect of all other sources of errors combined on the null direction of difference patterns is generally negligible compared to the effect of z-position errors, although instrumentation phase errors can sometimes shift the null an appreciable amount as well.
Table 2 shows the xy-and z-position error in various far-field parameters for a typical X-band and K-band antenna.
Again, note the strong influence that z-position or phase errors can have on the off-axis far-field parameters (sidelobe level, beamwidth, mainlobes of difference patterns) and on the null shift of the difference pattern.
Note also that the maximum possible null shift caused by z-position errors is independent of frequency.
73 The far-field errors caused by the inaccuracies of the receivers in measuring the phase and amplitude of the probe output are esti-mated in Section III.B.2 (see eqs.
(82), (83) and figures 11-13).
(It should be mentioned that instrumentation errors associated with converting analog to digital information is assumed negligible.) As would be expected, the instrumentation errors in measuring phase contribute to the far-field errors in exactly the same way as longi-tudinal or z-position errors of the scanner.
However, since the errors which a typical receiver introduces into the phase are small and increase monotonically with decreasing amplitude of the probe output, calculations show that their effect is often negligible throughout the far-field.
Although typical instrumentation errors in measuring phase are small and often introduce insignificant errors into the far-field, typical instrumentation errors in measuring amplitude can have a pronounced effect on the far-field (see figure 13), except for the far sidelobe region and in the null depth of difference patterns.
In general, if high accuracy is desired, the receiving system which measures the amplitude of the probe output should be calibrated and the calibration curves included as part of the computer program which deconvolutes the near-field data to get the far-field.
It is significant, however, that the instrumentation errors in measuring near-field amplitude have a relatively small affect on the null depth of difference patterns, as eq .
(82) and Table 3 demonstrate.
The reason for this is that the receiver distorts the opposite lobes of the near-field amplitude by approximately the same amount (see Appendix B) .
Table 3 also displays the amplitude error in various other far-field parameters for a typical microwave antenna.
The in-strumentation amplitude errors for 8 < 2X/£ do not depend directly upon frequency or the size of the test antenna, only upon the taper factor a of the near-field amplitude, the receiver inaccuracy N, R , and, except for the null depth of difference patterns, the inverse of the normalized far-field pattern g(r).
It should also be pointed out that the instrumentation amplitude errors in the far sidelobe region, > 10X/ imax , are relatively small, usually less than a few tenths of a dB for N, R less than a few thousands of a dB per dB 74 Whenever comparisons were possible, the expressions for both position and instrumentation errors (eqs.
(61), (64), (68), (82), and (83)) agreed well as an upper-bound with the results of the empirical error analysis of Newell et al .
[11,17,21] at the National Bureau of Standards, and with the error analysis performed by Rodrigue et al.
at the Georgia Institute of Technology with a hypothetical near-field distribution (see figures 11-13).
In Section III.C upper and "lower" bound expressions were derived for the far-field errors caused by multiple reflections (see eq.
(100)).
The upper and lower bounds were extremely far apart because the phase of the multiply reflected fields has a strong in-fluence on the far-field errors.
Since the phase of the multiply reflected fields for a given test and probe antenna interaction would, in general, be difficult to measure or estimate, it was con-cluded that the only reliable way to get an accurate estimate of the far-field errors from multiple reflections would be to use the fol-lowing straightforward but tedious procedure.
Take several near-field scans on parallel planes separated by about 1/4 wavelength or less.
Any deviations in the far-field patterns computed from the data on the separate scan planes would be caused primarily by the multiple reflections (assuming the scan area is large enough so that changes in finite scan errors are negligible).
If necessary, the effect of multiple reflections could probably be reduced appreciably by averaging the amplitude of the far-fields obtained from these different scan planes separated by a small fraction of a wavelength and covering a total change in separation distance of one wavelength In addition to the upper and lower limits of error, the far-field errors were derived for multiply reflected fields which satisfy a certain class of hypothetical near-field distributions (see eq.
(112) and figures 14 and 15) .
The major analytical results of this error analysis study can be combined into one long upper-bound expression for the fractional far-electric- field error n(6,(|0.
(6 and specify the angular spherical coordinates of the far-field pattern.) 75 n(e,cj)) , T max, n aAL 10 20 aAA' max A sine n probe max 11max 2N dB (3l) + 2 mr 2 input null shift of 2A shift difference pattern — max 7T£ 6 S +A X max Tmax radians (114a) (114b) (6 max a<kLj 2 + max (6 rn ) v max ; 8AF(6 S +A0 1 ) v max Y max ; r B 1 -< max (o-l) 2/g(e,<fr) (a-l)/2 , . max aAL 3A (sum patterns) (difference patterns) (sum and difference patterns) (sum patterns) (difference patterns) (sum and difference patterns) 10Amax 10Amax 10£max A 10£ 10A max <e< 2A ,max £max <e< I Equations (114) represent essentially an amalgamation of eqs .
(32), (64), (68), (82), (83) and (100), under the extreme condition that the various near-field errors combine in such a way as to create the maximum possible far-field errors.
If desired, the tighter upper-bound, eq .
(36), for the finite scan error could be used for the first term in eq .
(114a) instead of eq.
(32).
Recall from Section III. A. 3 that the finite scan error term represents a valid upper-bound halfway or more within the "solid angle" formed by the edges of the aperture and the boundary of the scan area (9 < _ 9 = ~-(90-y ) ; and r ' v "• 2 max 2 K 'max' outside this solid angle region the planar near-field scanning tech-nique cannot be relied upon with any confidence to yield accurate far-fields.
The instrumentation amplitude factor 3 is not given explicitly by eqs.
(114) in the region between 9 equal to 2A/£ and 10A/£ .
However 3 can be estimated in this region by connect-ing a straight line from its value at 2X/1 to its value at 10A/£ The detailed derivation of each of the terms in eq.
(114), except 76 n , and n.
.
, can be found in the part of the main text from probe input r which each particular term came.
The extra terms n , and n.
r probe input will be explained below.
The definition of the various parameters in eqs .
(114) can also be found from the preceeding main text: X = wavelength.
A = area of the antenna aperture.
TH £L X I = maximum width of the antenna aperture.
L = maximum width of the scan area.
y = maximum acute angle between the plane of the scan area 'max & r and any line connecting the edges of the aperture and scan area (0 = 90-y ) .
^ max 'max' X = the largest amplitude of the probe output at the edge of the scan area, measured in dB down from the maximum amplitude of probe output in the scan plane.
a = a "taper" factor- -equal to a minimum of 1.0 (for apertures of uniform amplitude and phase) and less than 5 for most tapered distributions found in practice.
(See eq.
(31) for the precise definition of a; for a difference pattern one should still use the taper factor of the constituent sum patterns .
) A = 2ttAP /A, where AP is the maximum amplitude of the max max ' max r transverse (xy) displacement errors within the effective scan area A .
(A is that part of the scan area over o^o ^ which the phase is fairly uniform.
For near-field scans parallel to the aperture A -A.) 8 = 2ttAz /A, where Az is the maximum amplitude of the max max ' max r longitudinal (z) displacement errors within the effective scan area A .
I ° Acb = the maximum instrumentation errors (expressed in radians) Y max y r j involved in measuring the phase of the probe output on the effective scan area A (see footnotes 10 and 13) .
o AF = fractional difference between the amplitude of the two main far-field lobes of the difference pattern (see Appendix B) .
Njt, = the maximum instrumentation errors involved in measuring the amplitude of the probe output --N, R is expressed in dB error per dB amplitude down from the maximum amplitude 77 on the scan area.
(The amplitude error is designated as zero at the maximum amplitude; see footnote 11.) mr z = the average ratio of the amplitude of the multiply re-flected probe output to the amplitude of the total probe output as the probe traverses the scan area.
(Its value can be estimated experimentally by changing the distance between the probe and test antenna at var-ious locations within the scan area, and calculating one-half the fractional peak to peak height of the varia-tions in amplitude which repeat about every A/2.) g(9,) = 1 for the center of the main beam, or beams if a difference pattern.) The superscripts "s" and "rn" refer to the "systematic" and "random" parts of the displacement errors respectively.
The instrumentation phase and amplitude errors (A\ , N, n ) are assumed systematic in I max ai5 i max nature.
The phase error Ac}) does not show up in n for 9 > A/(10£ ) in cix.
z because, as explained in Section III.B.2, the shape of the near-field instrumentation phase error is such that it has negligible effect in this off-axis region compared to the maximum effect of typical sys-s tematic z-position errors 6 .
Random position errors have non-r max v negligible effect only near the boresight direction of sum patterns.
mr The multiple reflection ratio e is enclosed in a box in eq.
(114a) to emphasize that for most antennas it represents an unreal-istically large upper bound.
The error n , (9,cb) simply represents the uncertainty in the probe v ,rJ r j r receiving characteristics of the probe in the direction corresponding to (9,(J)).
For example, if the receiving characteristic S'-.
of the probe (see reference ) was known to an accuracy of 1% for the direction (9,6), then n , would be .01 for that direction.
k.
r j y 'probe The error n .
.
arises in normalizing the amplitude of the input r probe output to the input power or amplitude |a | of the input mode to the test antenna.
Such a measurement is necessary whenever abso-lute values of the gain function are required.
Probably the simplest and most accurate method of performing this normalization is to con-nect the input waveguide of the test antenna directly to the output waveguide of the probe through a variable attenuator.
If, as ex-plained in Section III.B.2, the receiver which measures the amplitude of the probe output is specified arbitrarily to have zero error when the probe output is at its maximum amplitude on the scan area, then the normalization can be accomplished by measuring the attenuation needed to reduce the amplitude of the direct input from the trans-mitter to the level of the maximum amplitude of the probe output on the scan area.
Of course, "mismatch factors" of any consequence must also be measured.
The quantity n.
.
merely denotes the com-n 7 input J bined fractional error of the variable attenuator and of the devices used to measure the necessary mismatch factors.
By using a high precision attenuator, the fractional error n.
.
can usually be r ' input ' kept below a few thousandths.
Table 4 shows the total maximum possible error in a number of far- field parameters for a typical X-band and K-band antenna and a reasonably accurate scanning facility.
The table was computed from eqs.
(114) for the representative values of the near-and far-field parameters listed above the table.
It is emphasized that the values shown in Table 4 are the maximum possible upper-bounds to the far-field errors computed under the extreme condition that each of the sources of near-field error produce its maximum possible change in the far- field and then all these maximum changes in the far-field add in phase.
The z-position errors have been separated and under-lined when they represent the dominant contribution to the upper-bound errors, because whether or not these maximum possible z-position errors actually occur depends strongly on the far-field direction of interest and on the shape of the deviation in z-position throughout the scan area.
As mentioned above, a detailed analysis of this de-pendence will be included as part of a subsequent report by Newell .
79 The error Cn be + ^ ut ) in the receiving characteristic of the probe and in normalizing to the input power of the test antenna was chosen as . 1 dB in Table 4.
For the error in on-axis gain of the sum patterns, Table 4 shows that this contribution of .1 dB is about as large as all the other errors combined.
The same is true for the gain of the mainlobes of the difference pattern if the z-position part of the errors is ignored.
Thus, for the situation described by Table 4, greater accuracy in the calibration of the probe and in the measuring of the input power to the test antenna could be a first step in significantly reducing the errors in the direction of maximum gain.
It is interesting to compare the maximum equivalent reflected signal allowable in a conventional "far-field" range or anech-oic chamber to get accuracies comparable to those shown in Table 4 for the near-field scanning technique.
It is a simple matter to show that the maximum equivalent reflected signal (ERS) measured in dB down from the direct signal is related to the values of ri, R by n dB ERS = 20 log -X—~ (assuming small ri,J .
For example, near the boresight direction the ERS would have to be 33 dB down for the sum pattern and 15 dB down for the difference pattern to give the corresponding errors shown in Table 4.
Of course, there would also be errors on conventional ranges due to proximity effects, uncertainties in the calibration of the standard antenna, and instrumentation errors.
In conclusion, it can be seen from Tables 1-4 that for a reason-ably accurate scanning system no one source of error dominates over all the others in the boresight direction of both sum and difference patterns.
However, in the far-field region away from the boresight direction, but well within the solid angle region formed by the edge of the aperture and boundary of the scan area, the deviation in z-position of the scanner can, in principle, cause far-field errors which are much larger than the combined errors of all other sources.
In practice, however, these maximum possible errors seldom occur.
Moreover, it will be shown in the subsequent report by Newell 80 that the scanner can be designed and utilized to keep the off-axis, z-position errors far below the upper-bound values given by eqs .
(114) and shown in Tables 2 and 4.
Of course, beyond the solid angle region formed by the edge of the aperture and boundary of the scan area, it was shown in Section III. A that the far-field computed from the near-field data cannot be relied upon with any confidence.
Ef-2 E AE Figure 1.
Main beam of a hypothetical test antenna, Aperture of Test Antenna »-z Figure 2.
Schematic of scanning geometry 82 Test Antenna Figure 3.
Schematic of aperture antenna, Figure 4.
Definition of a , 6 , D .
6 m m m 83 P.Pm Dm /j m point / rm r^-] t "Xem R ^\ 9 >o'm »-0' j rl 0' % Figure 5.
Schematic of aperture and scan areas.
(Although A and A' are drawn parallel, it is not a necessary requirement of the theory.) e=tan~'(-/2-l) Reflector Figure 6.
Circular reflector antenna.
84 -5 -10 GO -a S " l5 <u -o B -20 "5.
<f -25 <-o 2-30 o a> >-35 UJ -40 -45 -80 -60 -40 -20 20 40 60 80 Probe X- position in cm Figure 7a.
Near-field centerline data (constrained lens) (z = 25 cm) .
-50 -40 -30 -20 -10 10 20 Probe X-position in cm 50 Figure 7b.
Near-field centerline data (reflector antenna) (z = 43.18 cm).
.15 CD T3 C .05 Q> o> c D t_ o c o o X o c " .05 o 15 Upper bound \ \ Newell & Crawford 2.6 24 2.2 2.0 1.4 1.0 (L /2a) -Scan Length /Aperture Diameter Figure 8a.
Change in gain vs.
decreasing scan length (constrained lens) CD c o .c o c "o CO c o .25 20 10 .05 .05 10 15 .20 .25 \ A\ s Newel Crav i a ford Upper boun d / \ i \ \ \N i s 2.05 1.85 1.65 1.45 1.25 1.05 .85 .65 45 .25 .05 (Lmox/2a)- Scan Length /Aperture Diameter Figure 8b.
Change in gain vs.
decreasing scan length (reflector antenna) 86 Figure 9b Figure 9a.
Constrained lens sum port near-field log amplitude, £=9.2 GHz, z-25.0 cm, no radome.
Near-field phase, constrained lens sum port, f = 9.2 GHz, z = 25 cm.
Figure 9c.
Constrained lens antenna and probe 87 288 252 216 d) CD w m <v IHO o c a; 144 in o si a.
IOR 100 -80 -60 -40 -20 20 40 60 80 100 Probe X-Position in cm Figure 9d.
Near-Field Centerline Data, z = 25 cm 88 ±1.4 ±1.2 ±1.0 ±0.8 ±0.6 ±0.4 ±0.2 Figure 10.
Position errors in on-axis gain.
±.5i— ±10 ±20 Figure 11.
Comparison with Rodrigue et al.
for random phase errors.
89 ±.25 ±20 CO "3 ±15 o I c m P- ±10 ±.05 Upper bound Newell mm ——— — ±1 ±2 ±3 ±4 , ±5 ±6 ±7 ±8 Degrees (±A<£ max ) Figure 12.
Comparison with Newell for quadratic phase errors.
±07 ±06 ±.05 ±04 CD ±.03 ±.02 ±.01 a=3.0 y a=2.5 /Rod / rigueetaL (a=3) >X / a =2.0 X / V Newell (a=2) .^ / a=l.5 Jr " X ±.001 ±.002 ±.003 ±.004 ±.005 ±.006 ±007 ±008 ±009 N^B (dB per dB down) Figure 13.
Amplitude errors in on-axis gain.
90 Figure 14.
On-axis error in far-field from multiple reflections.
a/X =I.O I.Oc env elope \ -K — 2>r<«h r 0° 6° 12° 18° 24° 30° 6 a/X =8.0 Figure 15.
Far-field errors from multiple reflections (a/A = 12) .
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•H o •H •H T3 CO o 03 X rH £ X I— 1 PL, 03 CU E 03 rH r— i c i xs 03 i rH r— 1 •H c •H CU d 3 3 rt o en PQ o Z z "'• CU u d d c !-H ca Jh E cu f-H CU P +-> CU +-> CO +-> Mh -m 03 Mh oj a, •H a \ 03 c c cu M d < d 03 CQ d 03 X o3 U • H p-X E-03 H O M-t (O !h O H LU T3 PL, i 03 •M O CU rH 03 95 Appendix A Near-Fields of a Circular Antenna with Uniform Aperture Distribution The purpose of this Appendix is to derive expressions for the near-field of a circular antenna of uniform aperture distribution.
Specifically, the amplitude and phase are constant within the aper-ture and zero outside.
The antenna is electrically large in the sense that the diameter of the aperture is many wavelengths across.
The transverse electric field to the right of the aperture is given exactly by eq.
(16).
If we assume that the aperture fields are linearly polarized in the x-direction, eq .
(16) becomes E (r) = x v J 2tt 9 z -E .
2tt a iklr-R' O 8 r r Q l a 7 i J . _ _ , R'dR'dtJ)' (Al) o o | r-R ' for the circular aperture of radius a and uniform amplitude E .
The element of area has been written in terms of the polar coordinates (R',cf)') of the vector R'.
In addition it can be proven from eq.
(15) that if the aperture fields have longitudinal components which are small compared to the transverse components, then the longitudinal components remain relatively small in the near-field (z << a 2 /A) , provided a/A >> 1.
The change of variable t = R'-R, where R is the transverse part of r, converts eq.
(Al) to E (r) x v J -2iE o 3z ikz 7T ik/t 2 (a) + z 2 / e da o Ra (A2b) The variables t , t, and t and the limit of integration a, are o 1 2 l defined in figures Ala and Alb.
Apparently, eqs .
(A2) were first derived by Schoch For large k, the integrals in eqs.
(A2) can be evaluated by the method of stationary phase.
The integrand of eq.
(A2a) has two stationary points, one at a = and one at a = tt.
The two terms in 96 the integrand of eq.
(A2b) each have a single stationary point at a = 0.
After carrying through the details of the method of station-ary phase, eqs .
(A2) become E (r) = E [e lkz +AI] Ra, where AI is given by AI = ^ M 2tt R i [k/z 2 +(R+a) 2 -£] i [k/z 2 + (R-a) 2 +J] (R+a)y^ 2 +(R+a) 2 (R-a) /z 2 + (R-a) 2 (A3a) (A3b) (A3c) Equation (A3b) is identical to eq .
(17) of the main text when the following substitutions are made: D 1 = /z 2 +(R+a) D 2 = /z 2 +(R-a) 2 a.
cosO-, = z/(R+a) cos6~ = z/(R-a) Strictly speaking, eqs.
(A3) are valid only as k = 2tt/A approaches infinity.
For finite k they represent first term approximations to an infinite asymptotic series, and the first term approximations are valid only in certain regions of the near-field.
These regions of validity can be estimated by returning to eqs.
(A2) .
First concen-trate on eq.
(A2a) .
In eq.
(A2a) t ranges from (a-R) to (a+R) .
Thus if the rate of change of t is somewhat uniform with a on the & o interval a = to it, the method of stationary phase will yield a good approximation when /(R+ a) 2 + z /(a-R) 2 +z 2 >> 1.
(A4) Unfortunately t does not change uniformly with a from a = to it, but it does so in the separate intervals to y and y to it.
Since t 2 = a 2 -R 2 at a = y, the condition (A4) must be replaced by the two conditions , 97 /(R+a) 2 + z 2 A 2 R 2 + z 2 > 3 (A5a) , , , , /a 2 -R 2 +z 2 -/(a-R) 2 +z 2 > 3.
(A5b) (Actually the right side of eqs .
(A5) should be >> 1 but experience with the method of stationary phase indicates it remains a good approximation down to where the quantities in brackets in eqs.
(A5) are just a few wavelengths -- specif ically 3A is chosen in eqs.
(A5).) Manipulation of eqs.
(A5) shows that they are satisfied for large a/A if 3A < R < [a- (12A/a)A] (A6a) z < ^y /R 2 -(3A) 2 .
(A6b) That is, eq.
(A3a) approximates eq .
(A2a) in the region defined by eqs.
(A6) .
A similar analysis with eq .
(A2b) shows that it is approximated by eq.
(A3b) in the region defined by R > [a+(12A/a) A] (A7a) a z < (R-a) 3A' (A7b) The regions of validity defined by eqs.
(A6) and (A7) are shown in figure A2 for an a/A equal to 12.
Each region is labeled by the equation which approximates the field in that region.
In addition to the regions defined by eqs.
(A6) and (A7) two other regions--one near the z-axis, R < 7a R < the larger of 2a z or 3(^) 2/3 A (A8a) (A8b) and one near the edge of the aperture are shown in figure A2 z > 2 r a z > Tra a-R R-a 1 < 4 a (A9a) (A9b) (A9c) 98 In the axis region defined by eqs .
(A8) , the exponent in the integrand of eq.
(A2a) can be approximated by the first two terms of its expansion in R/a, and integrated to give E (r) = E x v J o ikz ze i k /z 2 + a : J /z 2 ka R ^2+a 2 ^ (A10) z"+a' Similarly, in the edge region defined by eqs.
(A9) , the eq.
(A2b) can be integrated approximately to give E (r) = J x v J 2 ikz ze ik/z 2 + 2a 2 J /z 2 + 2; ka : Vz 2 + 2a 2^ (All) In the near-field in front of the aperture (z less than about Yy a 2 /A) , eqs.
(A3a) and (A10) reveal that the electric field can be approximated by a single equation, with E (r) = E [e lkz + AI ' ] , x v J o L J ' (A12a) f i[k/z 2 +(R+a) 2 -J] ( AI za /2 y { (R+a)/z 2 +(R+a) 2 i[k/z 2 +(R-a) 2 +|] ka R •/z^+CR+a) 2 (R-a)/z 2 +(R-a) 2 ka R Vz 2 + (R-a) and J defined as o J (z) = o v J (1 - i z 2 ) z < 1.46 z > 1.46.
(A12b) (A12c) Essentially, eqs.
(A12) represent the total electric field for the linearly polarized circular aperture in the region R < a-A (A13a) 1 a 2 z < 12 A (A13b) 99 The amplitude |E | and phase 6 of E v may be found from eqs.
CA121 , 1— [Q! cos(k/z 2 + (R+a) 2 -kz-|) -Q 2 cos (k/z 2 + (R-a) 2 -kz +}1 a E (A14a) kz za /2 coskz Q 1 sin(k/z 2 + (R+a) 2 -^) -Q sin(k/z 2 + (R-a) 2 - J) sinkz Q 1 cos(k/z 2 +(R+a) 2 -|) -Q 2 cos (k/z 2 + (R-a) 2 +j) with (A14b) J' ka /z 2 +(R+a) 2 R (A14c) (R+a)/z 2 + (R+a) 2 Q?
= T e ka p «j _ K u /z 2 +(R-a) 2 (A14d) (R-a)/z 2 +(R-a) 2 Equations (A14) hold (to a first order approximation) everywhere in the near-field region defined by eqs.
(A13) .
Except within a wavelength or so of the z-axis, eqs.
(A14) reveal that in this region 1 X the maximum amplitude fluctuations are about ± — / — , and the phase varies slightly from kz.
The amplitude of the electric field given by eq.
(A14a) is plotted in figure A3 for a/A = 12 and 15, and for increments of z/a from to y~ a/A.
The amplitude curves show large fluctuations within a wavelength of the points of maxima and minima along the z-axis.
These z-axis maxima and minima, which are a well-known phenomenon for the circular aperture of uniform distribution, are less pronounced for noncircular apertures, or for apertures with a tapered distribution (see reference , Section 1-F).
Figure A3 also shows that the amplitude variations across the aperture repeat about every wavelength in the very near-field (z < a/4) , but spread out as the distance from the aperture gets larger.
This behavior has been observed for tapered distributions as well, by the many near- field measurements on microwave antennas performed at the National Bureau of Standards 100 The phase in the near-field of a circular antenna with a/A = 12 is plotted in figure A4 .
Except near the zeros in on-axis amplitude, the phase across the beam is uniform to within a few degrees of oscillation which repeat about every wavelength in the very near-field.
101 (a) b) Figure Al .
Definition of t and a.
Aperture Figure A2.
Dotted line shows region in which eq .
(A12a) holds 102 A/\ f\s\P , r r (z/a = .75) \j .2 .4 .6 R/a w\A/ S. S". /\ / \J V ) --.
vv (z/a = .5) ' \/ c) .; > .4 .( R/a 3 .8 (z/a = .25) .2 .4 .6 R/a |E X I p rw\ A A /\ z' n ^V (z/a = 1.25) 2 4 6 R/a (z/a 1.0) r^sA?\P\/ 2 .4 6 R/a IE, A/\ f\~riA>k / 1/v^ (z/a= .75) vy .2 .4 .6 R/a IE J \i'>^V/I A fl A /I A n U/Q--5) ^/v .2 .4 .6 R/a (z/a = .25) .2 4 6 R/a (a) a/A = 12 ^ a/A = 1S Figure A3.
Near-field amplitude of circular antenna.
103 77 .
.2 .4 .6 .8 R/a (z/a=.5) 77 .
-7T 2 .4 R/a (z/a = .25) 6 .8 IT +.
77" .2 .4 6 .8 R/a (z/a=I.O) 77 .
-77 .2 .4 .6 .8 R/a (z/a=75) Figure A4 .
Near-field phase of circular antenna (a/A = 12).
104 Appendix B Position and Instrumentation Errors for the Null Depth of Difference Patterns 1 .
Position Errors The errors in null depth of a difference pattern caused by transverse displacement errors (AP) of the near-field scanner can be derived in the same way and are given by the same upper bound ex-pression as that of the transverse errors for sum patterns (eq.
(60a)).
However, the sum pattern derivation of Section III.B.l for longitudinal displacement errors cannot be applied to finding errors in the null depth of difference patterns.
The main reasons for this are twofold.
The first and most obvious is that the equation for difference patterns analogous to eq .
(48) may not be satisfiable be-cause the near- field of a difference pattern changes phase by 180° across the aperture.
Secondly, as the analysis below shows, the greatest effect of a longitudinal displacement error is a slight shift in the position of the null rather than a change in the depth of the null.
If the longitudinal position errors were zero and the perpen-dicular (e ) to the scan plane were parallel to the null axis, the far-field in the direction of the "null" is given in terms of the near-field by eq.
(10): null m 1 e ik(r-d) j p- d dp -(B1) Ao where, as usual, A refers essentially to that part of the scan area that covers the antenna aperture, and just the x-component of electric-field will be considered first.
Now if the amplitude of E (P,d) on opposite halves of the aper-x ture (and thus A ) were equal and the phase difference exactly 180 the field in the direction of the null axis would actually be zero.
In reality the amplitudes are not exactly equal and the phase dif-ference, even on the average, is not exactly 180°.
Specifically, we can write E (P,d) as 2v 105 Ex (P,d) i( +kd) (A +AA1 e (over one half, A ,) A.
A 01 A e x i( +kd+Aip ) ^ r ox r x' (over second half, A 2 ) (B2) with AA /A and Axp both << 1.
For simplicity has been chosen XX X ox constant, although it can be shown that the results do not change significantly if cf> is allowed to vary slightly across A .
Substi-0X iAip ° tution of eq.
(B2) into (Bl) and approximating e x by (1 + iAip ) yields ,null i(kr+c|) ) r y Y ox^ i Xr A AA dP x ol A / iAx Aij, x dP o2 (B3) for the field in the null direction.
Experimentally, it has been found at the NBS that for many if not all antennas operating in a difference mode the first integral in (B3) predominates , but for the sake of the error analysis we must retain both integrals.
When longitudinal displacement errors (Az) are introduced, eq (10) shows that eq .
(Bl) must be replaced by .null 1 x Ar i(kr+(J) ) ik(Az-sin9e D »P) ox / E fP,d) e K dP, A X o (B4) where E represents the null field computed from the actual near-field data containing errors in the z-position of the scanner.
6 can no longer be set equal to zero because the error Az may shift the angular position of the null axis.
After substituting eq.
(B2) into (B4) , expanding the exponential in a power series, and discarding error terms higher than second order, eq .
(B4) becomes, 106 .null' Jx i(kr+cf> ) r ox Ar / AA dP -/ iA Aip dP A^i A , X X ol o2 (B5) + / (±A +AA + ) A X X o ik(Az-6e D 'P) + k 2 Az0e p P (kAz) 2 _ Ck9eR -P) , 2 2 'R dP R k / Ax A^ x (Az-9e R -P) dP A o2 with AA in A , x ol AA in Ao2 and the + and -sign before A being used in the areas A , and A ~ 6 x to ol o2 respectively.
The first order error term in eq .
(B5) can be made zero by choosing 0=6 such that to s / (±Ax+AAx ) Az dP = s / (±Ax +AAx )(e R -P) dP o o (B6) corresponds essentially to the shift in the direction of the null caused by longitudinal position errors.
If we take the weighted i — i III £LX average value of [£P| at about I /4, eq.
(B6) implies that for most antennas Imax < Azmax ' ,max where I denotes the maximum width of the antenna aperture defining 6 = 2ttAz /X, eq.
(B7) becomes approximately, max max (B7) By (B8) With the choice of given by eq.
(B6) the amplitude, ,null ' r,null i x E"" | , of the error field in the null direction may be written by subtracting eq.
(B3) from eq.
(B5) , 107 pnull ' pnull x x <h k 2 / ± A A x Az0 e n P s R (Az) Ce s e R .P) 2 dP k / Ax AiP x (Az-6 s eR .P) dP o2 (B9) The first term in the first integral of eq .
(B9) can be made zero by merely shifting the reference plane from which Az is measured.
The third term in the first integral is identically zero because it is an odd function over the scan area A o c null ' ^null i .
k x x ' — Ar Thus eq .
(B9) reduces to k / A A^j, (Az-6 |P|) dP + / ± A (Az) A , X X S Z A X o2 o (BIO) The second integral in eq.
(BIO) is also negligible if we assume the rms value of (Az) is approximately equal on the "positive" and "nega-tive" sides of A .
Finally, since the maximum value of is 26 A/tt£, J s max ma) eq.
(BIO) becomes cnull' c null x x ave k 6 A max Ar Imax / A Aij, |P| dP A X X o2 x a i a ve o Alp max x Ar / A dP, A , X o2 (Bll) with AiJj denoting the average phase difference from it radians of the probe output between the positive and negative sides of the partial scan area which is perpendicular to the null axis .
The factor t— / A dP is approximately equal to the maximum Ko2 field in the mainbeams of the difference pattern.
And since an ex-pression analagous to eq.
(Bll) holds for the y-component of the field, we can write the fractional error ri(r) near the null of a difference pattern as ^ 1 6 max A^ave ^ r ) ' (B12a) where, as in the main text, g(r) is the ratio of the amplitude of the maximum far-electric-field to the far-electric-field at the given direction r.
Here r is essentially the direction of the null axis and thus g(r) the ratio of maximum far-field to null axis far-field.
Now eq.
(B12) is a very simple expression.
However, the average phase difference A^ of the probe output may be a difficult 108 quantity to estimate accurately from the near-field phase data.
For-tunately, it can be shown that Aip is related to the relative size of the two mainlobes in the far-field of the difference pattern.
Specifically, a straightforward but rather lengthy manipulation (which will not be shown here) of the near-field of a difference pat-tern reveals that Ail» can be approximated by r ave ^ r J Ail; -4AF y ave where AF is the fractional difference between the amplitude of the two main far-field lobes of the difference pattern.
For example, if the amplitude of one mainlobe is 10 on some linear scale and the amplitude of the other mainlobe is 10.1 then AF would equal (10.1-10)/10 or .01.
Equation (B12a) can now be written in the alternative form ^ i 46max AF ^' (B12b) 2 .
Instrumentation Errors The inaccuracies in measuring near-field phase have the same effect on the null-depth as the longitudinal position errors.
That is, A , the maximum instrumentation error in measuring phase, simply replaces 6 in eqs .
(B8) and (B12) .
IT13.X The inaccuracies in measuring amplitude affect the null depth differently, however, than the transverse position errors because the nonlinearities in the instrumentation distort the amplitude on each side of the difference pattern in nearly the same way and thus their effect on the null depth is much smaller than might first be expected.
Specifically, errors in measuring amplitude show up in the null depth through distortion of only the AA part (see eq.
(B2)) of the near field.
Carrying through an analysis similar to that per-formed above for z-position errors yields the following expression for the maximum change AE.
in null electric field caused by ampli-tude errors: |AEnull| 1 d| ; |AA|) dp £ dB max ol ?
(B13) Aol Xr 109 where AA is the maximum of AA across A , and N JT.
is the maximum max ol dB instrumentation error in measuring the amplitude of the probe output (see Section III.B.2).
If we approximate |E.
|, the x-component AAmaxA j of which is given in eq.
(B3) , by ^ , then the fractional z Ar error in null depth caused by instrumentation errors in measuring amplitude can be written (B14) Note that the instrumentation amplitude error in null depth does not depend on the null depth itself.
In addition, it can be shown that, unlike phase errors, the instrumentation amplitude errors have negligible effect on the angular position of the null direction.
110 References IEEE Standard Definitions of Terms for Antennas , IEEE Transac-tions on Antennas and Propagation, AP- 22 , 1 (January 1974).
DeSize, L.K.
and J.F.
Ramsay, in Microwave Scanning Antennas , Ed.
R.C.
Hansen, Academic Press--New York, London (1964), Vol.
1, Ch.
2-II.B.
Kerns, D.M., "Correction of Near-Field Antenna Measurements Made with an Arbitrary But Known Measuring Antenna," Electronics Letters, 6, 11, pp.
346-347 (28th May 1970).
See e.g.
Jackson, J.D., Classical Electrodynamics , John Wiley § Sons Inc., New York, London (1962), Section 9.6.
a Braunbek, W.
, "Zur Beugung an der Kreisscheibe , " Zeitschrift Fur Physik, 122, pp.
405-415 (1950).
b Ufimtsev, P. la., "Approximate Computation of the Diffraction of Plane Electromagnetic Waves at Certain Metal Bodies," Soviet Physics--Technical Physics, 2, 8, pp.
1708-1718 (August 1957).
a Keller, J.B., "Diffraction by an Aperture," Journal of Applied Physics, 2^, 4, pp.
426-444 (April 1957).
b Keller, J.B., "Geometrical Theory of Diffraction," Journal of the Optical Society of America, 5_2, 2, pp.
116-130 (February 1962) .
Van Kampen, N.G., "An Asymptotic Treatment of Diffraction Problems," Physica, 14, 9, pp.
575-589 (January 1949).
Keller, J.B., R.M.
Lewis, and B.D.
Seckler, "Diffraction by an Aperture II," Journal of Applied Physics, 2_8, 5, pp.
570-579 (May 1957) .
a Kouyoumjian, R.G., and P.H.
Pathak, "A Uniform Geometrical Theory of Diffraction for an Edge in a Perfectly Conducting Surface," Proceedings of the IEEE, 62, 11, pp.
1448-1461 (November 1974) .
b Hwang, Y.M.
and R.
G.
Kouyoumjian, "A Dyadic Coefficient for an Electromagnetic Wave Which is Rapidly-Varying at an Edge," URSI 1974 Annual Meeting, Boulder, Colorado.
See e.g., Born M.
, and E.
Wolf, Principles of Optics , Pergamon Press, Oxford, New York (1970), 4th Ed., Appendix III.
Ill Newell, A.C.
and M.L.
Crawford, "Planar Near-Field Measurements on High Performance Array Antennas," NBSIR 74-380, National Bureau of Standards, Boulder, Colorado (July 1974).
Kraus , J.D., Antennas , McGraw-Hill, Inc., New York, Toronto, London (1950), Chs .
3 and 12.
Hansen, R.C., (Ch.
1, Sect.
2-1 in ref ): A /A = G/G in Table IX, p.
66.
Rusch, W.V.T.
and P.D.
Potter, Analysis of Reflector Antennas , Academic Press, New York -London (1970), Section 2.52.
Ruze, John, "Antenna Tolerance Theory --A Review," Proceedings IEEE, 54, 4, pp.
633-640 (April 1966).
Rodrigue, G.P., E.B.
Joy, and C.P.
Burns, An Investigation of the Accuracy of Far-Field Radiation Patterns Determined from Near-Field Measurements , Report --Georgia Institute of Technology, Atlanta, Georgia (August 1973).
Newell, A.C, private communication, National Bureau of Stand-ards, Boulder, Colorado.
Kerns, D.M., "Scattering-Matrix Description and Near-Field Mea-surements of Electroacoustic Transducers," The Journal of the Acoustical Society of America, 5_7, 2, pp.
497-507 (February 1975) Schoch, V.A.
, "Betrachtungen liber das Schallfeld einer Kolbenmembran," Akustische Zeitschrift, 6, pp.
318-326 (1941).
Crawford, M.L., A.C.
Newell, J.W.
Greene, and A.D.
Yaghjian, "Experimental Design Study for a Near-Field Broadbeam Antenna Pattern Calibrator," AFAL-TR- 75-179 , Air Force Avionics Lab., Wright-Patterson Air Force Base, Ohio 45433 (to be published).
Newell, A.C, "Planar Near-Field Measurement Techniques on High Performance Arrays --Part II," Air Force Technical Report, Air Force Avionics Laboratory -- Wright Patterson Air Force Base, Ohio (to be published) Newell, A.C, R.C Baird, and P.F.
Wacker, "Accurate Measurement of Antenna Gain and Polarization at Reduced Distances by an Extrapolation Technique," IEEE Transactions on Antenna and Propagation, AP-21 , 4, pp.
418-431 (July 1973).
112 Kanda, M., "Accuracy Considerations in the Measurement of the Power Gain of a Large Microwave Antenna," IEEE Transactions on Antennas and Propagation, AP-23 , 3, pp.
407-411 (May 1975).
Appel -Hansen, J., "Reflectivity Level of Radio Anechoic Chambers," IEEE Transactions on Antennas and Propagation, AP-21 , pp.
490-498 (July 1973) .
Jensen, Frank, "Electromagnetic near- field-far-field correla-tions," Ph.D.
Dissertation, Technical University of Denmark, Lyngby, Denmark (July 1970).
ACKNOWLEDGMENT The author is grateful to Allen C.
Newell for his invaluable consultation throughout the preparation of this report.
Much appreciation is extended to John .W.
Greene and Douglas P.
Kremer for the information they provided about the NBS near-field scanning facilities.
Mark T.
Ma and Carl F.
Stubenrauch of the Department of Commerce, and Kenneth Grimm of the Air Force Avionics Laboratory contributed many helpful suggestions in reviewing the report.
Many thanks also go to Janet R.
Jasa for carefully and patiently typing the manuscript .
113 U.S. DEPT. OF COMM.
BIBLIOGRAPHIC DATA SHEET 1. PUBLK A 1 ION OK HI- PORT NO.
NBS Tech Mote 667 2.
Ciov't Accession No.
4. TITLE AND SUB' Upper-Bound Errors in Far-Field Antenna Parameters Determined from Planar Near-Field Measurements, Part I : Analysis 3. Recipient's Accession No.
5. Publication Date October 1975 6. Performing Organization Cod< 276.05 7. AU rHOR(S) Arthur D Y a g h j i a n 8. Performing Organ. Report No.
TN-667 9. PERFORMING ORGANIZATION NAME AND ADDRESS NATIONAL BUREAU OF STANDARDS, DEPARTMENT OF COMMERCE WASHINGTON, D.C. 20234 Bou 1 der Labs 10. Projccr/Task/Work Unit No.
2765276 11. Contract Grant No.
12. Sponsoring Organization Name and < omplete Address (Street, City, State, ZIP) Air Force Avionics Laboratory Air Force Wright Aeronautical Laboratories Air Force Systems Command Wright-Patterson Air Force Base, Ohio 45433 13. Type of Report & Period C overed 7/73 -7/74 14. Sponsoring Agency ( od 15, SUPPLEMENTARY NOTES 16. ABSTRACT (A 200-word or less factual summary of most significant inforniation.
If document includes a significant bibliography or literature survey, mention it here.) General expressions are derived for estimating the errors in the sum or difference far-field pattern of electrically large aperture antennas which are measured by the planar near-field scanning tech-nique.
Upper bounds are determined for the far-field errors produced by 1) the nonzero fields outside the finite scan area, 2) the inac-curacies in the positioning of the probe, 3) the distortion and non-linearities of the instrumentation which measures the amplitude and phase of the probe output, and 4) the multiple reflections.
Computa-tional errors, uncertainties in the receiving characteristics of the probe, and errors involved with measuring the input power to the test antenna are briefly discussed.
17. KEY WORDS (six to twelve entries; alphabetical order; capitalize only the first letter of the first key word unless a proper name; separated by semicolons) Antennas; error analysis; far^field pattern; near-field measurements; planar scanning; plane wave spectrum.
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Do Not Release to NTIS PX 1 Order From Sup. of Doc, U.S. Government Pointing Dlfice Washington, D.C. 20 402, SO Cat. No. CM.
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2611 | https://math.stackexchange.com/questions/4578882/given-the-vectors-a-and-b-such-that-ab-and-2a-b-are-perpendicular-and-a-b-and-4 | linear algebra - Given the vectors a and b such that a+b and 2a-b are perpendicular and a-b and 4a+b are perpendicular, find the angle between a and b - Mathematics Stack Exchange
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Given the vectors a and b such that a+b and 2a-b are perpendicular and a-b and 4a+b are perpendicular, find the angle between a and b
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(a+b)⋅(2 a−b)=0(a+b)⋅(2 a−b)=0
(a−b)⋅(4 a+b)=0(a−b)⋅(4 a+b)=0
2 a⋅a+a⋅b−b⋅b=0 2 a⋅a+a⋅b−b⋅b=0
4 a⋅a−3 a⋅b−b⋅b=0 4 a⋅a−3 a⋅b−b⋅b=0
b⋅b=2 a⋅a+a⋅b b⋅b=2 a⋅a+a⋅b then I found
a⋅a=2 a⋅b a⋅a=2 a⋅b
b⋅b=5 a⋅b b⋅b=5 a⋅b
Then I dont know how to continue to find an angle.
Can someone help. Appreciate that
linear-algebra
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edited Nov 17, 2022 at 17:56
MeltedStatementRecognizing
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asked Nov 17, 2022 at 16:27
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For some basic information about writing mathematics at this site see, e.g., here, here, here and here.Another User –Another User 2022-11-17 16:28:45 +00:00 Commented Nov 17, 2022 at 16:28
Does the OP realize that this is an exercise about inner products of vectors?M. Wind –M. Wind 2022-11-17 16:57:03 +00:00 Commented Nov 17, 2022 at 16:57
Hints: 1) The cosine of the angle equals the dot product divided by the product of the magnitudes. 2) The dot product of v v with itself equals the square of the magnitude of v v. Then just substitute and simplify to get the desired cosine.Ned –Ned 2022-11-17 23:52:11 +00:00 Commented Nov 17, 2022 at 23:52
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0=(a⃗+b⃗)⋅(2 a⃗−b⃗)=2 a⃗⋅a⃗−b⃗⋅b⃗+a⃗⋅b⃗=2 a 2−b 2+a b cos(θ)0=(a→+b→)⋅(2 a→−b→)=2 a→⋅a→−b→⋅b→+a→⋅b→=2 a 2−b 2+a b cos(θ)
0=(a⃗−b⃗)⋅(4 a⃗+b⃗)=4 a⃗⋅a⃗−b⃗⋅b⃗−3 a⃗⋅b⃗=4 a 2−b 2−3 a b cos(θ)0=(a→−b→)⋅(4 a→+b→)=4 a→⋅a→−b→⋅b→−3 a→⋅b→=4 a 2−b 2−3 a b cos(θ)
Multiply the first equation by 3 3, then add the second equation. It follows that 10 a 2=4 b 2 10 a 2=4 b 2, hence b=1 2 10−−√a b=1 2 10 a. Substitute this result into one of the two equations and you get cos(θ)=1 10 10−−√cos(θ)=1 10 10.
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answered Nov 18, 2022 at 4:37
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The Physics Classroom » Physics Tutorial » Sound Waves and Music » Reflection, Refraction, and Diffraction
Sound Waves and Music - Lesson 3 Behavior of Sound Waves
Reflection, Refraction, and Diffraction
Interference and Beats
The Doppler Effect and Shock Waves
Boundary Behavior
Reflection, Refraction, and Diffraction
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Like any wave, a sound wave doesn't just stop when it reaches the end of the medium or when it encounters an obstacle in its path. Rather, a sound wave will undergo certain behaviors when it encounters the end of the medium or an obstacle. Possible behaviors include reflection off the obstacle, diffraction around the obstacle, and transmission (accompanied by refraction) into the obstacle or new medium. In this part of Lesson 3, we will investigate behaviors that have already been discussed in a previous unit and apply them towards the reflection, diffraction, and refraction of sound waves.
Reflection and Transmission of Sound
When a wave reaches the boundary between one medium another medium, a portion of the wave undergoes reflection and a portion of the wave undergoes transmission across the boundary. As discussed in the previous part of Lesson 3, the amount of reflection is dependent upon the dissimilarity of the two media. For this reason, acoustically minded builders of auditoriums and concert halls avoid the use of hard, smooth materials in the construction of their inside halls. A hard material such as concrete is as dissimilar as can be to the air through which the sound moves; subsequently, most of the sound wave is reflected by the walls and little is absorbed. Walls and ceilings of concert halls are made softer materials such as fiberglass and acoustic tiles. These materials are more similar to air than concrete and thus have a greater ability to absorb sound. This gives the room more pleasing acoustic properties.
Reflection of sound waves off of surfaces can lead to one of two phenomena - an echo or a reverberation. A reverberation often occurs in a small room with height, width, and length dimensions of approximately 17 meters or less. Why the magical 17 meters? The effect of a particular sound wave upon the brain endures for more than a tiny fraction of a second; the human brain keeps a sound in memory for up to 0.1 seconds. If a reflected sound wave reaches the ear within 0.1 seconds of the initial sound, then it seems to the person that the sound is prolonged. The reception of multiple reflections off of walls and ceilings within 0.1 seconds of each other causes reverberations - the prolonging of a sound. Since sound waves travel at about 340 m/s at room temperature, it will take approximately 0.1 s for a sound to travel the length of a 17 meter room and back, thus causing a reverberation (recall from Lesson 2, t = d/v = (34 m)/(340 m/s) = 0.1 s). This is why reverberations are common in rooms with dimensions of approximately 17 meters or less. Perhaps you have observed reverberations when talking in an empty room, when honking the horn while driving through a highway tunnel or underpass, or when singing in the shower. In auditoriums and concert halls, reverberations occasionally occur and lead to the displeasing garbling of a sound.
But reflection of sound waves in auditoriums and concert halls do not always lead to displeasing results, especially if the reflections are designed right. Smooth walls have a tendency to direct sound waves in a specific direction. Subsequently the use of smooth walls in an auditorium will cause spectators to receive a large amount of sound from one location along the wall; there would be only one possible path by which sound waves could travel from the speakers to the listener. The auditorium would not seem to be as lively and full of sound. Rough walls tend to diffuse sound, reflecting it in a variety of directions. This allows a spectator to perceive sounds from every part of the room, making it seem lively and full. For this reason, auditorium and concert hall designers prefer construction materials that are rough rather than smooth.
Reflection of sound waves also leads to echoes. Echoes are different than reverberations. Echoes occur when a reflected sound wave reaches the ear more than 0.1 seconds after the original sound wave was heard. If the elapsed time between the arrivals of the two sound waves is more than 0.1 seconds, then the sensation of the first sound will have died out. In this case, the arrival of the second sound wave will be perceived as a second sound rather than the prolonging of the first sound. There will be an echo instead of a reverberation.
Reflection of sound waves off of surfaces is also affected by the shape of the surface. As mentioned of water waves in Unit 10, flat or plane surfaces reflect sound waves in such a way that the angle at which the wave approaches the surface equals the angle at which the wave leaves the surface. This principle will be extended to the reflective behavior of light waves off of plane surfaces in great detail in Unit 13 of The Physics Classroom. Reflection of sound waves off of curved surfaces leads to a more interesting phenomenon. Curved surfaces with a parabolic shape have the habit of focusing sound waves to a point. Sound waves reflecting off of parabolic surfaces concentrate all their energy to a single point in space; at that point, the sound is amplified. Perhaps you have seen a museum exhibit that utilizes a parabolic-shaped disk to collect a large amount of sound and focus it at a focal point. If you place your ear at the focal point, you can hear even the faintest whisper of a friend standing across the room. Parabolic-shaped satellite disks use this same principle of reflection to gather large amounts of electromagnetic waves and focus it at a point (where the receptor is located). Scientists have recently discovered some evidence that seems to reveal that a bull moose utilizes his antlers as a satellite disk to gather and focus sound. Finally, scientists have long believed that owls are equipped with spherical facial disks that can be maneuvered in order to gather and reflect sound towards their ears. The reflective behavior of light waves off curved surfaces will be studies in great detail in Unit 13 of The Physics Classroom Tutorial.
Diffraction of Sound Waves
Diffraction involves a change in direction of waves as they pass through an opening or around a barrier in their path. The diffraction of water waves was discussed in Unit 10 of The Physics Classroom Tutorial. In that unit, we saw that water waves have the ability to travel around corners, around obstacles and through openings. The amount of diffraction (the sharpness of the bending) increases with increasing wavelength and decreases with decreasing wavelength. In fact, when the wavelength of the wave is smaller than the obstacle or opening, no noticeable diffraction occurs.
Diffraction of sound waves is commonly observed; we notice sound diffracting around corners or through door openings, allowing us to hear others who are speaking to us from adjacent rooms. Many forest-dwelling birds take advantage of the diffractive ability of long-wavelength sound waves. Owls for instance are able to communicate across long distances due to the fact that their long-wavelength hoots are able to diffract around forest trees and carry farther than the short-wavelength tweets of songbirds. Low-pitched (long wavelength) sounds always carry further than high-pitched (short wavelength) sounds.
Scientists have recently learned that elephants emit infrasonic waves of very low frequency to communicate over long distances to each other. Elephants typically migrate in large herds that may sometimes become separated from each other by distances of several miles. Researchers who have observed elephant migrations from the air and have been both impressed and puzzled by the ability of elephants at the beginning and the end of these herds to make extremely synchronized movements. The matriarch at the front of the herd might make a turn to the right, which is immediately followed by elephants at the end of the herd making the same turn to the right. These synchronized movements occur despite the fact that the elephants' vision of each other is blocked by dense vegetation. Only recently have they learned that the synchronized movements are preceded by infrasonic communication. While low wavelength sound waves are unable to diffract around the dense vegetation, the high wavelength sounds produced by the elephants have sufficient diffractive ability to communicate long distances.
Bats use high frequency (low wavelength) ultrasonic waves in order to enhance their ability to hunt. The typical prey of a bat is the moth - an object not much larger than a couple of centimeters. Bats use ultrasonic echolocation methods to detect the presence of bats in the air. But why ultrasound? The answer lies in the physics of diffraction. As the wavelength of a wave becomes smaller than the obstacle that it encounters, the wave is no longer able to diffract around the obstacle, instead the wave reflects off the obstacle. Bats use ultrasonic waves with wavelengths smaller than the dimensions of their prey. These sound waves will encounter the prey, and instead of diffracting around the prey, will reflect off the prey and allow the bat to hunt by means of echolocation. The wavelength of a 50 000 Hz sound wave in air (speed of approximately 340 m/s) can be calculated as follows
wavelength = speed/frequency
wavelength = (340 m/s)/(50 000 Hz)
wavelength = 0.0068 m
The wavelength of the 50 000 Hz sound wave (typical for a bat) is approximately 0.7 centimeters, smaller than the dimensions of a typical moth.
Refraction of Sound Waves
Refraction of waves involves a change in the direction of waves as they pass from one medium to another. Refraction, or bending of the path of the waves, is accompanied by a change in speed and wavelength of the waves. So if the media (or its properties) are changed, the speed of the wave is changed. Thus, waves passing from one medium to another will undergo refraction. Refraction of sound waves is most evident in situations in which the sound wave passes through a medium with gradually varying properties. For example, sound waves are known to refract when traveling over water. Even though the sound wave is not exactly changing media, it is traveling through a medium with varying properties; thus, the wave will encounter refraction and change its direction. Since water has a moderating effect upon the temperature of air, the air directly above the water tends to be cooler than the air far above the water. Sound waves travel slower in cooler air than they do in warmer air. For this reason, the portion of the wavefront directly above the water is slowed down, while the portion of the wavefronts far above the water speeds ahead. Subsequently, the direction of the wave changes, refracting downwards towards the water. This is depicted in the diagram at the right.
Refraction of other waves such as light waves will be discussed in more detail in a later unit of The Physics Classroom Tutorial.
Jump To Next Lesson:
Natural Frequency
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2613 | https://stats.stackexchange.com/questions/467233/combinatory-probability-group-composed-by-different-balls-of-different-colors | combinatorics - Combinatory probability, group composed by different balls of different colors - Cross Validated
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Combinatory probability, group composed by different balls of different colors
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Let's say you have a group of M M balls of different colors in a box. For example, 20 balls are red, 15 are blue, 10 are green, 5 are grey, 5 are yellow and 5 violet, for a total of M=60 M=60 balls. You pick 1⩽n⩽M 1⩽n⩽M of them without replacement. The order of the colors does not count, so for instance, if n=2 n=2 and you pick red then gray, it is the same as picking gray then red.
How do I calculate the probability of all the possible outcomes for n n elements? Is there a general formula for this problem? In particular, if M M and n n are large then the number of possible combinations is large, so how can I find the most probable combinations?
probability
combinatorics
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edited May 19, 2020 at 22:47
Ben
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asked May 18, 2020 at 20:06
FabrizioFabrizio
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1 "Probability of all the possible outcomes" seems either too inclusive or too vague. Specifically, if 4 balls drawn without replacement, then probability of getting exactly 2 red, 1 green and 1 yellow is (20 2)(10 1)(5 1)(25 0)(60 4).(20 2)(10 1)(5 1)(25 0)(60 4). –BruceET Commented May 18, 2020 at 21:22
1 See Wikipedia on 'multinomial distribution' for more. –BruceET Commented May 18, 2020 at 21:30
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Yes, there is a general formula. Consider an urn containing M M balls, where M 1 M 1 balls have the color c 1 c 1, M 2 M 2 balls have the color c 2 c 2,..., M r M r balls have the color c r c r, and M 1+⋯+M r=M M 1+⋯+M r=M. If you draw a sample of size n library(extraDistr)
K <- 10 # sample size
x <- subset(expand.grid(red=0:20, blue=0:15, green=0:10, gray=0:5, yellow=0:5, violet=0:5), red+blue+green+gray+yellow+violet==K)
dim(x)
2625 6
head(x)
red blue green gray yellow violet
11 10 0 0 0 0 0
31 9 1 0 0 0 0
51 8 2 0 0 0 0
71 7 3 0 0 0 0
91 6 4 0 0 0 0
111 5 5 0 0 0 0
tail(x)
red blue green gray yellow violet
739201 0 0 0 2 3 5
753986 1 0 0 0 4 5
754006 0 1 0 0 4 5
754321 0 0 1 0 4 5
757681 0 0 0 1 4 5
776161 0 0 0 0 5 5
p <- dmvhyper(x, n=c(20,15,10,5,5,5), k=K)
max(p)
0.008930581
x[which.max(p),]
red blue green gray yellow violet
159646 3 2 2 1 1 1
dmvhyper(x[which.max(p),], n=c(20,15,10,5,5,5), k=K)
0.008930581
choose(20,3)choose(15,2)choose(10,2)555/choose(60,K)
0.008930581
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edited May 20, 2020 at 13:30
answered May 19, 2020 at 22:05
Image 8: Sergio's user avatar
SergioSergio
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\ Usually, comments are not for this but your answer is absolutely beautiful and impressive. I was able to get a result but yours is amazing and efficient. –Fabrizio Commented May 20, 2020 at 13:32
\ You are welcome. Happy to be useful! –Sergio Commented May 20, 2020 at 13:35
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. In the following case, I extracted 12 elements and I did a statistics using 10 million extractions.
library(parallel)
library(plyr)
name of the elements of the set
acolors=c("B","G", "O", "R", "Y", "Gr")
generating a real set with a certain composition
alist=c(rep("B",20), rep("G", 15), rep("O", 10), rep("R",5), rep("Y",5), rep("Gr",5) )
number of extraction to simulate
pulls=10000000
parallel version of the extraction
all_res=mclapply(1:pulls, function(x, alist, acolors){
ares=NULL
asamp=list(table(sample(alist, 12)))
for(ac in acolors){
if(is.na(asamp[ac])){
ares=c(ares,0)
}
else{
ares=c(ares,asamp[ac])
}
}
return(ares)
}, alist=alist, acolors=acolors, mc.cores=8)
tdata store the result of each extraction. Each column has a given name
corresponding to "acolors"
a line can look like 4,2,3,1,1,1, that mean 4 from color B, 2 from color G and so on...
tdata=as.data.frame(do.call(rbind, all_res))
colnames(tdata)=acolors
now we do the statistics of the result, we count how many times a given line is duplicated
stat_res=as.data.frame(ddply(tdata,.(B, G, O, R, Y, Gr),nrow))
we sort the data frame from the most probable to the least probable
stat_res=stat_res[order(stat_res$V1, decreasing = TRUE ),]
we calculate the frequency of each line
stat_res$frequency=stat_res$V1/sum(stat_res$V1)
```
With the results is then possible to use the binomial formula given by @BruceET and calculate the probability for the most frequent results.
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answered May 19, 2020 at 20:15
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2614 | https://www.ck12.org/flexi/precalculus/sums-of-geometric-series/ | Sums of Geometric Series
Arithmetic Series Counting with Permutations and Combinations
Concept Summary:
| |
| A geometric series is a sum of numbers whose consecutive terms form a geometric sequence, and it can be finite or infinite. The sum of a finite geometric series can be calculated using the formula: where is the first term, is the common ratio, and is the number of terms. An infinite geometric series can be either convergent or divergent, depending on the value of the common ratio If the infinite geometric series is divergent, and its sum does not go to a specific number. If |r| < 1, the infinite geometric series is convergent, and its sum can be calculated using the formula: A partial sum of an infinite sum is the sum of all the terms up to a certain point. |
Represent geometrically the following numbers on the number line :
Represent geometrically the following numbers on the number line :
Represent geometrically the following numbers on the number line :
Is the sequence geometric?
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What is the sum of the infinite geometric series?
What is the sum of an infinite geometric series?
Which geometric series converges?
Which geometric series represents 0.4444 as a fraction?
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Infinite Geometric Series Determine whether the infinite geometric series is convergent or divergent. If it is convergent, find its sum.
Infinite Geometric Series Determine whether the infinite geometric series is convergent or divergent. If it is convergent, find its sum.
Infinite Geometric Series Determine whether the infinite geometric series is convergent or divergent. If it is convergent, find its sum.
Infinite Geometric Series Determine whether the infinite geometric series is convergent or divergent. If it is convergent, find its sum.
Directions: Consider the following function: f(x) = (1-x)^{-2}. Find the Radius of Convergence for this series. If necessary, write INF for ∞. R=1
Directions: Consider the following function: f(x) = (1-x)^{-2}. Write out the Maclaurin series for f(x) (in sigma notation). f(x) = ∑_{n=0}^{∞} ____
Find a power series solution of the differential equation given below. Determine the radius of convergence of the resulting series, and use the series given below to identify the series in terms of familiar elementary functions. (10x-1)y' + 10y = 0
Determine the maximum value of the sum S, given by , over all sequences of nonnegative real numbers satisfying .
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When written out in full, the number has 4041 digits. What is the sum of the digits of this 4041-digit number?
Find the nth term of the sequence 1/1, 1/2, 1/3,...
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Evaluate: $$\sum_{n=1}^{6} 4(3)^{n-1}$$
Evaluate: S =
Find the missing term in the geometric sequence: 6, [?],
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2615 | https://www.immunopathol.com/Article/ipp-41724 | Association between metabolic syndrome and risk of endometrial cancer; a systematic review and meta-analysis
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eISSN: 2423-8015
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Submitted: 29 Jul 2024
Accepted: 17 Nov 2024
ePublished: 25 Nov 2024
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Immunopathol Persa. 2025;11(2): e41724.
doi:10.34172/ipp.2025.41724
Scopus ID:86000186538
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1 CITATION 1 Total citation 1 Recent citation n/a Field Citation Ratio n/a Relative Citation Ratio
Meta-analysis
Association between metabolic syndrome and risk of endometrial cancer; a systematic review and meta-analysis
Sheida Abbasi 1, Jalal Rezaei 2, Fariba Bazi 3, Moloud Alsadat Mousavi 4, Sadaf Rassouli 5, Zeinab Zamanpour 6, Elmira Hosseini 7, Marziyeh Noori 8, Zahra Hamidi Madani 9
1 Department of Gynecology and Obstetrics, School of Medicine, Tehran University of Medical Sciences, Tehran, Iran
2 Department of Critical Care Nursing, School of Nursing and Midwifery, Tehran University of Medical Sciences, Tehran, Iran
3 Department of Reproductive Health and Midwifery, Faculty of Medical Sciences, Tarbiat Modares University, Tehran, Iran
4 Department of Gynecology and Obstetrics, Imam Hossein Hospital, School of Medicine, Shahid Beheshti University of Medical Sciences, Tehran, Iran
5 Department of Gynecology and Obstetrics, Imam Khomeini Hospital, School of Medicine, Sari University of Medical Sciences, Sari, Iran
6 Department of Gynecology and Obstetrics, School of Medicine, Jundishapur University of Medical Sciences, Ahvaz, Iran
7 Department of Gynecology and Obstetrics, School of Medicine, Urmia University of Medical Sciences, Urmia, Iran
8 Department of Gynecology and Obstetrics, Shahid Akbarabadi Hospital, School of Medicine, Iran University of Medical Sciences, Tehran, Iran
9 Reproductive Health Research Center, Department of Gynecology and Obstetrics, School of Medicine, Guilan University of Medical Sciences, Rasht, Iran
Corresponding Author: Zahra Hamidi Madani, Email: Tannaz.hamidi@yahoo.com
Abstract
Introduction: Endometrial cancer is one of the most prevalent female malignancies, with various factors, including metabolic syndrome, contributing to its incidence. Thus, this study aims to evaluate the association between metabolic syndrome and the risk of endometrial carcinoma.
Materials and Methods:In this systematic review and meta-analysis, two independent authors searched electronic databases, including Cochrane, PubMed, ProQuest, Web of Science, and the Google Scholar search engine up to May 16, 2024. Data analysis was conducted using STATA 14 software with a significance level of P< 0.05 for all tests.
Results: A pooled analysis of 12 observational studies found that metabolic syndrome elevated the risk of endometrial carcinoma by 37% overall (OR: 1.37, 95% CI: 1.33, 1.42), 35% in cohort studies (OR: 1.35, 95% CI: 1.29, 1.42), and 40% in case-control studies (OR: 1.40, 95% CI: 1.33, 1.48). However, hypertension increased the risk of endometrial carcinoma by 25% (OR: 1.25, 95% CI: 1.18, 1.33), fasting hyperglycemia by 25% (OR: 1.25, 95% CI: 1.15, 1.37), hypertriglyceridemia by 17% (OR: 1.17, 95% CI: 1.13, 1.21), low high density lipoprotein (HDL) by 20% (OR:1.20, 95% CI: 1.12, 1.28), increased waist circumference by 59% (OR:1.59, 95% CI: 1.43, 1.77), pre-menopausal period by 67% (OR:1.67, 95% CI: 1.38, 2.02), and post-menopausal period by 61% (OR: 1.61, 95% CI: 1.17, 2.21). Likewise, obesity almost doubled the risk of endometrial carcinoma (OR: 2.13, 95% CI: 1.65, 2.75).
Conclusion: Metabolic syndrome increases the risk of endometrial carcinoma, with obesity being the most dangerous risk factor for endometrial cancer. Thus, managing metabolic disorders in women can be an important step toward reducing the incidence of endometrial cancer.
Registration: This study has been compiled based on the PRISMA checklist, and its protocol was registered on the PROSPERO (ID: CRD42024551509) and Research Registry (UIN: reviewregistry1838).
Keywords:Metabolic syndrome, Endometrial neoplasms, Endometrial carcinoma, Reaven syndrome X, Insulin resistance syndrome X, Endometrium cancer
Citation: Abbasi Sh, Rezaei J, Bazi F, Alsadat Mousavi M, Rassouli S, Zamanpour Z, Hosseini E, Noori M, Hamidi Madani Z. Association between metabolic syndrome and risk of endometrial cancer; a systematic review and meta-analysis. Immunopathol Persa. 2025;11(2):e41724. DOI:10.34172/ipp.2025.41724.
× Association between metabolic syndrome and risk of endometrial cancer; a systematic review and meta-analysis
Immunopathol Persa. 2025;11(2): e41724.
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2616 | https://www.jendodon.com/article/S0099-2399(05)60951-X/fulltext | Pulpal Pain Diagnosis—A Review - Journal of Endodontics
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Clinical ArticlesVolume 26, Issue 3p175-179 March 2000
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Pulpal Pain Diagnosis—A Review
I.B.Bender, DDS
I.B.Bender, DDS
Affiliations
Dr. Bender is Chairman Emeritus, Department of Dentistry, Albert Einstein Medical Center, and Professor Emeritus, School of Dental Medicine, University of Pennsylvania, Philadelphia, PA
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Affiliations & Notes Article Info
Dr. Bender is Chairman Emeritus, Department of Dentistry, Albert Einstein Medical Center, and Professor Emeritus, School of Dental Medicine, University of Pennsylvania, Philadelphia, PA
Footnotes:
This study was supported by a grant from the I. B. Bender Research Endowment Fund, Albert Einstein Medical Center (Philadelphia, PA).
DOI: 10.1097/00004770-200003000-00012 External LinkAlso available on ScienceDirect External Link
Copyright: © 2000 The American Association of Endodontists.
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Evidence gathered from our studies and the work of others appears to support the presence of two distinct nerve pain pathways in the dental pulp, represented by fast conducting A-delta and slow conducting C-fibers. Each of these types of fibers has different pain characteristics: A-delta fibers evoke a rapid, sharp, lancinating pain reaction, and C-fibers cause a slow, dull, crawling pain. Pain response thresholds vary in different regions of the tooth, and thermal, osmotic, ionic, and electric stimuli involve different mechanisms to provoke nerve excitation of the dental pulp. Evidence also points to the fact that the incidence of pain increases as the histopathosis worsens. On interrogation, patients who manifest severe or referred pain almost always give a previous history of pain in the tooth with the ache. Eighty percent of patients who give a previous history of pain manifest histopathologic evidence of chronic partial pulpitis with partial necrosis, the untreatable category, for which endodontics or extraction is indicated. The other 20% exhibit histopathosis of the pulp with slight inflammation to chronic partial pulpitis without necrosis, a treatable category. Clinically, one can determine the degree of pulp histopathosis by asking the patient about a previous history of pain in the involved tooth. This history of previous pain adds another dimension in diagnosis for the clinician as to whether the painful pulpitis is reversible. This information also aids in referred pain localization.
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