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8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.1.1 RF frequency accuracy (Precision) | Purpose Successful processing of OFDM signals requires that certain carrier frequency accuracy be maintained at the transmitter. Specific network operations modes such as SFN require high accuracy of the carrier frequency. Interface L, M Method The 8k mode of the DVB-T always has a continual pilot, with continuous phase along successive OFDM symbols, exactly at the channel centre (k = 3 408). Its frequency may be directly measured by any spectrum analyser that has an integrated counter and at least a resolution filter of 300 Hz or less (if necessary by utilizing a reference source of sufficient accuracy). The 2k mode has a continual pilot with continuous phase at k = 1 140. Its frequency may be directly measured by any spectrum analyser that has an integrated counter and at least a resolution filter of 300 Hz or less (if necessary by utilizing a reference source of sufficient accuracy). The centre channel frequency may be inferred by subtracting to the measured frequency: 8 MHz channels: 285 714 Hz i.e. [(1 140 - 852) × 4 464,2 857 = 1 285 714 Hz]. 7 MHz channels: 1 125 000 Hz i.e. [(1 140 - 852) × 3 906,25 = 1 125 000 Hz]. 6 MHz channels: 964 286 Hz i.e. [(1 140 - 852) × 4 464,2857 = 964 286 Hz]. NOTE: For 2k mode this method may have some inaccuracy if the sampling frequency of the modulator is not precise, however such error in the sampling frequency would need to be very high to significantly affect the centre channel measurement. Should more accuracy needed, the two outer continual pilots may be measured as indicated under clause 9.1.2 RF channel width, and the mean of the two values be calculated. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.1.2 RF channel width (Sampling Frequency Accuracy) | Purpose Channel width measurements are convenient for verification that sampling frequency accuracy is maintained at the modulator side. Interface L, M Method The occupied bandwidth of a COFDM modulated channel depends directly from the frequency spacing and this from the sampling frequency. The outermost carriers in a DVB-T signal are continual pilot carriers. Their frequencies are measured (see clauseE.1) and the difference between them should be compared to the nominal channel width of 7 607 142,857 Hz for 8 MHz channels, 6 656 250,000 Hz for 7 MHz channels and 5 705 357,143 Hz for 6 MHz channels. NOTE: Three decimal places are given here for completeness only. Accuracy of 1 Hz at 5 MHz means 0,2 × 10-6 per Hz, which may be enough for most cases of sampling frequency measurement. Measurement instruments should have better accuracy and resolution (typically in the order of ten times) than the required measurement accuracy. If the frequency of the outermost carriers is known, see clauses E.1.3 and E.1.4 for how to measure them, then the related values may be calculated as per table below. Denoting the outermost pilot frequencies as FL and FH appropriately the occupied bandwidth is OB = FH _ FL. The number of carriers is K, and for the 2k mode K-1 = 1 704 while for the 8k mode K-1 = 6 816. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 57 Table 9.2: Calculated values 8k mode 2k mode Occupied bandwidth FH - FL Frequency Spacing (FH - FL)/6 816 (FH - FL)/1 704 Useful duration 6 816/(FH - FL) 1 704/(FH - FL) Centre channel 1st IF (FH - FL) × 4 096/(K-1) (FH - FL) × 1 024/(K-1) Sampling Frequency (FH - FL) × 16 384/(K-1) (FH - FL) × 4 096/(K-1) |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.1.3 Symbol Length measurement at RF (Guard Interval verification) | Purpose Verification of the guard interval used in a received DVB-T signal may be carried out at RF level by careful frequency measurements. This measurement is valid in cases where there is an uncertainty on whether a modulator is correctly working and producing a signal with the expected or assigned Guard Interval. Interface L, M Method The scattered pilots produce a pulsed-like spectrum every third carrier in a DVB-T spectrum due to their repetition presence at the same phase and location every fourth symbol. The frequency difference between two contiguous spectral lines representing a scattered pilot represents the inverse of the time length of four consecutive DVB-T symbols. Measuring such frequency difference and dividing its inverse by 4 will provide the total symbol length TS of the measured signal. By subtracting the nominal useful symbol duration TU the length of the GI is found. See clause E.1 for details on the measurement procedure and symbol lengths. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.2 Selectivity | Purpose To identify the capability of the receiver to reject out-of-channel interference. Interface The measurement of the signal input level and the interferer should be carried out at the interface N, using interface W or X for the BER monitoring. Method The input power is adjusted to 10 dB above the minimum input power as defined in "Receiver sensitivity" (see clause 9.8). The C/I threshold needed for QEF operation after RS decoder (BER < 2 x 10-4 before RS decoder) should be measured as a function of the frequency of a CW interferer. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.3 AFC capture range | Purpose To determine the frequency range over which the receiver will acquire overall lock. Interface N, for the application of the test signal; Z, for the test of TS synchronization Method A signal is applied to the input of the receiver, at a level 10 dB above the minimum input power as defined in "Receiver sensitivity" (see clause 9.8). The signal is frequency shifted in steps (from below and above) towards a nominal value and the Sync_byte_error is verified according to clause 5.2.1 (Measurement and analysis of the MPEG-2 TS - First priority: necessary for decodability (basic monitoring)). ETSI ETSI TR 101 290 V1.4.1 (2020-06) 58 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.4 Phase noise of Local Oscillators (LO) | Purpose Phase noise can be introduced at the transmitter, at any frequency converter or by the receiver due to random perturbation of the phase of the oscillators. In an OFDM system the phase noise can cause Common Phase Error (CPE) which affects all carriers simultaneously, and which can be minimized or corrected by using the continual pilots. However the Inter-Carrier Interference (ICI) is noise-like, cannot be corrected. The effects of CPE are similar to any single carrier system and the phase noise, outside the loop bandwidth of the carrier recovery circuit, leads to a circular smearing of the constellation points in the I/Q plane. This reduces the operating margin (noise margin) of the system and may directly increase the BER. The effects of ICI are peculiar to OFDM and cannot be corrected for. This has to be taken into account as part of the total noise of the system. Interface Any access to Local Oscillators (LO), in transmitters, converters and receivers. Method Phase noise can be measured with a spectrum analyser, a vector analyser or a phase noise test set. Method for CPE Phase noise power density is normally expressed in dBc/Hz at a certain frequency offset from the local oscillator signal. It is recommended to specify a spectrum mask with at least three points (frequency offsets and levels), for example see figure 9.3. NOTE: See clauses A.4 and E.4 for additional information on phase noise measurements. See clause E.4.1 for some practical information. Carrier Figure 9.3: Possible mask for CPE measurements Method for ICI For the measurement of ICI, the use of multiples of the carrier spacing is recommended for the frequencies, fa, fb, fc. Table 9.3: Frequency offsets for 2 k and 8 k systems 2 k system 4,5 kHz 8,9 kHz 13,4 kHz 8 k system 1,1 kHz 2,2 kHz 3,4 kHz Typical use For manufacturing, incoming inspection and maintenance of modulators, transmitters, up/down converters and receivers, either professional or consumer type. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 59 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.5 RF/IF signal power | Purpose Signal power, or wanted power, measurement is required to set and check signal levels at the transmitter and receiver sites. Interface K, L, M, N, P Method The signal power of a terrestrial DVB signal, or wanted power, is defined as the mean power of the signal as would be measured with a thermal power sensor. In the case of received signals care should be taken to limit the measurement to the bandwidth at the wanted signal. When using a spectrum analyser or a calibrated receiver, it should integrate the signal power within the nominal bandwidth of the signal (n × fSPACING) where n is the number of carriers. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.6 Noise power | Purpose Noise is a significant impairment in a transmission network. Interface N,P Method The noise power (mean power), or unwanted power, can be measured with a spectrum analyser (out of service). The noise power is specified using the occupied bandwidth of the OFDM signal (n × fSPACING) where n is the number of carriers. NOTE: The term C/N should be calculated as the ratio of the signal power, measured as described in clause 9.5, to the noise power, measured as described in this clause. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.7 RF and IF spectrum | Purpose To avoid interfering with other channels, the transmitted RF spectrum should comply with a spectrum mask, which is defined for the terrestrial network. If the spectrum at the modulator output is defined by a spectrum mask, the same procedure can be applied to the IF signal (with no pre-correction active). Interface K, M Method This measurement is usually carried out using a spectrum analyser. The spectral density of a terrestrial DVB signal is defined as the long-term average of the time-varying signal power per unity bandwidth (i.e. 1 Hz). Values for other bandwidths can be achieved by proportional increase of the values for unity bandwidth. To avoid regular structures in the modulated signal a non-regular, e.g. a Pseudo-Random Binary Sequence (PRBS) - like or a programme type digital transmitter input signal is necessary. Care has to be taken that the input stage of the selective measurement equipment is not overloaded by the main lobe of the signal while assessing the spectral density of the side lobes, i. e. the out-of-band range. Especially in cases with very strong attenuation of the side lobes non-linear distortion in the measurement equipment can produce side lobe signals that mask the original ones. Selective attenuation of the main lobe has proven to be in principal a way to avoid this masking effects. However, as the frequency response of the band-stop filter has to be included in the evaluation, the whole measurement procedure may become somewhat complex. For the resolution bandwidth, the recommended values should not exceed 30 kHz. Preferred values are approximately 4 kHz. The measurement should be Noise-normalized to 4 kHz. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.8 Receiver sensitivity/dynamic range for a Gaussian channel | Purpose For network planning purposes, the minimum and maximum input powers for normal operation of a receiver have to be determined. Interface Test signals are applied and measured at interface N; interfaces W or X are used for the monitoring of BER before RS. Method The minimum and maximum input power thresholds for QEF (Quasi Error Free) operation after the RS decoder (i.e. BER < 2 × 10-4 before RS decoding) should be measured. The dynamic range is the difference between the measured values. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 60 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.9 Equivalent Noise Degradation (END) | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.9.0 General | Purpose END is a measure of the implementation loss caused by the network or the equipment where the reference is the ideal performance. Interface W or X for BER measurement; N, P or S for noise injection Method The END is obtained from the difference in dB of the C/N ratio needed to reach a BER of 2 × 10-4 before RS (outer) decoding, and the C/N ratio that would theoretically give a BER of 2 × 10-4 for a Gaussian channel (see annex A of ETSI EN 300 744 [i.9]). |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.9.1 Equivalent Noise Floor (ENF) | Purpose ENF is a measure of the implementation loss caused by the transmitting equipment where the reference is the ideal transmitter. Interface M for noise power measurement, W or X for BER measurement; N, P or S for noise injection Method The ENF is obtained from the measurement of additional noise needed to reach a BER of 2 × 10-4 before RS (outer) decoding, and the noise level that would theoretically give a BER of 2 × 10-4 for a Gaussian channel (see annex A of ETSI EN 300 744 [i.9]) as described in clause B.12. Note on END and ENF: The impact of the DVB-T transmitter on the overall system performance, when a certain DVB-T mode is being received by the reference receiver, via a Gaussian channel, is assessed by the measurement of the END. The reference receiver is in the present document defined as a DVB-T receiver which require a C/N which is 3,0 dB higher than the C/N figures indicated in ETSI EN 300 744 [i.9], on a Gaussian channel. The END is in the present document defined to be the difference between required C/N, for a BER of 2 × 10-4 after convolutional decoding on the reference receiver, using a real and an ideal DVB-T transmitter. The END is not only a characteristic of the transmitter itself but is also dependent on the used DVB-T mode and on the receiver implementation loss (this is why a fixed 3,0 dB receiver implementation loss is defined for the reference receiver). The END should not exceed [0,5] dB and should be independent of the selected guard interval. Depending on the requirements of the network operator typical END values fall in the range [0,1-0,4] dB. For the determination of the END value another parameter, the Equivalent Noise Floor ENF, can be used. As described in clause B.12, this should result in an improved accuracy for the END. As opposed to the END the ENF is relatively independent of the DVB-T mode used and on the receiver implementation loss and can therefore be used to characterize the transmitter by itself. Depending on whether there is a need for characterizing the DVB-T transmitter by itself, or whether there is a need to characterize its effect on a receiver, the ENF can sometimes be used as an alternative to END as a performance parameter. The influences of intermodulation and amplitude ripple are expected to dominate in practise in the performance parameter END. (The Group Delay response of a transmitter needs to be defined by network operators depending on the configuration in use (channel combiners, output filters, etc.).) ETSI ETSI TR 101 290 V1.4.1 (2020-06) 61 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.10 Linearity characterization (shoulder attenuation) | Purpose The "shoulder attenuation" can be used to characterize the linearity of an OFDM signal without reference to a spectrum mask. Interface M Method Apply the following procedure on the measured RF spectrum of the transmitter output signal: (a) Identify the maximum value of the spectrum by using a resolution bandwidth at approximately 10 times the carrier spacing. (b) Place declined, straight lines connecting the measurement points at 300 kHz and 700 kHz from each of the upper and lower edges of the spectrum. Draw additional lines parallel to these, so that the highest spectrum value within the respective range lies on the line. (c) Subtract the power value of the centre of the line (500 kHz away from the upper and lower edge) from the maximum spectrum value of (a) and note the difference as the "shoulder attenuation" at the upper and lower edge. (d) Take the worst case value of the upper and lower results from (c) as the overall "shoulder attenuation". NOTE: For a quick overview the value at e.g. 500 kHz can be measured directly provided that coherent interferers are not present. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.11 Power efficiency | Purpose To compare the overall efficiency of DVB transmitters. Interface M Method Power efficiency is defined as the ratio of the DVB output power to the total power consumption of the chain from TS input to the RF signal output including all necessary equipment for operation such as blowers, transformers etc. (and is usually quoted in % terms). The operational channel and the environmental conditions need to be specified. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.12 Coherent interferer | Purpose To identify any coherent interferer which may influence the reliability of the I/Q analysis or the BER measurements. Interface N or P Method The measurement is carried out with a spectrum analyser. The resolution bandwidth is reduced stepwise so that the displayed level of the modulated carriers (and of the unmodulated pilots, due to the influence of the guard interval) is reduced. The CW interferer is not affected by this process and can be identified after appropriate averaging of the trace. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.13 BER vs. C/N ratio by variation of transmitter power | Purpose To evaluate the BER performance of a transmitter as the Carrier to Noise (C/N) ratio is varied, with the measurement repeated for a range of mean transmitted output powers. This measurement can be used to compare the performance of a transmitter with theory or with other transmitters. Interface From F to U or from E to V Method A Pseudo-Random Binary Sequence (PRBS) is injected at interface F (or E). The various C/N ratios are established at the input of the test receiver by addition of Gaussian noise, and the BER of the received PRBS is measured at point V (or U) using a BER TEST Set. The measurement is repeated for a range of mean transmitted output power. If the ability to generate a PRBS at interface F (or E) is included in the transmitting equipment for test purposes, then it should be a 223 - 1 PRBS as defined by Recommendation ITU-T O.151 [i.12]. For the measurement of carrier and noise power, the system bandwidth is defined as n × fSPACING, where n is the number of active carriers (e.g. 6 817 or 1 705 carriers in an 8 MHz channel) and fSPACING is the frequency spacing of the OFDM carriers. NOTE: Transmitter back-off is defined as the ratio of the rated pulsed peak power of the transmitter to the mean power of the signal. The rated pulsed peak power is normally equivalent to the peak sync power of a standard B, D, G, H, I or K RF signal. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 62 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.14 BER vs. C/N ratio by variation of Gaussian noise power | Purpose To evaluate the BER performance of a receiver as the Carrier to Noise (C/N) ratio is varied by changing the added Gaussian noise power. This measurement can be used to compare the performance of a receiver with theory or with other receivers. For example to evaluate the influence of receiver noise floor. Interface From F to U or from E to V. Method A Pseudo-Random Binary Sequence (PRBS) is injected at interface F (or E). Various C/N ratios are established at the input of the receiver under test by addition of Gaussian noise and the BER of the received PRBS is measured at point V (or U) using a BER test set. A test transmitter should be able to generate the 223 - 1 PRBS as defined by Recommendation ITU-T O.151 [i.12]. For the measurement of carrier and noise power, the system bandwidth is defined as n × fSPACING where n is the number of active carriers i. e. 6 817 or 1 705 carriers and fSPACING is the frequency spacing of the OFDM carriers. NOTE: The bandwidth in an 8 MHz channel is approx. 7,61 MHz, in a 7 MHz channel system it is 6,66 MHz and 5,71 MHz in a 6 MHz channel. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.15 BER before Viterbi (inner) decoder | Purpose This measurement gives an in-service indication of the un-coded performance of the transmitter, channel and receiver. Interface V. Method The signal after Viterbi decoding in the test receiver is coded again using the same convolutional coding scheme as in the transmitter in order to produce an estimate of the originally coded data stream. This data stream is compared at bit-level with the signal which is available before Viterbi decoder. The measurement should be based on at least several hundred bit errors. Viterbi Decoder Delay Convolutional Coder Outer de-interleaver Comparison BER V W X Figure 9.4: BER measurement before Viterbi decoding |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.16 BER before RS (outer) decoder | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.16.0 General | Purpose The BER is the primary parameter which describes the quality of the digital transmission link. Interface W or X Method The BER is defined as the ratio between erroneous bits and the total number of transmitted bits. Two alternative methods are available; one for "Out of Service" and a second for "In Service" use. In both cases, the measurement should only be done within the Link Available Time (LAT) as defined in clause 6.2. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 63 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.16.1 Out of Service | The basic principle of this measurement is to generate within the channel encoder a known, fixed, repeating sequence of bits, essentially of a Pseudo-Random nature. In order to do this the data entering the sync-inversion/randomization function is a continuous repetition of one fixed TS packet. This sequence is defined as the null TS packet in ISO/IEC 13818-1 [i.1] with all data bytes set to 0x00; i.e. the fixed packet is defined as the four byte sequence 0x47, 0x1F, 0xFF, 0x10, followed by 184 zero bytes (0x00). Ideally this would be available as an encoding system option. The apparently obvious alternative of injecting a PRBS in the transmitter at the output of the RS encoder is not used because of the requirement to have sync bytes to ensure correct operation of the byte interleaver. Insertion after the byte interleaver is not appropriate because it is not then directly comparable with the in-service measurement. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.16.2 In Service | The basic assumption made in this measurement method is that the RS check bytes are computed for each link in the transmission chain. Under normal operational circumstances, the RS decoder will correct all errors and produce an error-free TS packet. If there are severe error-bursts, the RS decoding algorithm may be overloaded, and be unable to correct the packet. In this case the transport_error_indicator bit should be set, no other bits in the packet should be changed, and the 16 RS check bytes should be recalculated accordingly before re-transmission on to another link. The BER measured at any point in the transmission chain is then the BER for that particular link only. The number of erroneous bits within a TS packet will be estimated by comparing the bit pattern of this TS packet before and after RS decoding. If the measured value of BER exceeds 10-3 then the measurement should be regarded as unreliable due to the limits of the RS decoding algorithm. Any TS packet that the RS decoder is unable to correct should cause the calculation to be restarted. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.17 BER after RS (outer) decoder (Bit error count) | Purpose To gain information about the pattern with which bit errors occur. Interface Z Method The same principle as used for the "Out of service" measurement of the "BER before the RS decoder" described in clause 9.16.1, with the modification that the result is presented as an error count rather than a ratio. The receiver only has to compare the received TS packets with the Null packets as defined in clause A.1.2. This method is applicable for cases where the BER before RS decoder is lower than approximatively 10-3. This can be used as one parameter for the estimation of the quality of the transmission link as it was defined by the operator, or for localization of specific problems. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.18 IQ signal analysis | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.18.1 Introduction | The IQ analysis can be applied on single carriers of the OFDM signal as well as on groups of carriers. If groups of carriers are under consideration all received symbols of this group can be superimposed in order to get one common constellation diagram. Since the scattered pilot carriers, the continual pilot carriers and the TPS carriers are transmitted in a different modulation scheme it is recommended to exclude these carriers from the IQ analysis or apply a specific IQ analysis. Assuming: - a constellation diagram of M symbol points and K carriers under consideration with 0 < K ≤ KMAX + 1 and KMAX + 1 is the total number of active OFDM carriers (i.e. 1 705 or 6 817 carriers); - a measurement sample of N data points, where N is sufficiently larger than M × K to deliver the wanted measurement accuracy; and ETSI ETSI TR 101 290 V1.4.1 (2020-06) 64 - the co-ordinates of each received data point j being Ij + δIj, Qj + δQj where I and Q are the co-ordinates of the ideal symbol point and δI and δQ are the offsets forming the error vector of the data point (as long as the respective carrier is a "useful" one). The following six parameters can be calculated, which give an in-depth analysis of different influences, all deteriorating the signal. Modulation Error Ratio (MER) and the related Error Vector Magnitude (EVM) are calculated from all N data points without special pre-calculation for the data belonging to the M symbol points. With the aim of separating individual influences from the received data, for each point i of the M symbol points the mean distance di and the distribution σi can be calculated from those δIj, δQj belonging to the point i. From the M values {d1, d2, ... dM} the influences/parameters: - Origin offset/Carrier suppression (CS); - Amplitude Imbalance; and - Quadrature Error (QE) (only for 2 k modes since the centre carrier needs to carry a complete constellation which is not the case in an 8k system where the centre carrier is a continual pilot) can be extracted and removed from the di values, allowing to calculate the Residual Target Error (RTE) with the same algorithm as the System Target Error (STE) from {d1, d2, ... dM}. From the statistical distribution of the M clouds the parameters: - Phase Jitter (PJ); and - coherent interferer (if it is dominant) may be extracted. The remaining clouds (after elimination of the above two influences) are assumed to be due to Gaussian noise only and are the basis for calculation of the signal-to-noise ratio. The parameter may include - besides noise - also some other disturbing effects, like small coherent interferers or residual errors from the channel correction. When using the interfaces S or T filtering of the signal before the interface should be considered. The parameters Origin offset/Carrier suppression (CS), Amplitude Imbalance (AI) and Quadrature Error (QE) are typical performance parameters of the modulator. The other parameters are also influenced by the transmission system and the receiver/demodulator. It should be noted that the channel estimation/channel correction mechanism can have an impact on the measurement results. This is particularly true for measurements in the field or under simulated but realistic reception conditions. For measurements taken at the output of a transmitter this impact of the channel estimation/channel correction mechanism is negligible. For comparison of measurement results, information on the character of the channel estimation/channel correction mechanism should be provided. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 65 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.18.2 Modulation Error Ratio (MER) | Purpose To provide a single "figure of merit" analysis of the K carriers. Interface S, T and H Method The carrier frequency of the OFDM signal and the symbol timing are recovered. Origin offset of the centre carrier (e.g. caused by residual carrier or DC offset), Quadrature Error (QE) and Amplitude Imbalance are not corrected. A time record of N received symbol co-ordinate pairs ( ) j j Q I ~ , ~ is captured. For each received symbol, a decision is made as to which symbol was transmitted. The error vector is defined as the distance from the ideal position of the chosen symbol (the centre of the decision box) to the actual position of the received symbol. This distance can be expressed as a vector ( ) j j Q I δ δ , . The sum of the squares of the magnitudes of the ideal symbol vectors is divided by the sum of the squares of the magnitudes of the symbol error vectors. The result, expressed as a power ratio in dB, is defined as the MER. ( ) ( ) dB Q I Q I MER N j j j N j j j + + × = = = 1 2 2 1 2 2 10 log 10 δ δ It should be reconsider that MER is just one way of computing a "figure of merit" for a vector modulated signal. Another "figure of merit" calculation is Error Vector Magnitude (EVM) defined in annex C of the present document. It is also shown in annex C that MER and EVM are closely related and that one can generally be computed from the other. MER is the preferred first choice for various reasons itemized in annex C of the present document. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.18.3 System Target Error (STE) (void) | Figure 9.5: Definition of Target Error Vector (TEV) (void) |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.18.4 Carrier Suppression (CS) | Purpose A residual carrier is an unwanted coherent signal added to the centre carrier of the OFDM signal. It may have been produced by dc offset voltages of the modulating I and/or Q signal or by crosstalk from the modulating carrier within the modulator. Interface S and T. Method Search for systematic deviations of all constellation points of the centre carrier and isolate the residual carrier. Calculate the Carrier Suppression (CS) from the formula: × = RC sig P P CS 10 log 10 where PRC is the power of the residual carrier and Psig is the power of the centre carrier of the OFDM signal (without residual carrier). NOTE: Not applicable for 8k modes (see clause 9.18.1). ETSI ETSI TR 101 290 V1.4.1 (2020-06) 66 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.18.5 Amplitude Imbalance (AI) | Purpose To separate the QAM distortions resulting from Amplitude Imbalance (AI) of the I and Q signal from all other kind of distortions. Interface S and T. Method Calculate the I and Q gain values vI and vQ from all points in a constellation diagram eliminating all other influences. Calculate Amplitude Imbalance (AI) from vI and vQ. NOTE 1: Since the allocation of I and Q to the axis in the complex plane is unambiguous for a DVB-T signal, the parameter AI can convey the information which component dominates. Therefore, this definition differs slightly from the one given in clause 6.9.5. ν ν I Q I Q Q I Q I v v v v v v if if AI > × − ≥ × − = % 100 1 % 100 1 { ( ) ( ) ( ) ( ) ( ) ( ) i Q i I i N j j Q i M i i Q i i Q N j j I i M i i I i i I d d d Q N d Q d Q M I N d I d I M = + = + = = + = = = = = 9.18.3) subclause in given as d of component - (Q 1 1 9.18.3) subclause in given as d of component - (I 1 1 i 1 1 i 1 1 δ ν δ ν NOTE 2: Not applicable for 8k modes (see clause 9.18.1). ETSI ETSI TR 101 290 V1.4.1 (2020-06) 67 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.18.6 Quadrature Error (QE) | Purpose The phases of the two carriers feeding the I and Q modulators have to be orthogonal. If their phase difference is not 90 a typical distortion of the constellation diagram results. It is assumed that the value derived from the centre carrier is representative for the whole signal. Interface S and T. Method Search for the constellation diagram error shown in figure 9.6 and calculate the value of the phase difference Δϕ = ϕ1 - ϕ2 after having eliminated all other influences and convert this into degrees: ( ) [ ]° − × ° = ϕ ϕ π 2 1 180 QE I Q Decision Boundary Signal Point Decision Boundary Box 1 ϕ 2 ϕ 90°+QE Figure 9.6: Distortion of constellation diagram resulting from I/Q Quadrature Error (QE) NOTE: Not applicable for 8k modes (see clause 9.18.1). ETSI ETSI TR 101 290 V1.4.1 (2020-06) 68 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.18.7 Phase Jitter (PJ) | Purpose The PJ of an oscillator is due to fluctuations of its phase or frequency. Using such an oscillator to modulate a digital signal results in a sampling uncertainty in the receiver, because the carrier regeneration cannot follow the phase fluctuations. The signal points are arranged along a curved line crossing the centre of each decision boundary box as shown in figure 9.7 for the four "Corner Decision Boundary Boxes". Q I "Corner Decision Boundary Box" for calculation of the Phase Jitter Arc section through a Figure 9.7: Position of "Arc section" in the constellation diagram to define PJ (example: 64-QAM) Interface S and T. Method Phase Jitter can be calculated theoretically using the following algorithm: 1) Calculate the angle between the I-axis of the constellation and the vector to the received symbol ) , ( rcvd rcvd Q I : rcvd rcvd I Q arctan 1 = φ 2) Calculate the angle between the I-axis of the constellation and the vector to the corresponding ideal symbol ) , ( ideal ideal Q I : ideal ideal I Q arctan 2 = φ Phi 2 instead of Phi 1 3) Calculate the error angle: 2 1 φ φ φ − = E 4) From these N error angles calculate the RMS phase jitter: = = − = N i N i E E i i N N PJ 1 2 1 2 2 1 1 φ φ However, the following method may be more practical: The first approximation of the "Arc Section" of a "Corner Decision Boundary Box" is a straight line parallel to the diagonal of the "Decision Boundary Box". Additionally the curvature of the Phase Jitter (PJ) trace has to be taken into account when calculating the standard deviation of the PJ. The mean value of the PJ is calculated in degrees. ( ) × − × × ° = d M PJ PJ 1 2 arcsin 180 σ π [°] where M = Order of QAM and 2d = Distance between two successive boundary lines Within the argument of the arc sine function, the standard deviation of the Phase Jitter is referenced to the distance from the centre of the "Corner Decision Boundary Box" to the centre point of the QAM signal. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 69 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.19 Overall signal delay | Purpose To measure and adjust the signal delay of an OFDM transmitter to a given value so that the transmitters in an SFN can be synchronized. Interface A, M. Method (a) The total delay between the MPEG TS input of the transmitter under test and the MPEG TS output of a test receiver is established by measuring the time delay required to match the input and output data patterns. If the delay of the test receiver is known then the transmitter signal delay can be derived. Alternatively, the delay of the test receiver could be expressed relative to the delay of a reference receiver. This would avoid the need to measure the absolute delay of any receiver. (b) A more direct method may be to define a transmitter test mode in which the occurrence of a Mega-frame Initialization Packet (MIP) at the MPEG TS input causes a trigger pulse (see ETSI TS 101 191 [i.14]). The trigger pulse is made available for connection to an oscilloscope and also used to "arm" the modulator. At the start of the next mega-frame the modulator transmits a null symbol (or a defined pulse in the time domain) rather than the normal data. The delay between the trigger pulse and the RF null (or pulse) is measured. (c) The delay of a transmitter could be expressed relative to the delay of a reference transmitter. For the measurement a reduced amplitude sample is taken from both transmitters and adjusted to have similar level (< 3 dB difference), the samples are combined in a RF linear adder and the output is fed to a spectrum analyser. Typically the spectrum formed will have lobes due to the difference of delays in the two transmitters. The inverse of the frequency width of the lobes represents the relative delay between the transmitters. Two drawbacks has to be taken in account: 1) the delay is absolute, that is, it gives no indication of which transmitter has the longer delay; 2) the accuracy is related to the ability of identifying the minimal values of the lobes and the accuracy of the measurement. NOTE 1: The delay of a transmitter may be considered as the addition of various parts including the physical delays of the analogue part of the OFDM signal, including the path length to the antenna. Also the buffers used for signal conditioning (TS bitrate adaptation to the sampling frequency of the transmitter) and other intermediate buffers in the OFDM spectrum calculation (IFFT) may differ from manufacturer to manufacturer. NOTE 2: In cases of single frequency networks, the SFN adapter at the transmitter site may be considered as integral part of the modulator transmitter. It may calculate the delay, from the value of the STS (Synchronization Time Stamp) to the 1 pps used as reference, in different way from manufacturer to manufacturer and add differences in the delays that have to be included in the measurement result. It is recommended to use a test Transport Stream with embedded MIP data, and real-time calculation of the STS. See clause E.16 for test set-up, measurement description and example of results. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 70 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.20 SFN synchronization | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.20.1 MIP_timing_error | Purpose A necessary precondition for SFN synchronization is that the Synchronization Time Stamp (STS) values inserted in the Mega-frame Initialization Packet (MIP) are correct. This test checks that successive STS values are self-consistent. See ETSI TS 101 191 [i.14]. Interface A, Z (especially Transport Stream between the "SFN adapter" and "SYNC system" as defined in [i.14]). Method Locate the MIP in three successive mega-frames numbered M, M+1 and M+2. Extract the synchronization_time_stamp field from each MIP (STSM, STSM+1 and STSM+2). In general, the difference between any two consecutive STS values will be the duration of one mega- frame minus some multiple (including zero) of the time between GPS pulses. Even without knowing the precise duration of the mega-frame, the duration is constant and the following can be derived: STSM+2 - STSM+1 = STSM+1 - STSM + nT where T is 1s and n is any integer. Calculate nT from the above formula and check it is an integral number of seconds to within a user defined accuracy. This test can be performed continually on each successive set of 3 mega-frames, {M+1, M+2, M+3}, {M+2, M+3, M+4} etc. The test result should be discarded if the mega-frame size changes over the set of three mega-frames. NOTE: The mega-frame size changes, for example, with the change of the DVB-T transmission mode. This would normally result in a resynchronization. NOTE: Figure 9.8 is an illustration of the timing relationship between mega-frames and the GPS one second pulses. This shows how the synchronization_time_stamp (STS) is calculated. Consider STSM+1 and STSM+2. In this case it is quite clear that: STSM+2 - STSM+1 = duration of one mega-frame In the case of STSM and STSM+1, a 1s pulse has passed by and the equivalent equation is: (STSM+1 + 1) - STSM = duration of one mega-frame GPS 1s pulses Mega- frame M I P M I P M I P M I P M I P STS values 1s 2s 3s 4s STSM STSM+1 STSM+2 STSM+3 STSM+4 M M+1 M+2 M+3 M+4 Figure 9.8: Megaframe/GPS pulse timing relationship ETSI ETSI TR 101 290 V1.4.1 (2020-06) 71 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.20.2 MIP_structure_error | Purpose This test verifies that the syntax of the MIP complies with the specification in ETSI TS 101 191 [i.14]. Interface A, Z Method For each transport packet carried on PID 0x15 in the transport stream, the following checks are performed: The transport_packet_header should comply with ETSI TS 101 191 [i.14] clause 6, table 1, and ISO/IEC 13818-1 [i.1] clause 2.4.3.2, tables 2 and 3. All length fields should be consistent to provide a proper length packet. This includes section_length (which also should not exceed 182), individual_addressing_length (which should match the length of the loops for each transmitter), function_loop_length (which should match the sum of the size of each of the functions), function_length (which should match the proper length of the function based upon the function tag). The synchronization_time_stamp and the maximum_delay should be in the range of 0x0 to 0x98967F. The CRC_32 field should match the CRC calculated for the MIP data. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.20.3 MIP_presence_error | Purpose This test verifies that the MIP is inserted into the transport stream only once per mega- frame. Interface A, Z Method The following checks are performed: Extra MIP – For every MIPN (where N > 1), signal an error if it arrives within the number of packets indicated by the pointer field of MIPN-1. Missing MIP - For each MIP received, calculate the mega-frame size from the parameters in the tps_mip. The latest two values of the mega-frame size are stored. After every MIPN is received (where N > 1), signal an error if a MIPN+1 is not received before K + R packets are received after MIPN, where K is the pointer value of MIPN and R is mega-frame size in packets from the previous MIPN-1. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.20.4 MIP_pointer_error | Purpose The MIP insertion can be at any location in the mega-frame. If the insertion is periodic as defined in the MIP, the MIP location in the mega-frame is constant over time. The MIP can be used to determine the mega-frame size and where each mega-frame starts and ends in the transport stream thanks to the pointer field verified by this test. Interface A, Z Method For each MIP received, calculate the mega-frame size from the parameters in the tps_mip. The latest three values of the mega-frame size are stored. For every MIPN that is received (where N > 2), signal an error if the pointer value (PN) of MIPN does not hold in the following equation: PN = PN-1 + MFN-2 - (iN - iN-1) Where MFN-2 is the size of the Nth mega-frame in packets but is calculated from MIPN-2, and iN is the packet index for MIPN. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 72 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.20.5 MIP_periodicity_error | Purpose In the case of a periodic MIP insertion (as defined in ETSI TS 101 191 [i.14] clauses 5 and 6), the pointer value should remain constant, as well as the number of packets between each MIP. Interface A, Z Method The following checks are performed: Compare the current pointer field in MIPN with the pointer field in the MIPN-1. It is an error if they are different, unless the mega-frame size changed between N and N-1. The number of packets between each MIP (iN - iN-1) should also be constant unless the mega-frame size changes. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.20.6 MIP_ts_rate_error | Purpose In a SFN network the modulator settings are transmitted by the tps_mip (see ETSI TS 101 191 [i.14] clause 6, table 3). These settings determine the transmission mode and in this way the bit rate of the Transport Stream. This test verifies that the actual Transport Stream data rate is consistent with the DVB-T mode defined by the tps_mip. Interface A, Z Method For each MIP received, calculate the data rate of the transmission mode - given by tps_mip setting and compare it with the actual data rate of the Transport Stream. Signal an error if the following equation is correct: Max_deviation ≤ | TS_data_rate - [(IFFT_clock_freq × tpl /204 × c × m × (uc/tc))/(1 + g)] | Where: • Max_deviation e.g. 10 kb/s; maximum deviation between actual TS_data_rate and data rate of the transmission mode given by tps_mip. The value results from the smallest difference of TS data rates which can be determined by two correct tps_mip settings for different modes. • TS_data_rate actual data rate of the Transport Stream measured by a test instrument according to clause 5.3.3.2. • IFFT_clock_freq 64/7 MHz (for 8 MHz channel bandwidth), 64/8 MHz (for 7 MHz channel bandwidth) 48/7 MHz (for 6 MHz channel bandwidth) given by tps_mip P12 and P13 • tpl transport packet length 188 or 204 byte • c code rate 1/2, 2/3, 3/4, 5/6 or 7/8 given by tps_mip P5,P6 and P7 • m 2 (for QPSK), 4 (for 16 QAM) or 6 (for 64 QAM) given by tps_mip P0 and P1 • uc useful_carriers 1512 (for 2k), 6 048 (for 8k) given by tps_mip P10, P11 (see note) • tc total_carriers 2 048 (for 2k), 8 192 (for 8k) given by tps_mip P10, P11 (see note) • g guard interval 1/4, 1/8, 1/16 or 1/32 given by tps_mip P8, P9 NOTE: The term (uc/tc) can be replaced by a constant value since uc2k/tc2k = uc8k/tc8k. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 73 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 9.21 System Error Performance | Purpose: The System Error Performance describes the performance of the digital transmission from the input of the MPEG-2 TS signal into the DVB Baseline system to the MPEG-2 TS output of this Baseline system. Interfaces: A, Z, M: with reference receiver (e.g. Transmitter measurement). N: with reference receiver (e.g. coverage measurements). Method: The measurement of System Error Performance is based on a subset of the error events defined in clause 5.4: • Errored Second (ES) or Errored Time Interval (ETI), • Severely Errored Second (SES) or Severely Errored Time Interval (SETI). The used time interval T for identification of these events depends on the aim of the measurement. Time intervals longer or shorter than 1 second may be considered appropriate in certain circumstances. Evaluation of Error Performance Parameters Error performance should only be evaluated whilst the transmission is in the available state (see also clause 6.1). To evaluate error performance parameters from events, a certain measurement interval (MI) has to be used. This measurement interval depends on the specific aim of the measurement. Possible measurement intervals corresponding to special applications are proposed in table 9.4. In general the error performance is the ratio of number of true events to the total number of time intervals T during the measurement interval. Consequently derived performance parameters are: • Errored Second Ratio (ESR) or Errored Time Interval Ratio (ETIR); • Severely Errored Second Ratio (SESR) or Severely Errored Time Interval Ratio (SETIR). Table 9.4: Examples of Measurement Intervals MI Length of Measurement Interval (MI) Application 5 s - applicable for analysis of mobile reception 20 s - Coverage Check - recommended minimum measurement interval for receiver comparison 5 minutes - possible resolution for 1 hour analysis. 1 hour - possible resolution for daily fluctuations analysis 10 Recommendations for the measurement of delays in DVB systems Void. 11 Measurements for the second generation terrestrial (DVB-T2) system |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.1 Introduction | The DVB-T2 system as it is addressed in the following clauses, spans from the input Interface A to the output Interface D. Both these interfaces carry MPEG2 Transport Streams ("TS"). ETSI ETSI TR 101 290 V1.4.1 (2020-06) 74 Figure 11.1: Block diagram of a typical DVB-T2 chain [i.27] The following clauses specify a number of tests and measurements at the interfaces A, B, C, Cn and D. For measurements at Interface A and D, see clause 5 for the parameters measured at the other interfaces see clauses 11.2 and 11.3. The Interfaces Cn are introduced in addition to the interfaces defined in ETSI EN 302 755 [i.27] to accommodate all specified measurements for DVB-T2. With regard to the signalling, it is recommended that a measurement instrument should display the signalled information as readable text and abbreviations. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2 Measurements at the DVB-T2 Modulator Interface (T2-MI) | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.1 Introduction | The DVB-T2 Modulator Interface specification ETSI TS 102 773 [i.24] defines the format of T2-MI packets. The T2-MI format enables the operation of Single Frequency Networks SFNs by taking the scheduling decisions at the central point of the T2 gateway. The output from the T2 Gateway is forwarded to all modulators in the SFN so that all modulators receive identical information. For more details refer to the T2-MI standard [i.24] Interfaces Cn ETSI ETSI TR 101 290 V1.4.1 (2020-06) 75 Figure 11.2: The T2-MI protocol stack [i.24] Figure 11.3 T2-MI packet format [i.24] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.2 Measurements of the syntax of T2 MI packets | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.2.1 T2MI_packet_type_error_1 | Purpose Two of the various packet types are mandatory for each T2 frame: 1016 (L1-current data), 2016 (DVB-T2 Timestamp). If L1 repetition, in-band signalling (IBS) or Time-Frequency-Slicing (TFS) is indicated in the L1-current data, an L1-future packet (1116) should also be present. If any of these packet_type is not present in each T2 frame, a T2MI_packet_type_error_1 is signalled. Interface Interface B "T2-MI" Method Comparison of the decoded packet_type value with the list of mandatory values. Reference Clauses 5.1 and 5.4 of ETSI TS 102 773 [i.24] ETSI ETSI TR 101 290 V1.4.1 (2020-06) 76 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.2.2 T2MI_packet_type_error_2 | Purpose The number of BB-frames (packet_type 0016) relating to a PLP in a given T2 frame should match the value of PLP_NUM_BLOCKS signalled in the dynamic signalling of both the L1-current and L1-future (when present). The signalled values of frame_idx and superframe_idx for BB-frame packets should be consistent with the time interleaver parameters specified in the configurable signalling of the L1-current. NOTE: • BB-frames are only mandatory for a given PLP if they are signalled in the L1-current data. • In a case where the Interleaving Frame spans more than one T2-frame (PI>1), there is a possibility that there will be no BB frames present with certain frame_idx values since the frame_idx always refers to the first T2 frame of the Interleaving Frame. • This could also occur if the bit-rate for a given PLP falls to zero in a given T2 frame. Interface Interface B "T2-MI" Method Comparison of the number of BB frames and their values of frame_idx and superframe_idx with the information from the L1 signalling. Reference Clauses 5.1 and 5.4 of ETSI TS 102 773 [i.24] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.2.3 T2MI_packet_count_error | Purpose This error indicates a discontinuity of T2-MI packets Interface Interface B "T2-MI" Method Comparison of the decoded packet_count value of the received T2-MI packet with the packet_count value of the previous packet. The case of receiving the first packet of a transmission, for which no specific packet_count value is required, needs to be considered. Reference Clause 5.1 of ETSI TS 102 773 [i.24] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.2.4 T2MI_CRC_error | Purpose The CRC32 check indicates if the content of the respective T2-MI packet is corrupted. It is calculated across all other bits in the packet (both header and payload plus any padding). Interface Interface B "T2-MI" Method According to annex A of ETSI TS 102 773 [i.24] Reference Clause 5.1 of ETSI TS 102 773 [i.24] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.2.5 T2MI_payload_error | Purpose The T2MI_payload_error is signalled if the decoded plp_id in T2-MI packets with packet_type 0016 is not included in the list (L1 post_signalling/configurable) of plp_id for the T2-MI signal. Interface Interface B "T2-MI" Method Comparison of the decoded plp_id of T2-MI packets with packet_type 0016 with list of possible values. Reference Clause 5.2.1 of ETSI TS 102 773 [i.24] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.2.6 T2MI_plp_num_blocks_error | Purpose The number of FEC blocks in an Interleaving Frame for a PLP as signalled in the Dynamic L1-post signalling should be consistent with the number of BB frame packets. Interface Interface B "T2-MI" Method This error indication is set if the number of received BB frame packets does not match the signalled value. Reference Clause 7.2.3.2 of ETSI EN 302 755 [i.27] and clause 5.2.1 of ETSI TS 102 773 [i.24] ETSI ETSI TR 101 290 V1.4.1 (2020-06) 77 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.2.7 T2MI_transmission_order_error | Purpose The T2MI_transmission_order_error is signalled if the packet_types are in a wrong ordering and position inside a T2 frame. Interface Interface B "T2-MI" Method The required ordering is DVB-T2 Timestamp (2016) -> P2 bias balancing cells (1216, if present) -> L1_Current (1016) -> L1_future (1116, if present) -> change of frame_idx Reference Clause 5.4 of ETSI TS 102 773 [i.24] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.2.8 T2MI_DVB-T2_Timestamp_error | Purpose The T2MI_DVB-T2_Timestamp_error signals a wrong timestamp inside a superframe Interface Interface B "T2-MI" Method The T2MI_DVB-T2_Timestamp_error is signalled if at least one T2_timestamp has a different value than the other DVB-T2 timestamp inside a single superframe_(superframe_idx). Reference Clause 5.2.7.1 of ETSI TS 102 773 [i.24] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.2.9 T2MI_DVB-T2_Timestamp_discontinuity | Purpose The T2MI_DVB-T2_Timestamp_discontinuity signals a non-increasing timestamp (not relevant with null timestamps) Interface Interface B "T2-MI" Method The T2-MI timestamps are compared for subsequent superframes and the error is indicated if the difference is not equal to the duration of the superframe. NOTE: Timestamp_discontinuity should take into account a time window of what should be the next time stamp and the received timestamp. Reference Clause 5.7 of ETSI TS 102 773 [i.24] 11.2.2.10 T2MI_T2_frame_length_error Purpose The T2 frame length derived from parameters signalled in L1 is not longer than 250 ms. Interface Interface B "T2-MI" Method The error is indicated if the T2 frame length derived from L1 signalling parameters is over 250 ms. Reference Clause 5.2.4 of ETSI TS 102 773 [i.24] and clause 7.2 of ETSI EN 302 755 [i.27] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.3 Checks on the T2-MI MIP (Modulator Information Packet) | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.3.1 T2MI_MIP_timestamp_error | Purpose The value of the timestamp of the T2-MI MIP (t2_timestamp_mip) with PID = 1516 should not be lower than the value of the DVB-T2 timestamp. Interface Interface B "T2-MI" Method The error is indicated if the timestamp of the T2-MI MIP has a lower value than the value of the DVB-T2 timestamp. Reference Annex B of ETSI TS 102 773 [i.24] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.3.2 T2MI_MIP_individual_addressing_error | Purpose The consistency of the data contained in the individual_addressing_byte is checked against the bytes of the individual_addressing_data field of a T2-MI packet of type 2116. Interface Interface B "T2-MI" Method The error is indicated if the comparison shows inconsistencies. Reference Clauses B.2.1 and 5.2.8 of ETSI TS 102 773 [i.24] ETSI ETSI TR 101 290 V1.4.1 (2020-06) 78 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.3.3 T2MI_MIP_continuity_error | Purpose The insertion of at least one complete T2-MIP within a T2 superframe is required. Interface Interface B "T2-MI" Method The error is indicated if the usage of MIP is signalled, and a superframe without a complete T2-MIP packet is found. Reference Clause B.2.2 of ETSI TS 102 773 [i.24] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.3.4 T2MI_MIP_CRC_error | Purpose The CRC32 bits of the T2-MIP are checked to establish that the T2-MIP packet is uncorrupted. Interface Interface B "T2-MI" Method The error is indicated if the CRC check shows that the T2-MIP packet is corrupted. Reference Clause B.2.1 of ETSI TS 102 773 [i.24] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.4 Check on consistency of T2-MI signalling information | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.4.1 T2MI_bandwidth_consistency_error | Purpose The bandwidth signalled in the T2-MI DVB-T2 Timestamp and the determining parameters signalled in L1 (i.e. fft mode, guard interval, pilot pattern, number of OFDM data symbols and PLP specific code rate, modulation, fec type, num_blocks, baseband mode, issy information, null packet deletion, in-band signalling, num other plp inband) should be such that the stream can be transmitted in the pertaining channel. Interface Interface B "T2-MI" Method The error is indicated if the maximum possible bit rate which is calculated from the T2-MI DVB-T2 Timestamp and the determining L1 parameters is lower than the bit rate of the stream encapsulated in the T2-MI packets. Reference Clause 5.2.7 of ETSI TS 102 773 [i.24] and clause 7.2 of ETSI EN 302 755 [i.27] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.4.2 T2MI_DVB-T2_Timestamp_leap_second_error | Purpose The T2MI_DVB-T2_Timestamp_leap_second_error signals an error in leap second value signalled in utco field in T2 Timestamp. Interface Interface B "T2-MI" Method Leap second value in T2-MI timestamp is compared to known leap second value as published by IERS (International Earth Rotation Service). Reference Clause 5.2.7 of ETSI TS 102 773 [i.24] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.5 Measurements at T2-MI transport layer | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.5.1 Encapsulation of T2-MI packets into MPEG-2 TS (ASI) streams | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.5.1.0 General | T2-MI packets are encapsulated into 188-bytes MPEG-2 TS packets according to "Data Piping" mechanism (ETSI EN 301 192 [i.33], clause 4). This allows to reuse existing telecom links based on ASI. Figure 11.4: Encapsulation of T2-MI packets into MPEG-2 TS (ASI) ETSI ETSI TR 101 290 V1.4.1 (2020-06) 79 The addition of a minimum of PSI (PAT and PMT) is suggested, to help detecting alarms on the ASI link. The "classic" Reed-Solomon (188,204) FEC can be (optionally) adopted. 11.2.5.1.1 Parameters relevant to transport of T2-MI packets over MPEG-2 TS Measurements of the MPEG-2 TS stream carrying T2-MI packets can be done according to Clause 5 of the present document. However, the TS in this case is only used as a container for the T2-MI packets according to data piping mechanism, and therefore only a subset of the parameters are relevant for the measurements. 11.2.5.1.2 Informative parameters Parameter Notes Reference Total TS bit-rate The total bit-rate of the TS carrying the T2-MI data includes the TS header and PAT, PMT, stuffing, etc. The default bit-rate measurement profile should be MGB1. For special cases, MGB5 with a user- defined measurement interval should be used. Clause 5.3.3.2 T2-MI bit-rate Bit-rate corresponding to the PID carrying the T2-MI packets (data piping). The default bit-rate measurement profile should be the net TS bit-rate without header and without Adaptation Field. RS(188,204) FEC Present/Absent 11.2.5.1.3 Integrity parameters Subset of relevant 1st priority parameters, with reference to clause 5.2.1: No. Indicator Notes 1.1 TS_sync_loss 1.2 Sync_byte_error 1.3/1.3a PAT_error PAT is suggested but not mandatory in ETSI TS 102 773 [i.24] 1.4 Continuity_count_error 1.5/1.5a PMT_error PMT is suggested but not mandatory in ETSI TS 102 773 [i.24] 1.6 PID_error Since the T2-MI over TS is considered as a data service, a user-defined repetition rate of 5 s or more may be suitable. Subset of relevant 2nd priority parameters, with reference to clause 5.2.2: No. Indicator Notes 2.1 Transport_error It should be possible to disable the indication of the Transport_error if the T2-MI transport system does not support this function. 2.3 PCR_error If signalled according to ETSI TS 102 773 [i.24], annex G 2.4 PCR_accuracy_error If signalled according to ETSI TS 102 773 [i.24], annex G ETSI ETSI TR 101 290 V1.4.1 (2020-06) 80 Subset of relevant 3rd priority parameters, with reference to clause 5.2.3: No. Indicator Notes 3.4/3.4a Unreferenced_PID Relevant if PMT is present |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.2.5.2 Encapsulation of T2-MI packets into IP streams | 11.2.5.2.1 Parameters relevant to transport of T2-MI packets over IP 11.2.5.2.1.0 General MPEG-2 TS packets carrying the T2-MI data are encapsulated into IP using the RTP/UDP/IP protocol stack, according to DVB-IPTV standard (ETSI TS 102 034 [i.25]). A number of TS packets can be allocated per IP packet, limited by the maximum size of the IP datagram. Care should be taken not to exceed the underlying maximum transmission unit (MTU) of the network, in order to avoid fragmentation. 11.2.5.2.1.1 Informative parameters Parameter Explanation and notes Reference Source IP address Source address of the IP stream carrying the T2-MI packets IETF RFC 791 [i.34] Source IP port Source port of the UDP stream carrying the T2-MI packets IETF RFC 768 [i.35] Destination IP address Destination address of the IP stream carrying the T2-MI packets. It is normally a multicast group IETF RFC 791 [i.34], IETF RFC 3171 [i.36] Destination IP port Destination port of the UDP stream carrying the T2-MI packets IETF RFC 768 [i.35] IP stream bit-rate Total bit-rate associated to the IP stream carrying the TS with T2-MI packets No. of TS packets per IP frame Number of TS packets encapsulated in one IP frame Clause 7.1.1 of ETSI TS 102 034 [i.25] FEC type Type of FEC applied to protect the T2-MI stream over the IP linik. Possible values: 'None', 'SMPTE 2022-1', 'Raptor' Annex E of ETSI TS 102 034 [i.25] SMPTE FEC rows (L) Number of rows of the SMPTE 2022- 1 FEC matrix (if FEC is applied) Clause 8.1 of SMPTE 2022-1 [i.26] SMPTE FEC columns (D) Number of columns of the SMPTE 2022-1 FEC matrix (if FEC is applied) Clause 8.1 of SMPTE 2022-1 [i.26] SMPTE FEC IP port Destination port of the UDP stream carrying the SMPTE FEC packets (if FEC is applied). This port should be equal to destination IP port + 2 IETF RFC 768 [i.35], Clause 8.1 of SMPTE 2022-1 [i.26] SMPTE FEC bit-rate rate associated to the stream carrying SMPTE 2022-1 packets NOTE: Typical value for number of TS packets encapsulated in one IP frame over Ethernet is 7. In ISO/IEC 13818-1 [i.1] it is not required that this number keeps constant along the stream; however, a variation of this value is not compatible with SMPTE 2022-1 [i.26] FEC (and in fact such variation is not allowed by SMPTE 2022-2 [i.43]). 11.2.5.2.1.2 Integrity parameters 11.2.5.2.1.2.0 General • Media Loss Rate (MLR) or Lost IP frames • Corrected IP frames ETSI ETSI TR 101 290 V1.4.1 (2020-06) 81 • Delay Factor 11.2.5.2.1.2.1 Media Delivery Index - Media Loss Rate (MDI-MLR) Purpose Number of IP packets lost per second, after error recovery mechanisms, if any. This parameter is part of the MDI (Media Delivery Index) parameter. Error if > 0. Interface Interface B "T2-MI" – IP transport layer Method The Continuity Counter field of the inner MPEG-2 TS stream can be used. Under error free conditions, the sequence numbers increment. Counting any missing sequence numbers every second will produce the packets lost per second. Alternatively, the RTP layer information can be used as a supplement for the UDP protocol. RTP also uses sequence numbers and this number also increments with every packet. The same method as above, of counting missing sequence numbers may be used. Reference Clause 3.2 of IETF RFC 4445 [i.37] 11.2.5.2.1.2.2 Lost IP frames Purpose Alternative option to represent packet loss with respect to MLR. Cumulative number of IP packets lost per second, after error recovery mechanisms, if any. Error if > 0. Interface Interface B "T2-MI" – IP transport layer Method The Continuity Counter field of the inner MPEG-2 TS stream can be used. Under error free conditions, the sequence numbers increment. Counting any missing sequence numbers every second will produce the packets lost per second. Alternatively, the RTP layer information can be used as a supplement for the UDP protocol. RTP also uses sequence numbers and this number also increments with every packet. The same method as above, of counting missing sequence numbers may be used. Reference Clause 3.2 of IETF RFC 4445 [i.37] 11.2.5.2.1.2.3 Corrected IP frames Purpose The value of Lost IP frames + Corrected IP frames gives an indication of the capabilities of the FEC mechanism. Interface Interface B "T2-MI" – IP transport layer Method In case of FEC, number of IP packets corrected by the FEC mechanism. Reference ETSI ETSI TR 101 290 V1.4.1 (2020-06) 82 11.2.5.2.1.2.4 Media Delivery Index - Delay Factor (MDI-DF) Purpose Maximum difference, observed at the end of each media stream packet, between the arrival of media data and the drain of media data, over a calculation interval (typically 1 s). This assumes the drain rate is the nominal constant traffic rate for constant bit rate. The DF gives a good approximation of the needed buffer at receiving side to compensate for network jitter cumulated over the calculation interval. This parameter is part of the Media Delivery Index (MDI) parameter. Interface Interface B "T2-MI" – IP transport layer Method Consider a virtual buffer VB used to buffer received packets of a stream. When a packet P(i) arrives during a calculation interval, compute two VB values, VB(i,pre) and VB(i,post), defined as: • VB(i,pre) = sum (Sj) - MR × Ti; where j=1..i-1 • VB(i,post) = VB(i,pre) + Si where Sj is the media payload size of the jth packet, Ti is the relative time at which packet i arrives in the interval, and MR is the nominal media rate. VB(i,pre) is the Virtual Buffer size just before the arrival of P(i). VB(i,post) is the Virtual Buffer size just after the arrival of P(i). The initial condition of VB(0) = 0 is used at the beginning of each measurement interval. A measurement interval is defined from just after the time of arrival of the last packet during a nominal period (typically 1 second) to the time just after the arrival of the last packet of the next nominal period. During a measurement interval, if k packets are received, then there are 2 × k + 1 VB values used in deriving VB(max) and VB(min). After determining VB(max) and VB(min) from the 2k + 1 VB samples, DF for the measurement interval is computed and displayed as: • DF = [VB(max) - VB(min)]/MR Reference Clause 3.1 of IETF RFC 4445 [i.37] |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3 Measurements for DVB-T2 baseline system | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.0 General | This clause lists a number of measurements for the DVB-T2 baseline system. The parameters are mainly measured at interface C "DVB-T2" or at interface C1 "T2-IQ" as in figure 11.1. A list of the main application area of the DVB-T2 measurement parameters described in this clause is given in table 11.1. The measurements in clause 6.1 "System availability" (Interface D in a DVB-T2 system, figures 9.2) and clause 6.2 "Link availability" (Interface C5 in a DVB-T2 system, figure 9.2) are also applicable. Table 11.1: DVB-T2 measurement parameters and their application area Measurement parameter Transmitter Network Receiver In-service measure- ment |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.1 RF measurements | X X |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.1.1 RF frequency accuracy | X X X |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.1.2 RF occupied bandwidth | X X X |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.2 Selectivity | X |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.3 AFC capture range | X |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.4 Phase noise of Local Oscillators (LO) | X X 11.3.5 RF/IF signal power X X X X 11.3.6 MISO Group Power Ratio X X X X 11.3.7 Noise Power X X X X 11.3.8 RF and IF spectrum X X X 11.3.9 Receiver sensitivity/dynamic range for a Gaussian channel X 11.3.10 Linearity characterization (shoulder attenuation) X X 11.3.11 Power efficiency X X 11.3.12 PAPR effect X 11.3.13 P1 Symbol Error Rate X X 11.3.14 BER before LDPC (inner) decoder X X 11.3.15 Number of LDPC iterations X X ETSI ETSI TR 101 290 V1.4.1 (2020-06) 83 Measurement parameter Transmitter Network Receiver In-service measure- ment 11.3.16 BER before BCH (outer) decoder X X 11.3.17 Baseband Frame Error Rate BBFER X X 11.3.18 Errored Second Ratio ESR X X 11.3.19 IQ signal analysis X X X 11.3.19.2 Modulation Error Ratio (MER) X X X X 11.3.19.3 Signal to Interference Noise Ratio (SINR) X X X X 11.3.19.4 Carrier Suppression (CS) X X 11.3.19.5 Carrier Phase X X 11.3.19.6 Amplitude Imbalance (AI) X X 11.3.19.7 Quadrature Error (QE) X X 11.3.20 SFN synchronization X X 11.3.21 L1 signalling error X X X 11.3.22 RMS Delay-Spread (RMS-DS) X X X 11.3.23 Maximum Excess Delay (MED) X X X 11.3.24 Receiver Buffer Model (RBM) validation test X 11.3.25 Relative power Level during the non-P1 part of the FEF (RLF_non_P1) X X NOTE 1: The term 'In-service measurement' is understood as a measurement that does not require a specific test signal but can be carried out with a normal DVB-T2 signal. NOTE 2: As an In-service measurement in an unoccupied channel, the measurement of Noise Power can provide an overview of the man-made noise conditions in a certain channel or frequency band. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 84 Figure 11.5: Simplified block diagram of a DVB-T2 transmitter Figure 11.6: Simplified block diagram of a DVB-T2 receiver ETSI ETSI TR 101 290 V1.4.1 (2020-06) 85 11.3.1 RF measurements |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.1.0 General | The measurement of some basic parameters of the DVB-T2 OFDM signal may be carried out at the RF layer with a test receiver, a spectrum analyser or similar instruments. 11.3.1.1 RF frequency accuracy Purpose Successful processing of OFDM signals requires that certain carrier frequency accuracy be maintained at the transmitter. Specific network operations modes such as SFN require high accuracy of the carrier frequency. Interface C Method The measurement of the RF frequency accuracy determines the centre frequency of the signal, i.e. the positioning of the signal in the RF channel. a) Spectrum analyser method: the centre frequency is derived from the frequencies measured for the continual pilots and/or the edge pilots. b) Test receiver method: the centre frequency is derived from the digital samples after the test receiver has synchronized to the incoming DVB-T2 signal. In this case, the accuracy is typically expressed as 'Carrier Offset' and given in Hz or ppm. Reference Clauses 9.2.4, 9.2.5 of ETSI EN 302 755 [i.27]. NOTE: In any case, the usage of a high-precision reference frequency, e.g. 10 MHz, may be helpful. 11.3.1.2 RF occupied bandwidth Purpose The measurement of the occupied bandwidth allows the verification of the correct sampling frequency at the modulator. Interface C Method The occupied bandwidth is calculated from the measurements of the frequencies of the edge pilots and/or continual pilots of the DVB-T2 signal. a) Spectrum analyser method: if the frequency of the edge carriers is known then the related values for the occupied bandwidth may be calculated. Denoting the edge pilot frequencies as Fmin and Fmax the occupied bandwidth is appropriately OB = Fmax - Fmin+ 1/TU. b) Test receiver method: the occupied bandwidth is derived from the digital samples after the test receiver has synchronized to the incoming DVB-T2 signal. Reference Clause 9.2.5 of ETSI EN 302 755 [i.27]. 11.3.2 Selectivity Purpose To identify the capability of the receiver to reject out-of-channel interference. Interface The measurement of the signal input level and the interferer should be carried out at the interface C, using interface C3 or C4 for the BER monitoring. Method The input power is adjusted to 10 dB above the minimum input power as defined in "Receiver sensitivity" (see clause 11.3.9). The C/I threshold needed for QEF operation should be measured as a function of the frequency of a CW interferer. The failure point is defined as ESR5. ESR (Errored Second Ratio) is defined in clause 11.3.18. One errored second during a time interval of 20 seconds is given as ESR5 (5 % of the seconds are errored). Alternatively, the QEF point is used that is defined as BER after LDPC = 10-7. Since this measurement is time-consuming, the BER after LDPC = 10-4 is measured and the associated C/N value is increased by 0,2 dB. This corresponds to a BER after BCH = 10-11. Reference Clause 6.1.2 of ETSI EN 302 755 [i.27]. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 86 11.3.3 AFC capture range Purpose To determine the frequency range over which the receiver will acquire overall lock. Interface C for the application of the test signal; C1 or C2 for the test of receiver synchronization Method A signal is applied to the input of the receiver, at a level 10 dB above the minimum input power as defined in "Receiver sensitivity" (see clause 11.3.9). The signal is frequency shifted in steps (from below and above) towards a nominal value, whilst forcing the receiver to re-acquire after each step. Correct reception is assumed if after each step: a) the receiver can synchronize to the applied DVB-T2 signal, or if this is not indicated, b) the failure point is defined as ESR5. ESR (Errored Second Ratio) is defined in clause 11.3.18. One errored second during a time interval of 20 seconds is given as ESR5 (5 % of the seconds are errored). Alternatively, the QEF point is used that is defined as BER after LDPC = 10-7. Since this measurement is time-consuming, the BER after LDPC = 10-4 is measured and the associated C/N value is increased by 0,2 dB. This corresponds to a BER after BCH = 10-11. Reference Clause 5.1.7 of ETSI EN 302 755 [i.27]. 11.3.4 Phase noise of Local Oscillators (LO) Purpose Phase noise can be introduced at the transmitter, at any frequency converter or by the receiver. In an OFDM system the phase noise can cause Common Phase Error (CPE) which affects all carriers simultaneously, and which can be minimized or corrected by using the continual pilots. However the noise-like Inter-Carrier Interference (ICI) cannot be corrected. This measurement may be useful for manufacturing, incoming inspection and maintenance of modulators, transmitters, up/down converters and receivers, either professional or consumer type. Interface Any access to Local Oscillators (LO), in transmitters, converters and receivers. Method Phase noise can be measured with a spectrum analyser, a vector analyser or a phase noise test set. Reference n/a |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.5 RF/IF signal power | Purpose Signal power, or wanted power, measurement is required to set and check signal levels at the transmitter and receiver sites. Interface C Method The signal power of a DVB-T2 signal is defined as the mean power of the signal as would be measured with a thermal power sensor. In the case of received signals care should be taken to limit the measurement to the bandwidth at the wanted signal. When using a spectrum analyser or a calibrated receiver, it should integrate the signal power within the nominal bandwidth of the signal (n × fSPACING) where n is the number of carriers. Note that some spectrum analyser may not automatically compensate for the applied Resolution Bandwidth so that a manual correction may be applicable. Reference n/a |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.6 MISO Group Power Ratio | Purpose The MISO Group Power Ratio (MGPR) is required to check the presence of both MISO Groups within a network. Interface C Method The MGPR is defined as the RF signal power for MISO Group 1 divided by the RF signal power for MISO Group 2. NOTE: The value should be given in dB. The MGPR may be negative if the signals of MISO Group 2 are received with higher input level than the signals of MISO Group 1. The MGPR takes the value of ±∞in the presence of one MISO Group. For the display of the power levels of the paths of both MISO groups, the use of different colours for the figures or the diagram components (e.g. power level over delay) is recommended. Reference Clause 9.1 of ETSI EN 302 755 [i.27]. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 87 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.7 Noise power | Purpose Noise is a significant impairment in a transmission network. Interface C (RF or IF) Method The noise power (mean power), or unwanted power, can be measured with a spectrum analyser (out of service). The noise power is specified using the occupied bandwidth of the OFDM signal (n × fSPACING) where n is the number of carriers. NOTE: The Carrier-to-Noise ratio C/N should be calculated as the ratio of the signal power, measured as described in clause 11.3.5, to the noise power, measured as described in this clause. Reference n/a |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.8 RF and IF spectrum | Purpose To avoid interfering with other channels, the transmitted RF spectrum should comply with a spectrum mask, which is defined for the terrestrial network. If the spectrum at the modulator output is defined by a spectrum mask, the same procedure can be applied to the IF signal (with no pre-correction active). Interface C (RF or IF) Method This measurement is usually carried out with a spectrum analyser. The spectral density of a terrestrial DVB signal is defined as the long-term average of the time-varying signal power per unity bandwidth (i.e. 1 Hz). Values for other bandwidths can be achieved by proportional increase of the values for unity bandwidth. To avoid regular structures in the modulated signal a non-regular, e.g. a Pseudo-Random Binary Sequence (PRBS) is applied as an input signal to the modulator. For the resolution bandwidth, the recommended values should not exceed 30 kHz. Preferred values are approximately 3 kHz. The measurement should be Noise-normalized to 3 kHz. Reference Clause 10 of ETSI EN 302 755 [i.27]. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 11.3.9 Receiver sensitivity/dynamic range for a Gaussian channel | Purpose For network planning purposes, the minimum and maximum input powers for normal operation of a receiver have to be determined. Interface Test signals are applied and measured at interface C; interfaces C3 or C4 are used for the monitoring of BER or PER. Method The minimum and maximum input power thresholds for QEF reception should be measured. The failure point is defined as ESR5. ESR (Errored Second Ratio) is defined in clause 11.3.18. One errored second during a time interval of 20 seconds is given as ESR5 (5 % of the seconds are errored). Alternatively, the QEF point is used that is defined as BER after LDPC = 10-7. Since this measurement is time-consuming, the BER after LDPC = 10-4 is measured and the associated C/N value is increased by 0,2 dB. This corresponds to a BER after BCH = 10-11. The dynamic range is the difference of the input power between the measured values in dB. Reference Clause 6.1.2 of ETSI EN 302 755 [i.27]. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 88 11.3.10 Linearity characterization (shoulder attenuation) Purpose The shoulder attenuation can be used to characterize the linearity of an OFDM signal. Interface C Method The following procedure is applied to the measured RF spectrum of the transmitter output signal: a) Identify the maximum value of the spectrum by using a resolution bandwidth at approximately 10 times the carrier spacing. b) Place declined, straight lines connecting the measurement points at 300 kHz and 700 kHz from each of the upper and lower edges of the spectrum. Draw additional lines parallel to these, so that the highest spectrum value within the respective range lies on the line. c) Subtract the power value of the centre of the line (500 kHz away from the upper and lower edge) from the maximum spectrum value of (a) and note the difference as the "shoulder attenuation" at the upper and lower edge. d) Take the worst case value of the upper and lower results from (c) as the overall "shoulder attenuation". NOTE: For a quick overview the value at e.g. 500 kHz can be measured directly provided that coherent interferers are not present. The method described here is used for DVB-T2 signals with a nominal channel width of 5, 6, 7, 8 or 10 MHz. The reference frequency points are the same for non-extended and extended modes (for reasons of protection of adjacent channels). The recommended method is the same as for DVB-T for reason of comparability. If a spectrum mask is used, it should be always the spectrum mask of the non-extended mode, even if the signal uses extended bandwidth. For DVB-T2 signals with a nominal channel width of 1,7 MHz an appropriate spectrum masks has to be defined. If such a spectrum mask is not specified, the measurement points should be set 150 kHz below or above the respective edge carrier. Figure 11.7: Measured spectrum of different DVB-T2 transmission modes Reference Clause 10 of ETSI EN 302 755 [i.27]. +300 kHz +700 kHz +500 kHz ETSI ETSI TR 101 290 V1.4.1 (2020-06) 89 11.3.11 Power efficiency Purpose To compare the overall efficiency of DVB transmitters. Interface C Method Power efficiency is defined as the ratio of the DVB output power to the total power consumption of the chain from TS input to the RF signal output including all necessary equipment for operation such as blowers, transformers etc. (and is usually quoted in % terms). The operational channel and the environmental conditions need to be specified. Reference n/a 11.3.12 PAPR effect Purpose To measure the PAPR effect in the OFDM signal by applying one of the PAPR methods (ACE Active Constellation Extension or TR Tone Reservation). The method is to measure the performance difference without the PAPR technique and with the PAPR. The parameters of interest are the power increase, CCDF, MER performance increase. Interface C (modulator output, transmitter output) Method CCDF effect (modulator output) Note that the CCDF impact should be measured before the amplification in order to measure the effect of amplifier saturation. Comparison of CCDF charts (cumulative complementary distribution function) and indicating the difference in dB in the CCDF chart for a specified probability (e.g. 10-7). Special attention should be given to the requirement that the instrument settings should not be changed between the two measurements (with and without PAPR) and no overload or clipping is effecting the OFDM signal. The number of samples for such a comparison should be >= 107. Figure 11.8: Example of CCDF measurement of PAPR effect ETSI ETSI TR 101 290 V1.4.1 (2020-06) 90 Reference Clause 9.6 of ETSI EN 302 755 [i.27]. At transmitter output: Power increase: Signal Mean power is first measured before and after the PAPR is applied. For the tone reserved carriers, it means that it is measured with empty tone reserved carriers (before PAPR is applied) and after PAPR is applied. The power increase should be kept as low as possible. MER improvement: MER is first measured before and after the PAPR is applied. For the tone reserved carriers, it means that it is measured with empty tone reserved carriers (before PAPR is applied) and after PAPR is applied. For ACE case, MER measurement method should follow the procedure described in annex L. 11.3.13 P1 Symbol Error Rate Purpose This measurement gives an indication of the P1 Symbol Error rate. Interface C5 Method The data transmitted within the S1 and S2 field are decoded and compared to the correct data. A P1 symbol is erroneous, if at least one bit error within the S1 or the S2 field occurred. The measurement of this variable allows an estimation of the T2 signal level if the actual payload data is not decodable. The correct values of the S1 and S2 field can e.g. be obtained by the S1/S2 field transmitted within the L1-Pre data. Reference Clause 6.1.2 of ETSI EN 302 755 [i.27]. 11.3.14 BER before LDPC (inner) decoder Purpose This measurement gives an in-service indication of the un-coded performance of the transmitter. Since the residual BER contributions from transmitter and (test) receiver contribute to the result of this measurement, the contribution of the (test) receiver should be negligible when validating a transmitter. Interface C3 Method The BER before LDPC is measured separately for each PLP. It allows the identification of sporadic bit errors in a transmitter output signal. The averaging period for the calculation of the BER before LDPC should be set so that sporadic bit errors are not averaged out. Reference Clause 6.1.2 of ETSI EN 302 755 [i.27]. 11.3.15 Number of LDPC iterations Purpose This measurement gives an in-service indication of the quality of the received signal and the computational resources activated for the LDPC decoder. Since the result of this measurement is largely dependent on the actual LDPC decoder implementation, results can only be compared when taken from the same test instrument. Interface C3 Method The number of LDPC iterations is measured separately for each PLP. The end of the iterations is reached when the number of remaining errors is lower or equal than the error correction capability of the following BCH decoder, or when the maximum number of LDPC iterations is reached. An error-free signal requires a minimum of one iteration of the LDPC decoder. The average of the Number of LDPC iterations should be calculated over 1 second, and the maximum value during 1 second should also be displayed together with the average value. In case the data rate is very low and frames of the respective PLP are received at longer intervals, periods for averaging and display of maxima should be set accordingly. Reference Clause 6.1.2 of ETSI EN 302 755 [i.27]. NOTE: It is recommended to provide an indication if the LDPC decoder does not converge. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 91 11.3.16 BER before BCH (outer) decoder Purpose The BER is the primary parameter which describes the quality of the digital transmission link. It provides a quick indication of potential problems, especially in cases were sporadic errors would lead only to small increases of BER after BCH. Interface C4 Method The BER before BCH is measured separately for each PLP. The calculation can be based on the re- encoded signal that is available after the BCH decoder. The BER is defined as the ratio between erroneous bits and the total number of transmitted bits. The time interval for this calculation should be definable. Reference Clause 6.1.1 of ETSI EN 302 755 [i.27]. 11.3.17 Baseband Frame Error Rate BBFER Purpose To gain information about the number of baseband frames which are effected by bit errors. Interface C5 Method The BBFER is measured separately for each PLP. A Baseband Frame is erroneous, if an uncorrectable error has been discovered and indicated by the error flag by the BCH decoder. The parameter is either given as a ratio or as the number of erroneous BB frames per second. Reference Clause 5.1.7 of ETSI EN 302 755 [i.27]. 11.3.18 Errored Second Ratio ESR Purpose To obtain statistical information about the link quality. Interface C5 Method Time intervals of the length of 1 second are defined according to the system clock reference. Those seconds during which a Baseband Frame Error occurred are marked as 'errored'. The ratio of Errored Seconds to the total time elapsed or to a period set by the test receiver is indicated as ESR. In practice, a time interval of 20 seconds is used. Reference Clause 5.1.7 of ETSI EN 302 755 [i.27]. 11.3.19 IQ signal analysis 11.3.19.1 Introduction The IQ analysis can be applied on single carriers of the OFDM signal as well as on groups of carriers. If a group of carriers is evaluated all received symbols of this group can be superimposed in order to obtain the respective measurement parameters or the constellation diagram. The definitions for the I/Q related parameters are based on the following assumptions: - a constellation diagram of M symbol points and K carriers under consideration with 0 < K ≤ KMAX +1 and KMAX + 1 is the total number of active OFDM carriers; - a measurement sample of N data points, where N is sufficiently larger than M × K to deliver the wanted measurement accuracy; and - the co-ordinates of each received data point j being Ij + δIj, Qj + δQj where I and Q are the co-ordinates of the ideal symbol point and δI and δQ are the offsets forming the error vector of the data point (as long as the respective carrier is a "useful" one). ETSI ETSI TR 101 290 V1.4.1 (2020-06) 92 11.3.19.2 Modulation Error Ratio (MER) Purpose To provide a "figure of merit" analysis for L1 Signalling data and each PLP of the DVB-T2 signal, typically at a transmitter output (for assessing the quality of the transmitted signal) or in a fixed location in a SFN (for identifying severe distortions in the set-up of the transmitters forming the SFN). Interface C2 (i.e. after equalization) Method The carrier frequency of the OFDM signal and the symbol timing are recovered. Origin offset of the centre carrier (e.g. caused by residual carrier or DC offset), Quadrature Error (QE) and Amplitude Imbalance are not corrected. A time record of N received symbol co-ordinate pairs ( ) j j Q I ~ , ~ is captured. For each received symbol, a decision is made as to which symbol was transmitted. The error vector is defined as the distance from the ideal position of the chosen symbol (the centre of the decision box) to the actual position of the received symbol. This distance can be expressed as a vector ( ) j j Q I δ δ , . The sum of the squares of the magnitudes of the ideal symbol vectors is divided by the sum of the squares of the magnitudes of the symbol error vectors. The result, expressed as a power ratio in dB, is defined as the MER. ( ) ( ) dB Q I Q I MER N j j j N j j j + + × = = = 1 2 2 1 2 2 10 log 10 δ δ It should be reconsider that MER is just one way of computing a "figure of merit" for a vector modulated signal. Another "figure of merit" calculation is Error Vector Magnitude (EVM). MER and EVM are closely related and one can generally be computed from the other. For both parameters, MER and EVM, the display of the values as a function of frequency or carrier number can be very helpful. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 93 11.3.19.3 Signal to Interference Noise Ratio (SINR) Purpose To provide a "figure of merit" analysis for L1 Signalling data and each PLP of the DVB-T2 signal with special focus on field measurements where the MER is not applicable. Interface C1 (i.e. before equalization) Method The carrier frequency of the OFDM signal and the symbol timing are recovered. Origin offset of the centre carrier (e.g. caused by residual carrier or DC offset), Quadrature Error (QE) and Amplitude Imbalance are not corrected. A time record of N received complex QAM cells jd~ is captured at the input of the cell de-interleaver without equalization. For each received QAM cell, a decision is made which QAM symbol jdˆ was transmitted. This value is then multiplied by the estimated channel transfer function j Hˆ , which defines the ideal position of the received QAM cell. The absolute squared value 2 ˆ ˆ j j d H ⋅ defines its signal power. In the absence of any disturbance, the value j j d H ˆ ˆ ⋅ should ideally match jd~ . The absolute squared value of the delta equals the interference plus noise power. The result, expressed as a power ratio in dB: dB d d H d H SINR N j j j j N j j j − ⋅ ⋅ × = = = 1 2 1 2 10 ~ ˆ ˆ ˆ ˆ log 10 It should be mentioned that the SINR is very similar to the MER, except that no equalization to the input data is performed. This avoids the effect of noise amplification in frequency selective channels. The display of the SINR values as a function of frequency or carrier number can be very helpful. 11.3.19.4 Carrier Suppression (CS) Purpose A residual carrier is an unwanted coherent signal added to the centre carrier of the OFDM signal. It may have been produced by DC offset voltages of the modulating I and/or Q signal or by crosstalk from the modulating carrier within the modulator. Interface C2 Method Search for systematic deviations of all constellation points of the centre carrier and isolate the residual carrier. Calculate the Carrier Suppression (CS) from the formula: × = RC sig P P CS 10 log 10 where PRC is the power of the residual carrier and Psig is the power of the centre carrier of the OFDM signal (without residual carrier). 11.3.19.5 Carrier Phase (CPh) Purpose The measurement of the phase of the residual carrier which is an unwanted coherent signal added to the centre carrier of the OFDM signal, allows for an automatic and efficient calibration of a modulator which optimizes the Carrier Suppression (CS). Interface C2 Method This parameter describes the phase of a not perfectly suppressed carrier (see Carrier Suppression). When the Carrier Suppression is infinite, a carrier phase cannot be measured. The reference for the Carrier Phase is the I axis. A residual carrier that is directed towards positive I values has the carrier phase of 0°. A residual carrier that is directed towards positive Q values has the carrier phase of 90°. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 94 11.3.19.6 Amplitude Imbalance (AI) Purpose To separate the QAM distortions resulting from Amplitude Imbalance (AI) of the I and Q signal from all other kind of distortions. Interface C2 Method Calculate the I and Q gain values vI and vQ from all points in a constellation diagram eliminating all other influences. Calculate Amplitude Imbalance (AI) from vI and vQ v v v v v v v v I Q I Q Q I Q I if if AI > × − ≥ × − = % 100 1 % 100 1 { ( ) ( ) ( ) ( ) ( ) ( ) i Q i I i N j j Q i M i i Q i i Q N j j I i M i i I i i I d d d Q N d Q d Q M I N d I d I M = + = + = = + = = = = = Error) Target System 9.18.3 clause in given as d of component - (Q 1 1 Error) Target System 9.18.3 clause in given as d of component - (I 1 1 i 1 1 i 1 1 δ ν δ ν ETSI ETSI TR 101 290 V1.4.1 (2020-06) 95 11.3.19.7 Quadrature Error (QE) Purpose The phases of the two carriers feeding the I and Q modulators have to be orthogonal. If their phase difference is not 90° a typical distortion of the constellation diagram results. Interface C2 Method Search for the constellation diagram error shown in Figure 11.9 and calculate the value of the phase difference Δϕ = ϕ1 - ϕ2 after having eliminated all other influences and convert this into degrees: ( ) [ ]° − × ° = ϕ ϕ π 2 1 180 QE I Q Decision Boundary Signal Point Decision Boundary Box 1 ϕ 2 ϕ 90°+QE Figure 11.9: Distortion of constellation diagram resulting from I/Q Quadrature Error (QE) ETSI ETSI TR 101 290 V1.4.1 (2020-06) 96 11.3.20 SFN synchronization Purpose To measure and adjust the signal delay of an OFDM transmitter to a given value so that the transmitters in an SFN can be synchronized. Interface C1 Method Based on a sufficient number of I/Q samples, the channel response of the received DVB-T2 signal is calculated. The measurement of the SFN synchronization depends on the location where this measurement is executed. The SFN synchronization is related to two independent parameters: a) Time synchronization: All signals received from the certain transmitters at the location of interest should be received within the Guard Interval. It is the task of the network planning that also echoes from these signals will not infringe this rule. Method: The complex channel impulse response is measured. In addition, the Guard Interval is indicated. Also pre-echoes are taken into account correctly, according to the positioning of the FFT window. The indication of the Guard Interval should always start with the start of the FFT window position. b) Frequency synchronization: The center frequency of all signals received from the certain transmitters should be within a certain limit. In practice, 1 Hz has been established as a good and practicable target value for this limit in case of an 8 MHz channel and an 8k FFT size. For other channel bandwidth and other FFT sizes, e.g.16k and 32k signals, a proportionally higher accuracy may be required. The SFN frequency offset can be calculated as a relative value. It should be referenced to the signal path that provides the maximum power. Method: The complex channel impulse response is measured from which the frequency offset of each echo is derived and indicated. Also pre-echoes have to be taken into account correctly. A positive SFN frequency offset value means that the local center frequency of the related echo is higher than the local center frequency of the reference signal. 11.3.21 L1 signalling error Purpose The consistency of the L1 data is essential for the successful decoding of the DVB-T2 signal by the receiver. Interface C3 Method The receiver checks the CRC within the corresponding L1-pre and L1-post data. 11.3.22 RMS Delay-Spread (RMS-DS) Purpose The performance of OFDM based systems depends on the characteristics of the transmission channel. The extent of the multipath propagation of the transmission channel is characterized by means of the root means square (RMS) of the channel delay spread. Interface C1 Method The RMS Delay-Spread is directly measured from the channel impulse response. It is calculated from all i multipath components with power i P and excess delay iτ that exceed a specified power threshold (e.g. -15 dB, -20 dB, -25 dB) with respect to the strongest component. The excess delay (or delta) i τ is defined as the delta of the multipath component i with respect to the strongest signal component. The calculation should be limited to the Guard Interval to avoid ambiguities. = = = = = − ⋅ = n i i n i i i d n i d i i n i i RMS P P with P P 1 1 1 2 2 1 ) ( ) ( 1 τ τ τ τ τ ETSI ETSI TR 101 290 V1.4.1 (2020-06) 97 11.3.23 Maximum Excess Delay (MED) Purpose The performance of OFDM based systems depends on the characteristics of the transmission channel. The largest multipath delay of the transmission channel is characterized by means of the Maximum Excess Delay (MED). Interface C1 Method The Maximum Excess Delay is directly measured from the channel impulse response. It is defined as the maximum delay max τ of all multipath components that exceed a specified power threshold of e.g. -15 dB, -20 dB, -25 dB with respect to the strongest component. The maximum excess delay max τ is defined with regard to the first arrived component above the specified power threshold. 11.3.24 Receiver Buffer Model (RBM) validation test Purpose This test applies a test stream for an out-of-service test to the receiver to validate the proper functioning of the receiver buffer. Interface C Method A DVB-T2 signal is applied to the input of a DVB-T2 receiver. This DVB-T2 signal contains a test stream (e.g. V&V number 7xx). The test stream is generated (typically in the T2 gateway) and includes a modified Common PLP. The modified Common PLP for this test does not contain the tables and service information as usual, but an externally applied video/audio test stream. The failure point is defined as ESR5. ESR (Errored Second Ratio) is defined in clause 11.3.18. One errored second during a time interval of 20 seconds is given as ESR5 (5 % of the seconds are errored). 11.3.25 Relative power level during the non-P1 part of the FEF (RLF_non_P1) Purpose The purpose of the test is to measure the average power level of the non-P1 signal (excluding the rise time/fall time between the P1 and adjacent waveforms) of the FEF relative to the power of the T2 signal, excluding the FEF. This procedure allows for in-service measurements of co-channel interference (CCI) when empty FEFs are applied. Interface C Method The DVB-T2 system makes it possible to broadcast a FEF part carrying a non-T2 waveform at potentially another power level, including an empty FEF. Empty FEF means null power during non-P1 part of the FEF. A DVB-T2 signal is generated including a FEF part, which contains some other waveform than DVB- T2, e.g. T2 TX-SIG or an empty FEF. During the FEF period the power level of the non-P1 part of the FEF is measured. In addition the power level of the DVB-T2 signal (excluding the FEF part) is measured. A fast enough power sensor gated properly, or a spectrum analyser capable for channel power measurements, or a DVB-T2 receiver can be used for this measurement. NOTE: Note that DVB-T2 receivers of the 1st generation should ignore FEFs. 12 Measurements for the second generation cable system (DVB-C2) |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.1 Introduction | The DVB-C2 system as it is addressed in the following clauses, spans from the input interface (MPEG TS or GSE stream) to the output interface (RF signal) of the DVB-C2 transmitter. NOTE: The development of DVB-C2 modulator interface is not finalized at the time of completion of these Measurement Guidelines. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 98 Figure 12.1: High level C2 modulator block diagram [i.28] DVB-C2 supports TS, any packetized and continuous input formats as well as the so called Generic Stream Encapsulation (GSE). All input streams are multiplexed into a Baseband Frame format. The Forward Error Correction (FEC) scheme is applied to these Baseband Frames. In line with the other DVB-X2 systems, DVB-C2 uses a combination of LDPC and BCH codes, which is a very powerful FEC providing about 6 dB improvement of signal-to-noise ratio (SNR) with reference to DVB-C. Appropriate Bit-Interleaving schemes optimize the overall robustness of the FEC system. Extended by a FEC Frame header, those frames compose the payload of Physical Layer Pipes (PLP). One or several of such PLPs are multiplexed into a Data Slice. A two-dimensional interleaving (in time and frequency domain) is applied to each slice enabling the receiver to eliminate impacts of burst impairments and frequency selective interference such as single frequency ingress. One or several Data Slices compose the payload of a C2-frame. The Frame Building process includes inter alia the insertion of Continual and Scattered Pilots. The first symbol of a DVB-C2 frame, the so-called "Preamble", carries the signalling data. A DVB-C2 receiver will find all relevant configuration data about the structure and the technical parameters of the DVB-C2 signal in the signalling data block in the Preamble as well as in the headers of the PLPs. In the following step the OFDM symbols are generated by means of an Inverse Fast Fourier Transformation (IFFT). A 4K-IFFT algorithm is applied generating a total of 4 096 sub-carriers, 3 409 of which are actively used for the transmission of data and pilots within a frequency band of 8 MHz. The guard interval used between the OFDM symbols has a relative length of either 1/128 or 1/64 in reference to the symbol length (448 μs). Figure 12.2: High-level block diagram of the signal processing defined for the DVB-C2 transmitting end [i.29] ETSI ETSI TR 101 290 V1.4.1 (2020-06) 99 The following clauses specify a number of tests and measurements at the modulator output interface and the corresponding input interface at the receiver side. With regard to the signalling, it is recommended that a measurement instrument should display the signalled information (DVB-C2 Layer 1 and Layer 2 signalling) as readable text and abbreviations. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2 Measurements for DVB-C2 baseline system | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.0 General | This clause lists a number of measurements for the DVB-C2 baseline system. A list of the main application area of the DVB-C2 measurement parameters described in this clause is given in table 11.1. The measurements in clause 6.1 "System availability" and clause 6.2 "Link availability" are also applicable at the TS interfaces of transmitter and receiver. Table 12.1 DVB-C2 measurement parameters and their application area Measurement parameter Transmitter Network Receiver In-service measurement 11.3.1 RF measurements 11.3.1.1 RF frequency accuracy X X X 11.3.1.2 RF occupied bandwidth X X 11.3.3 AFC capture range X 11.3.4 Phase noise of Local Oscillators (LO) X X 11.3.5 RF/IF signal power X X X X 11.3.7 Noise Power X X X X 11.3.8 RF and IF spectrum X X X 11.3.9 Receiver sensitivity/dynamic range for a Gaussian channel X 11.3.10 Linearity characterization (shoulder attenuation) X 11.3.14 BER before LDPC (inner) decoder X X 11.3.15 Number of LDPC iterations X X 11.3.16 BER before BCH (outer) decoder X X 11.3.17 Baseband Frame Error Rate BBFER X X 11.3.19 IQ signal analysis 11.3.19.2 Modulation Error Ratio (MER) X X X 11.3.19.6 Amplitude Imbalance (AI) X X 11.3.19.7 Quadrature Error (QE) X X 11.3.21 L1 signalling error X X X 11.3.24 Receiver Buffer Model (RBM) validation test X NOTE: As an In-service measurement in an unoccupied channel, the measurement of Noise Power can provide an overview of the conditions in a certain frequency band. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.1 RF measurements | |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.1.0 General | The measurement of some basic parameters of the DVB-C2 OFDM signal may be carried out at the RF layer with a test receiver, a spectrum analyser or similar instruments. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 100 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.1.1 RF frequency accuracy | Purpose Successful processing of OFDM signals requires that certain carrier frequency accuracy needs to be maintained at the transmitter with respect to the relative and absolute frequency position of the signal/carriers. Interface RF Method The measurement of the RF frequency accuracy determines the relative carrier offset regarding its carrier spacing. Furthermore, the measurement determines the absolute frequency/carrier number offset of the signal with respect to its configured frequency location (absolute carrier position). a) Test receiver method: relative carrier offset is derived from the digital samples after the test receiver has synchronized to the incoming DVB-C2 signal. In this case, the accuracy is typically expressed as 'Carrier Offset' and given in Hz or ppm. Furthermore, the absolute frequency offset is typically given in number of carriers. b) Spectrum analyser method: the relative carrier offset is derived from the frequencies measured for the continual pilots and/or the edge pilots which are continuously (per OFDM symbol) transmitted at a boosted amplitude in contrary to the scattered pilots. The absolute frequency offset can only be determined in knowledge of the DVB-C2 signal configuration, i.e. the Layer 1 Signalling and the START_FREQUENCY parameter, or by pattern matching of the continual pilot positions. NOTE: The assumption is that the frequency offset is not greater than the frequency difference between adjacent pilots. Reference Clauses 9.6.3, 9.6.4 of ETSI EN 302 769 [i.28]. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.1.2 RF sampling frequency | Purpose The measurement of the occupied bandwidth allows the verification of the correct sampling frequency at the modulator. Interface RF Method The occupied bandwidth (7,61 MHz or 5,71 MHz respectively) is calculated from the measurements of the frequencies of the edge pilots and/or continual pilots of the DVB-C2 signal. a) Spectrum analyser method: if the frequencies of the edge carriers of the DVB-C2 spectrum boundary are known then the related values for the occupied bandwidth may be calculated. Denoting the edge pilot frequencies as Fmin and Fmax the occupied bandwidth is appropriately OB = Fmax - Fmin + 1/TU. NOTE: If the measurement is based on pilots that are closer to each other than 7,61 MHz or 5,71 MHz in the frequency domain, a reduction of the accuracy of the calculated occupied bandwidth should be considered. Notches in the channel where the measurement is carried out need to be taken into account. b) Test receiver method: the occupied bandwidth is derived from the digital samples after the test receiver has synchronized to the incoming DVB-C2 signal. Reference Clauses 9.6.3, 9.6.4 of ETSI EN 302 769 [i.28]. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 101 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.2 AFC capture range | Purpose To determine the frequency range over which the receiver will acquire overall lock. Interface RF input for the application of the test signal Method • Set RF tuning frequency in the C2 signalling to L1 START_FREQUENCY + OFDM bandwidth/2 in the modulator AND demodulator (C2 'Absolute OFDM' requirement) as a first step; • then repeat the procedure as described hereafter: A signal is applied to the input of the receiver, at a level 10 dB above the minimum input power as defined in "Receiver sensitivity" (see clause 11.3.9). The signal is frequency shifted in steps (from below and above) towards a nominal value, whilst forcing the receiver to re- acquire after each step. Correct reception is assumed if after each step: a) the receiver can synchronize to the applied DVB-C2 signal, b) the Frame Error Rate or the rate of visible errors is below the failure point which is defined as 1 frame error or visible error in 20 seconds. • note the signal could be frequency shifted at the modulator or the receiver RF tuner, if shifted at the modulator then the RF frequency needs to be decoupled from the L1 signalling (L1 signalling SHOULD NOT change while the LO frequency is changed otherwise the demodulator will not acquire to the signal even if within the AFC capture range) • OFDM bandwidth/2 = 1 704 × 64/(7 × 4 096) MHz for 8 MHz tuning window (or OFDM bandwidth/2 = 1 704 × 48/(7 × 4 096) MHz for 6 MHz tuning window) • for tuning window > 8 MHz or 6 MHz respectively adjust tuning frequency to centre point between the two outermost OFDM subcarriers of the C2 system Reference Clause 10.1.1.1 of ETSI TS 102 991 [i.29]. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.3 Phase noise of Local Oscillators (LO) | Purpose Phase noise can be introduced by the transmitter, by any frequency converter or by the receiver. In an OFDM system the phase noise can cause Common Phase Error (CPE) which affects all carriers simultaneously, and which can be minimized or corrected by using the continual pilots. However the noise-like Inter-Carrier Interference (ICI) cannot be corrected. This measurement may be useful for manufacturing, incoming inspection and maintenance of modulators, transmitters, up/down converters and receivers, either professional or consumer type. Interface Any access to Local Oscillators (LO), in transmitters, converters and receivers. Method Phase noise can be measured with a spectrum analyser, a vector analyser or a phase noise test set. Reference n/a |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.4 RF/IF signal power | Purpose Signal power, or wanted power, measurement is required to set and check signal levels at the transmitter and receiver sites. Interface RF Method The signal power of a DVB-C2 signal is defined as the mean power of the signal between the outermost pilots of the 7,61 MHz (or of the 5,71 MHz) signal. The measurement is typically carried out with a spectrum analyser or a test receiver that integrates the signal power between a lower and an upper limit on the frequency axis. NOTE: The same procedure can be applied to wider DVB-C2 signals, e.g. 32 MHz at the output of a modulator/transmitter. If the signal includes notches, they have to be empty, i.e. not filled with other signals. Reference n/a ETSI ETSI TR 101 290 V1.4.1 (2020-06) 102 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.5 Noise power | Purpose Noise is a significant impairment in a transmission network. Interface RF Method The noise power , or unwanted power, can be measured with a spectrum analyser (out of service). The noise power is specified using the occupied bandwidth of the C2 signal, i.e. 7,61 MHz or 5,71 MHz. The measurement can also be carried out as an in-service measurement by inserting DVB-C2- compliant notches where the noise power is measured by a spectrum analyser or a test receiver. NOTE: The Carrier-to-Noise ratio C/N should be calculated as the ratio of the signal power, measured as described in clause 11.3.5, to the noise power, measured as described in this clause. Reference n/a |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.6 RF and IF spectrum | Purpose To avoid interfering with other channels, the transmitted RF spectrum should comply with a spectrum mask, which is defined for the cable network, typically by the cable network operator. Interface RF Method This measurement is usually carried out with a spectrum analyser or a test receiver. The spectral density of a DVB-C2 signal is defined as the long-term average of the time-varying signal power per unity bandwidth (i.e. 1 Hz). Values for other bandwidths can be achieved by proportional increase of the values for unity bandwidth. For the resolution bandwidth, the recommended values should not exceed 30 kHz. Preferred values are 3 kHz or lower. The measurement should be Noise-normalized to the same bandwidth. Reference Clause 10.3 of ETSI EN 302 769 [i.28]. |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.7 Receiver sensitivity/dynamic range for a Gaussian channel | Purpose For network planning purposes, the minimum and maximum input powers for normal operation of a receiver have to be determined. Interface Test signals are applied at the RF input and measured at Baseband level (or by expert viewing on a screen). Method The minimum and maximum input power thresholds for good reception should be measured by stepwise changes of the input power. For 'good reception', the Frame Error Rate or the rate of visible errors is below the failure point which is defined as 1 frame error or visible error in 20 seconds. Both measurement should provide results that are very close to the cut-off point as determined by the chosen FEC parameters (typically within 0,1 dB). The dynamic range is the difference between the maximum and minimum input power in dB. Reference n/a |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.8 Linearity characterization (shoulder attenuation) | Purpose The linearity of an OFDM signal characterizes the level of interference in the adjacent bands. The requirements are normally defined by the cable network operator who also is likely to define acceptable levels for ACLR (Adjacent Channel Leakage Ratio), out-of-band emissions and spurious emissions. Interface RF (Modulator/transmitter output) Method Spectrum analyser or test receiver measurement Reference n/a ETSI ETSI TR 101 290 V1.4.1 (2020-06) 103 |
8db1a1fae619f4f804112b02d36a85ca | 101 290 | 12.2.9 BER before LDPC (inner) decoder | Purpose This measurement gives an in-service indication of the un-coded performance of the transmitter. Since the residual BER contributions from transmitter and (test) receiver contribute to the result of this measurement, the contribution of the (test) receiver should be negligible when validating a transmitter. Interface Baseband Method The BER before LDPC is measured separately for each PLP. It allows the identification of sporadic bit errors in a transmitter output signal if the averaging period for the calculation of the BER before LDPC is set so that sporadic bit errors are not averaged out. Reference Clause 6.1 of ETSI EN 302 769 [i.28]. 12.2.10 Number of LDPC iterations Purpose This measurement gives an in-service indication of the quality of the received signal and the computational resources activated for the LDPC decoder. Since the result of this measurement is largely dependent on the actual LDPC decoder implementation, results can only be compared when taken from the same test instrument. Interface Baseband Method The number of LDPC iterations is measured separately for each PLP. The end of the iterations is reached when the number of remaining errors is lower or equal than the error correction capability of the following BCH decoder, or when the maximum number of LDPC iterations is reached. An error-free signal requires a minimum of one iteration of the LDPC decoder. The average of the number of LDPC iterations should be calculated over 1 second, and the maximum value during 1 second should also be displayed together with the average value. In case the data rate is very low and frames of the respective PLP are received at longer intervals, periods for averaging and display of maxima should be set accordingly. Reference Clauses 10.9, 11.2 of ETSI TS 102 991 [i.29]. NOTE: It is recommended to provide an indication if the LDPC decoder does not converge. 12.2.11 BER before BCH (outer) decoder Purpose The BER is the primary parameter which describes the quality of the digital transmission link. It provides a quick indication of potential problems, especially in cases where sporadic errors would lead only to small increases of BER after BCH. Interface Baseband Method The BER before BCH is measured separately for each PLP. The calculation can be based on the re- encoded signal that is available after the BCH decoder. The BER is defined as the ratio between erroneous bits and the total number of transmitted bits. The time interval for this calculation should be definable. Reference Clause 8.4.3.2 of ETSI EN 302 769 [i.28]. 12.2.12 Baseband Frame Error Rate BBFER Purpose To gain information about the number of baseband frames which are affected by bit errors. Interface Baseband Method The BBFER is measured separately for each PLP. A Baseband Frame is erroneous, if an uncorrectable error has been discovered and indicated by the error flag by the BCH decoder. The parameter is either given as a ratio or as the number of erroneous BB frames per second. Reference Clause 5.1.6 of ETSI EN 302 769 [i.28]. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 104 12.2.13 IQ signal analysis 12.2.13.1 Introduction The IQ analysis can be applied to single carriers of the OFDM signal as well as to groups of carriers. If a group of carriers is evaluated all received symbols of this group can be superimposed in order to obtain the respective measurement parameters or the constellation diagram. The definitions for the I/Q related parameters are based on the following assumptions: - a constellation diagram of M symbol points and K carriers under consideration with 0 < K ≤ KMAX +1 and KMAX + 1 is the total number of active OFDM carriers; - a measurement sample of N data points, where N is sufficiently larger than M × K to deliver the wanted measurement accuracy; and - the co-ordinates of each received data point j being Ij + δIj, Qj + δQj where I and Q are the co-ordinates of the ideal symbol point and δI and δQ are the offsets forming the error vector of the data point (as long as the respective carrier is a "useful" one). ETSI ETSI TR 101 290 V1.4.1 (2020-06) 105 12.2.13.2 Modulation Error Ratio (MER) Purpose To provide a "figure of merit" analysis for L1 Signalling data [MER(L1p2)] and each PLP [MER(PLP), MER(Common-PLP)] of the DVB-C2 signal, typically at a transmitter output (for assessing the quality of the transmitted signal) or in a fixed location in a network. Interface Internal I/Q interface (test receiver or modulation analyser) Method The carrier frequency of the OFDM signal and the symbol timing are recovered. Origin offset of the centre carrier (e.g. caused by residual carrier or DC offset), Quadrature Error (QE) and Amplitude Imbalance are not corrected. A time record of N received symbol co-ordinate pairs ( ) j j Q I ~ , ~ is captured. For each received symbol, a decision is made as to which symbol was transmitted. For the calculation of MER(PLP), the information derived from the LDPC decoder should be used to calculate the ideal symbol position transmitted ( ) j j Q I , . The error vector is defined as the distance from the ideal position of the chosen symbol to the actual position of the received symbol. This distance can be expressed as a vector ( ) j j Q I δ δ , . The sum of the squares of the magnitudes of the ideal symbol vectors is divided by the sum of the squares of the magnitudes of the symbol error vectors. The result, expressed as a power ratio in dB, is defined as the MER. ( ) ( ) dB Q I Q I MER N j j j N j j j + + × = = = 1 2 2 1 2 2 10 log 10 δ δ NOTE: In OFDM systems, a symbol consists of many OFDM subcarriers. The calculation for one symbol means, that all subcarriers should be considered that are included in this symbol. Thus, a single symbol contains a plurality of error vectors. It should be reconsidered that MER is just one way of computing a "figure of merit" for a vector modulated signal. Another "figure of merit" calculation is Error Vector Magnitude (EVM). MER and EVM are closely related and one can generally be computed from the other. Error Vector Magnitude (EVM) is defined as: ( ) ( ) % 100 1 2 2 1 2 2 ⋅ + + = = = N j j j N j j j RMS Q I Q I EVM δ δ For both parameters, MER and EVM, the display of the values as a function of frequency or carrier number can be very helpful. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 106 12.2.13.3 Amplitude Imbalance (AI) Purpose To separate the QAM distortions resulting from Amplitude Imbalance (AI) of the I and Q signal from all other kind of distortions. Interface Internal I/Q interface (test receiver or modulation analyser) Method Calculate the I and Q gain values vI and vQ from all points in a constellation diagram eliminating all other influences. Calculate Amplitude Imbalance (AI) from vI and vQ v v v v v v v v I Q I Q Q I Q I if if AI > × − ≥ × − = % 100 1 % 100 1 { ( ) ( ) ( ) ( ) ( ) ( ) i Q i I i N j j Q i M i i Q i i Q N j j I i M i i I i i I d d d Q N d Q d Q M I N d I d I M = + = + = = + = = = = = Error) Target System 9.18.3 clause in given as d of component - (Q 1 1 Error) Target System 9.18.3 clause in given as d of component - (I 1 1 i 1 1 i 1 1 δ ν δ ν ETSI ETSI TR 101 290 V1.4.1 (2020-06) 107 12.2.13.4 Quadrature Error (QE) Purpose The phases of the two carriers feeding the I and Q modulators have to be orthogonal. If their phase difference is not 90° a typical distortion of the constellation diagram results. Interface Internal I/Q interface (test receiver or modulation analyser) Method Search for the constellation diagram error shown in Figure 12.3 and calculate the value of the phase difference Δϕ = ϕ1 - ϕ2 after having eliminated all other influences and convert this into degrees: ( ) [ ]° − × ° = ϕ ϕ π 2 1 180 QE I Q Decision Boundary Signal Point Decision Boundary Box 1 ϕ 2 ϕ 90°+QE Figure 12.3: Distortion of constellation diagram resulting from I/Q Quadrature Error (QE) 12.2.14 L1 signalling error Purpose The consistency of the L1 data is essential for the successful decoding of the DVB-C2 signal by the receiver. Interface Baseband Method The receiver checks the CRC within the corresponding L1 signal. Reference Clause 8.3.2 of ETSI EN 302 769 [i.28] 12.2.15 Receiver Buffer Assumptions (RBA) validation test Purpose This test applies a test stream for an out-of-service test to the receiver to validate the proper functioning of the receiver buffer. Interface Baseband Method A DVB-C2 signal is applied to the input of a DVB-C2 receiver. This DVB-C2 signal contains a test stream (e.g. VV017,018,019 (M-PLP)). & ISSY The failure point is defined as: the Frame Error Rate or the rate of visible errors is below 1 frame error or visible error in 20 seconds.. Reference n/a NOTE: These test streams exercise the buffer but there are no specific tests defined or proposed yet to test buffer limits. Therefore, this clause is kept in the present document as a placeholder and can be revised if such test streams become available. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 108 Annex A: General measurement methods A.1 Introduction It is recommended that manufacturers add the test mode described in this annex to certain professional grade cable and satellite broadcast equipment. This recommendation is relevant to equipment that implements the channel encoding schemes defined in ETSI EN 300 429 [i.6] (cable) and ETSI EN 300 421 [i.5] (satellite). The purpose of the recommended test mode is to simplify out of service testing of systems and system components by making the channel encoder able to generate a known, fixed, repeating bit sequence of an essentially pseudo-random nature. The central requirement is that when the channel encoder is in the test mode, the data entering the sync inversion/randomization function is a continuous repetition of one fixed TS packet. The fixed packet is defined as the four byte sequence 0x47, 0x1f, 0xff, 0x10, followed by 184 zero bytes (0 x 00). This form of data is a refinement of the null TS packet definition in ISO/IEC 13818-1 [i.1]. A.2 Null packet definition This clause summarizes the null packet definition from ISO/IEC 13818-1 [i.1] and then describes how the definition has been extended for the purpose of the recommended test mode. ISO/IEC 13818-1 [i.1] defines a null TS packet for the purposes of data rate stuffing. Table A.1 shows the structure of a null TS packet using the method of describing bit stream syntax defined in clause 2.4.3.3 of ISO/IEC 13818-1 [i.1]. This description is derived from tables 2-3 Transport Header (TH) in ISO/IEC 13818-1 [i.1]. The abbreviation "bslbf" means "bit string, left bit first", and "uimsbf" means "unsigned integer, most significant bit first". The column titled "Value", gives the bit sequence for the recommended null packet. A null packet is defined by ISO/IEC 13818-1 [i.1] as having: - payload_unit_start_indicator = "0"; - PID = 0x1FFF; - transport_scrambling_control = "00"; - adaptation_field_control value = "01". This corresponds to the case "no adaptation field, payload only". The remaining fields in the null packet that should be defined for testing purposes are: - transport_error_indicator which is "0" unless the packet is corrupted. For testing purposes this bit is defined as "0" when the packet is generated; - transport_priority which is not defined by ISO/IEC 13818-1 [i.1] for a null packet. For testing purposes this bit is defined as "0"; - continuity_counter which ISO/IEC 13818-1 [i.1] states is undefined for a null packet. For testing purposes this bit field is defined as "0000"; - data_byte which ISO/IEC 13818-1 [i.1] states may have any value in a null packet. For testing purposes this bit field is defined as "00000000". ETSI ETSI TR 101 290 V1.4.1 (2020-06) 109 Table A.1: Null TS packet definition Syntax No. of bits Identifier Value null_transport_packet(){ sync_byte 8 bslbf "01000111" transport_error_indicator 1 bslbf "0" payload_unit_start_indicator 1 bslbf "0" transport_priority 1 bslbf "0" PID 13 uimsbf "1111111111111" transport_scrambling_control 2 bslbf "00" adaptation_field_control 2 bslbf "01" continuity_counter 4 uimsbf "0000" for (I = 0;i<N;i++) { data_byte 8 bslbf "00000000" } } A.3 Description of the procedure for "Estimated Noise Margin" by applying statistical analysis on the constellation data Instead of adding real noise to the received signal this method uses statistical analysis and an iterative search algorithm to estimate the added noise power to reach the critical BER. 1) Demodulate the signal to produce a statistically significant sequence of data records. Each record represents the state of the demodulated I and Q components at a decision instant. 2) Compute the average noise power as the mean square of the error vectors and calculate the estimated Savg/Navg ratio. ( ) ( ) + + × = = = N j j j N N j j j N Q I Q I SNR 1 2 2 1 1 2 2 1 10 log 10 σ σ The σIj and σQj are the error vector co-ordinates which represent the offset from the co-ordinates of the centre (mean value) of the actual received data for a specific constellation point, to the actual received data point j. If only Gaussian noise is present as an impairment the "centre (mean value) of the actual received data for a specific constellation point" is identical to the ideal symbol point. N is the number of data points in the measurement sample. 3) Compute the additional noise power Nstep required to degrade the computed SNR by a certain amount. The value Nstep is usually determined by the iterative optimization procedure which is used. 4) For each data record in the sample compute the distances d from the true position of the signal at the decision instant to each of the decision boundaries with adjacent cells. For each of the directions +I, -I, +Q, -Q that would cause a symbol error, convert the distance to the decision boundary into the number of standard deviations (k) of a normal distribution with a variance corresponding to the added noise power. The variance of the added noise power is: step N = 2 σ ETSI ETSI TR 101 290 V1.4.1 (2020-06) 110 and the normalized standard deviation corresponding to the distance dI+ is for example: + + = I I d k σ 5) Compute the probability QS of a symbol error for each distribution tail due to an erroneous state transition in the relevant direction. ( ) ∞ − = k s dx x k Q 2 exp 2 1 2 π or ( ) = 2 2 1 k erfc k Qs 6) Compute the number of bit errors that the erroneous state transition would cause and calculate the bit error probability QB. One symbol error may result in more than one bit error for transitions across either the I or Q axis. Sum the individual QB values and divide by the number of points in the sample to get the average probability of a bit error. 7) Repeat the steps 4 to 6 for incremental values of noise power until the critical BER is found and calculate the noise margin: Noise Margin + × = avg added N N dB 1 log 10 ) ( 10 A.4 Set-up for RF phase noise measurements using a spectrum analyser The noise performance of the carrier can be characterized as the ratio of the measured power in one noise sideband component, on a per hertz of bandwidth spectral density basis, to the total signal power: ( ) × = l otal_signa power_of_t se_only) deband,pha ity(one_si power_dens fm 10 log 10 α in (dBc/Hz) and fm is the frequency distance away from the carrier. For this measurement it is assumed that contributions from amplitude modulation to the noise spectrum are negligible compared to those from frequency modulation and that ΔB, the measurement bandwidth, is much smaller than fm. A spectrum analyser with a noise measurement option is able to measure the power within 1 Hz bandwidth. If this is not available the resolution bandwidth should be as small as possible and the video bandwidth has to be 10 or 20 times smaller in order to get sufficient averaging of the noise over time. For example: carrier frequency: 36 MHz fm = 10 kHz ΔB = Equivalent Noise Bandwidth (ENB) of the resolution bandwidth filter: 270 Hz video bandwidth: 10 Hz or 30 Hz NOTE 1: Spectrum analysers typically use near Gaussian filters for the resolution bandwidth with a 20 % tolerance. The Equivalent Noise Bandwidth (ENB) is equal to the bandwidth of the filter measured at -3,4 dB, (by actually measuring the filter of the spectrum analyser, the 20 % tolerance factor is eliminated). ETSI ETSI TR 101 290 V1.4.1 (2020-06) 111 Then the following conversion to 1 Hz bandwidth can be applied: ( ) B log 10 er signal_pow r_in_DB noise_powe log 10 10 10 Δ × − × ≅ m f α + 2,5 dB in [dBc/Hz] NOTE 2: The 2,5 term accounts for the correction of 1,05 dB due to narrowband envelope detection and the 1,45 dB due to the logarithmic amplifier. Having measured α(fm) for various values of fm an estimation of equivalent peak phase deviation and frequency deviation is possible by using sinusoidal analogy: α(fm) ≅ 20 × log10(Δϕrms/ 2 ) in [dB/Hz] with Δϕ in [rad/Hz] The square root of the sum of all noise densities within the frequency range of interest will give the equivalent RMS phase noise error vector in the I/Q plane. An estimation can be done if the phase noise power slope may be approximated by the density function: [ ] Hz W f a Y b 1 = with 10 _ _ ] [ decade per dB slope b = (b > 0) and b f N a 1 0× = where ( ) = 10 0 1 10 f N α Then the total double-side-band phase noise power within the frequency range of interest (f1,f2) can be approximated by: ( ) ( ) ( ) − − = = − − − − 2 1 1 2 1 1 1 1 1 2 1 2 f f b b b f f b a df f a Noise Phase DSB For the normalized RMS error vector (carrier = 1) it follows: RMS Quadrature Error Vector ( ) ( ) ( ) ph b b f f b a σ = − − = − − 1 2 1 1 1 1 1 2 [ ] rad ph σ ϕ σ arctan ≅ Δ (for carrier = 1) A.5 Amplitude, phase and impulse response of the channel The amplitude, phase and impulse response can be derived from the equalizer tap coefficients. The use of a good equalizer that is designed to cope with the echo profile defined in clause B.14 is recommended to get accurate results in case of high linear distortions. The capabilities to derive the channel response from the equalizer tap coefficients depend on the structure of the equalizer. Especially the channel response in the Nyquist slope of the signal cannot be measured exactly with a T-spaced equalizer. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 112 A.6 Out of band emissions The out of band emissions can be measured using a spectrum analyser. The resolution bandwidth should be low enough to detect peaks in the out of band spectrum. The video filter should be at least 10 times lower than the resolution bandwidth for sufficient averaging of the noise-like signal. Figure A.1: Spectrum mask as defined in ETSI EN 300 429 [i.6] ETSI ETSI TR 101 290 V1.4.1 (2020-06) 113 Annex B: Examples for test set-ups for satellite and cable systems B.0 Introduction Even if not demonstrated in the diagrams of this clause and also not mentioned in the explanations the receiver may be a part of the measurement device. In this case all the interfaces defined in figure 4.2 are internal ones, where the measurement device has access to. B.1 System availability See clause 6.1. Because this measurement is based on the error_indicator_flag in the TS header set in any previous stage including the last stage of the transmission chain the signal at interface Z should be used. Figure B.1: Test set-up for system availability B.2 Link availability See clause 6.2. This measurement monitors the performance of an individual link. Therefore the RS information should be created and be correct at the start point of the link. The measurement set-up may rely on the overload information coming from the RS decoder in the receiver at interface X or on the transport_error_indicator in the header of the TS packets at interface Z. Figure B.2: Test set-up for link availability B.3 BER before RS B.3.0 General See clause 6.3. The measurement can be done as out service measurement or as in service measurement. In both cases the measurement time is an important parameter. This parameter should be selectable within a wide range by the user. Preferably the measurement should display the BER as a function of measurement time. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 114 B.3.1 Out of service measurement See clause 6.3.1. When the BER is measured out of service Null packets as defined in clause A.2 should be created and transmitted to the receiving site. At the receiving site the signal at the interface W is compared against the pre-calculated values. The time window for the BER measurement should be selectable by the user. Figure B.3: Test set-up for out of service BER measurement before RS decoding B.3.2 In service measurement See clause 6.3.2. In this case no special signal should be inserted in the transmitter. The measurement only relies on the results of the RS decoder. The measurement can be done by using the signals at the interfaces W and Z. Figure B.4: Test set-up for out of service BER measurement before RS decoding B.4 Event error logging See clause 6.4. This measurement relies on information coming from different parts of the receiver like tuner, RS decoder or a demultiplexer. Typically the receiver will be a part of the measurement device because it is not expected that all this information will be available at a standard receiver. Figure B.5: Test set-up for event error logging ETSI ETSI TR 101 290 V1.4.1 (2020-06) 115 B.5 Transmitter symbol clock jitter and accuracy See clause 6.5. This measurement requires a symbol clock output at the modulator. To this interface an appropriate frequency counter and/or jitter and wander analyser can be connected. Figure B.6: Test set-up for transmitter symbol clock measurement B.6 RF/IF signal power See clause 6.6. The signal power can be measured directly at the interfaces N or P or by using a calibrated splitter. If needed an appropriate filter should be used. Figure B.7: Test set-up for RF/IF level measurement B.7 Noise power B.7.0 General See clause 6.7. Typically all the power present in a channel which is not part of the signal can be regarded as unwanted noise. It can be produced from different origination and be of the form of random noise (thermal), pseudo-random (digitally modulated interfering carriers) or periodic (Continuous Waves CW or narrowband interferences), the first two are called non- coherent and the periodic ones are termed as coherent. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 116 B.7.1 Out of service measurement For doing this measurement the carrier should be switched off. The measurements can be done at interface N (RF level) or at interface P (IF level). Noise level can be measured with a spectrum analyser or any other appropriate device. If a power metre is used the equivalent noise bandwidth should be taken into account. In this case of out-of-service measurement, all different types of noise are measured simultaneously, and the measured result can be termed as unwanted power. Figure B.8: Test set-up or out-of-service noise level measurement B.7.2 In service measurement For the "in service measurement" eye diagrams or IQ constellation diagram derived from I and Q signals available at interface T should be employed. In this case of "in service measurement", it is possible to determine the type of the noise by applying the I/Q signal analysis (see clause 6.9). Figure B.9: Test set-up for in-service noise level measurement B.8 BER after RS See clause 6.8. The set-up is equivalent to clause 6.3 BER before RS. The comparison is done after RS at interface Y. B.9 I/Q signal analysis See clause 6.9. For this measurement eye diagrams or IQ constellation diagram derived from I and Q signals available at interface T should be employed. Figure B.10: Test set-up for I/Q signal analysis ETSI ETSI TR 101 290 V1.4.1 (2020-06) 117 B.10 Service data rate measurement The set-up is equivalent to B.1 The measurement is based on the TS only. B.11 Noise margin B.11.0 General See clause 7.1. Purpose To provide an indication of the reliability of the transmission channel (i.e. cable network), the noise margin measurement is a more useful measure of system operating margin than a direct BER measurement due to the steepness of the BER curve. Interface The reference interface for the noise injection is the RF interface (N, input of receiver). For practical implementation, other interfaces can be used, provided equivalence to the described set-up is ensured. Test set-up Figure B.11 shows the recommended test set-up for the measurement of noise margin. Figure B.11: Test Set-up for noise margin measurement B.11.1 Recommended equipment 1 I/Q baseband signal source for 64 QAM; S Switch (to switch off modulation); 2 I/Q modulator; 3 RF generator (see clause B.11.2 below, remark 1) (level and frequency adjustable); 4 cable network (see clause B.11.2, remark 2); 5 noise source (flat within the required measurement range) (see clause B.11.2, remark 3); 6 adjustable attenuator in 0,1 dB (max. 0,5 dB) steps; 7, 8 directive couplers (see clause B.11.2, remark 4); ETSI ETSI TR 101 290 V1.4.1 (2020-06) 118 9 spectrum analyser; 10 reference receiver with good equalizer (see clause B.11.2, remark 5); 11 counter of BER. B.11.2 Remarks and precautions 1) Adjust RF carrier level so that non-linear distortion (i.e. CW, CSO, CTB) has no impact to BER measurement. 2) Pay attention to the amplitude response of the noise spectrum. If it is not white Gaussian spectrum (flat amplitude response) figure B.12 takes care to measure: a) If the effect produced by the thermal random noise is the wanted measurement, then take the measurement at the lowest level found in the wanted band (P4 in figure B.12), because it is the closest approximation to the random white thermal noise, then normalize the result to the full bandwidth of the channel, defined by the symbol rate x(1 + α). b) If the mean unwanted power is to be reported in the measurement, then integrate the spectrum with a suitable spectrum analyser or use a power metre with the appropriate filter as per clause B.7.1. Figure B.12: Amplitude response of the noise spectrum 3) If a noise source with broadband output spectrum is used, avoid any affect to BER measurement by non-linear distortion due to an overload of the reference receiver's input amplifier stage. 4) Usual power splitters are allowed if sufficient matching at all ports is ensured for all measurement conditions (i.e. high attenuation in adjustable attenuator). 5) Influence of linear distortion of the cable network to the BER measurement should be negligible. B.11.3 Measurement procedure Step 1: Add noise to modulated cable network output until BER is 10-4. Step 2: Switch off modulation with (S); Measure Noise power N1 (dBm) beside carrier (Δ f ≥ 0,5 MHz). Step 3: Switch off noise source (5); Measure Noise power N2 (dBm) beside carrier. Step 4: Compute Noise Margin (NM): NM = N1 − N2 (dB) NOTE: Due to step 2, the measurement of noise margin is to be done under out of service conditions. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 119 B.12 Equivalent Noise Degradation (END) Figure B.13: Test set-up for END measurement Procedure for the measurement of one point in figure B.13: 1) Measure the power of the DVB signal with a power metre. If this is not possible due to signals in the neighbouring channels, use a calibrated spectrum analyser. 2) Remove the wanted input signal and terminate the input. 3) Add noise to obtain the same level on the spectrum analyser. Now C/N = 0 dB. 4) Add the wanted input signal and increase the attenuation of the noise until a BER of 10-4 is measured. The value, for which the attenuation was increased, is the C/N for the given BER. 5) END is the difference between the measured C/N and the theoretical value of C/N for a BER of 10-4. Proposed settings for the spectrum analyser: RBW = 30 kHz, VBW < 300 Hz. B.13 BER vs. Eb/N0 The BER versus Eb/N0 will be measured using the test set-up described above. C/N measurements can be converted to Eb/N0 using the following formula: (m) 10log N C /N E 0 b 10 − = NOTE: For consideration of FEC overhead, see also clauses 7.5, 8.2, G.5, G.6 and G.7. B.14 Equalizer specification High order modulations such as 64 QAM are very sensitive to distortions. The eye aperture is so small that any perturbation can seriously disturb the reception of the signal. In the case of the DVB modulation formats, this problem is increased by the low value of the roll-off factor (0,15). In a real network, if no special processing is carried out in the receiver, the eyes appear completely closed, and no synchronization is possible. This is why all cable receivers, professional or not, are equipped with equalizers. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 120 Some of the most common impairments met on cable networks are echoes due to equipment impedance mismatching, or filtering effects. These impairments appear as perturbations of the frequency response (or impulse response) of the channel, and are corrected by the equalizer which is a form of adaptive filter. Equalizers are very efficient for linear distortions, but cannot combat those of a non-linear nature. They combat fixed frequency interference, which is equivalent to intermodulation products of analogue television signals. Equalizers have a large influence on the clock or carrier recovery systems, since these can use equalized signals. Thus the overall behaviour of the receiver depends on the performance of the equalizer. Most of the measurements specified in the present document are carried out after equalization. The first reason is that the signal is too impaired before equalization to obtain meaningful measurement results. Moreover, as most of the distortion at that point would be removed in any practical receiver, such measurements may not be relevant. The consequence of this is that measurement results are dependent on the equalizer response. This also means that equipment with different equalizer architectures will have different performance characteristics. This situation is not acceptable, and has led to the specification of the equalizer. The specification of an equalizer is a difficult task, because there is a large number of types of equalizer, due to the range of algorithms for the updating of coefficients, and the different filter architectures (time based, frequency based, recursive or non-recursive). In addition, the performance of future equipment should not be limited by any specification here. This is why a convenient solution is to specify the overall performance of the receiver as regards a perturbation typically corrected by the equalizer, specifically - echoes. The specification has to be defined so that the reference perturbation does not affect the measurement. The minimum level of perturbation that the equalizer will have to correct can then be defined. A solution is to set the minimum level of an echo that will not degrade the equivalent noise degradation of the incoming signal of more than 1 dB. This measurement is carried out for the worst case phase shift of the echo. Figure B.14: Specification of an equalizer Figure B.14 gives a possible equalizer specification which is subject to verifications in real systems. In some cases, when the likely response of a consumer receiver to network signals is studied, it is appropriate to have an equalizer in the measurement equipment whose performance is close to that of the consumer receiver. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 121 B.15 BER before Viterbi decoding This measurement should be based on the I and Q signals at interface T. If an external measurement device is used the signals at interfaces T and V are needed. The set-up is equivalent to figure B.9. B.16 Receive BER vs. Eb/N0 The measurement is based on transmission of Null packets as defined in clause A.2. At the receiving site noise is added at one of the interfaces N, P or R. The spectrum analyser is used for checking that the normal noise level is well below the added noise. The measurement itself is done either within the receiver or at one of the interfaces T, V or Y depending whether BER before Viterbi, after Viterbi or after RS should be evaluated. In case of interface Y, RS decoding should be deactivated in order to reduce the duration of the measurement. Figure B.15: Test set-up for BER vs. Eb/N0 measurement B.17 IF spectrum The output of the modulator should be directly connected to the spectrum analyser. In addition it is also possible to use a (calibrated) splitter. Figure B.16: Test set-up for IF spectrum measurement ETSI ETSI TR 101 290 V1.4.1 (2020-06) 122 Annex C: Measurement parameter definition C.1 Definition of Vector Error Measures Modulation Error Ratio (MER) is defined as: ( ) ( ) ( ) ( ) dB Q I Q I dB Q I Q I MER N j j j N j j j N j j j N j j j + + × = + + × = = = = = 1 2 2 1 2 2 10 1 2 2 1 2 2 10 log 20 log 10 δ δ δ δ Error Vector Magnitude (EVM) is defined as: ( ) % 100 1 2 max 1 2 2 × + = = S Q I N EVM N j j j RMS δ δ Where I and Q are the ideal co-ordinates, δI and δQ are the errors in the received data points. N is the number of data points in the measurement sample. Smax is the magnitude of the vector to the outermost state of the constellation. C.2 Comparison between MER and EVM To compare the two measures it is easier to write them both as simple ratios, clearly the use of decibels and percentages is not central to the definition. Taking MER first, the simple voltage ratio (MERV) is: ( ) ( ) + + = = = N j j j N j j j V Q I Q I MER 1 2 2 1 2 2 δ δ and multiplying both numerator and denominator by N 1 gives: ( ) ( ) + + = = = N j j j N j j j V Q I N Q I N MER 1 2 2 1 2 2 1 1 δ δ ETSI ETSI TR 101 290 V1.4.1 (2020-06) 123 Now looking at EVM as a simple voltage ratio (EVMV), the following can be written: ( ) max 1 2 2 1 S Q I N EVM N j j j V = + = δ δ EVM and MER are related such that: ( ) max max 1 2 2 / 1 1 S S V S Q I N EVM MER rms N j j j V V = = + = × = or V MER EVM V V × = 1 If the peak to mean voltage ratio, V, is calculated over a large number of symbols (10 times the number of points in the constellation is adequate if the modulation is random) and each symbol has the same probability of occurrence then it is a constant for a given transmission system. The value tends to a limit which can be calculated by considering the peak to mean of all the constellation points. Table C.2 lists the peak- to-mean voltage ratios for the DVB constellation sizes. Table C.1: Peak-to-mean ratios for the DVB constellation sizes QAM format Peak-to-mean voltage ratio (V) 16 1 341 32 1 303 64 1 527 C.3 Conclusions regarding MER and EVM MER and EVM measure essentially the same quantity and easy conversion is possible between the two measures if the constellation is known. When expressed as simple voltage ratios MERV is equal to the reciprocal of the product of EVMV and the peak-to-mean voltage ratio for the constellation. MER is the preferred measurement for the following reasons: - The sensitivity of the measurement, the typical magnitude of measured values, and the units of measurement combine to give MER an immediate familiarity for those who have previous experience of C/N or SNR measurement. - MER can be regarded as a form of Signal-to-Noise ratio measurement that will give an accurate indication of a receiver's ability to demodulate the signal, because it includes, not just Gaussian noise, but all other uncorrectable impairments of the received constellation as well. - If the only significant impairment present in the signal is Gaussian noise then MER and SNR are equivalent. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 124 Annex D: Exact values of BER vs. Eb/N0 for DVB-C systems Exact values of BER vs. Eb/N0 for DVB-C systems (see figure 7.2). Table D.1: Exact values of BER vs. Eb/N0 for DVB-C systems Eb/N0 Pb 10 0,025 48 10,5 0,020 72 11 0,016 46 11,5 0,012 74 12 0,009 582 12,5 0,006 981 13 0,004 909 13,5 0,003 319 14 0,002 147 14,5 0,001 323 15 0,000 771 6 15,5 0,000 423 5 16 0,000 217 1 16,5 0,000 103 1 17 4,499e-005 17,5 1,783e-005 18 6,351e-006 18,5 2,006e-006 19 5,537e-007 19,5 1,314e-007 20 2,634e-008 20,5 4,365e-009 21 5,846e-010 21,5 6,166e-011 22 4,974e-012 This assumes that the relationship between BER and Symbol Error Rate (SER) is given by the formula: SER m BER × = 1 ETSI ETSI TR 101 290 V1.4.1 (2020-06) 125 Annex E: Examples for the terrestrial system test set-ups E.0 Introduction Due to the essential differences in the modulation method used for the terrestrial system some of the test methods are different with respect to those used for cable and/or satellite. Even if not demonstrated in the diagrams of this clause and also not mentioned in the explanations, the receiver may be a part of the measurement device. In this case all the interfaces defined in figure 9.2 are internal ones, which the measurement device has access to. E.1 RF frequency accuracy E.1.0 General See clause 9.1. DVB-T Tx Reference Signal Source Spectrum Analyser DUT L, M Figure E.1: RF frequency accuracy set-up The measurement is to be done with a spectrum analyser. The signal can be picked up at interface L (IF) or M (RF), eventually by means of an aerial, or at interface N, if the received signal can be maintained stable enough for the measurement purposes, and applied to a spectrum analyser. Care should be taken at interfaces L or M not to overdrive the maximum allowed input signal for the spectrum analyser. E.1.1 Frequency measurements in DVB-T (void) Table E.1 (void) Figure E.2: Examples of 8k centre channel measurement… (void) E.1.2 Measurement in other cases (void) Table E.2 (void) Figure E.3: Examples of 8k carrier k = 48… (void) ETSI ETSI TR 101 290 V1.4.1 (2020-06) 126 E.1.3 Calculation of the external pilots frequency when they do not have continual phase. (void) Table E.3 (void) Figure E.4: Examples of 2k carrier k = 1 704… (void) Figure E.5: Examples of 8k carrier k = 6 816… (void) Figure E.6: Examples of 2k carrier k = 0… (void) Figure E.7: Examples of 2k carrier k = 0… (void) Table E.4 (void) E.1.4 Measuring the symbol length and verifying the Guard Interval (void) Table E.5 (void) Figure E.8: Examples of 8k carrier k = 6 813… (void) Table E.6 (void) Table E.7 (void) Table E.8 (void) Figure E.9: Examples of 2k carrier k = 1 701… (void) E.1.5 Measuring the occupied bandwidth, and calculation of the frequency spacing and sampling frequency The occupied bandwidth depends directly from the frequency spacing and this from the sampling frequency. If the frequency of the external pilots is known, see above on how to measure them, then the related values may be calculated as per table below. Denoting the outermost pilot frequencies as FL and FH appropriately the occupied bandwidth is OB = FH _ FL. The number of carriers K, and for 2k mode K-1 = 1 704 while for 8k mode K-1 = 6 816. Table E.9 Calculated value Nominal value (8 MHz Channels) 8k mode 2k mode 8k mode 2k mode Occupied bandwidth FH - FL 7,60714285714285714285714285714286… MHz Frequency Spacing (FH - FL)/6 816 (FH - FL)/1 704 1 116,0714285…Hz 4 464,2857142…Hz Useful duration 6 816/(FH - FL) 1 704/(FH - FL) 896 μs 224 μs Centre channel 1st IF (FH - FL) × 4 096/(K-1) (FH - FL) × 1 024/(K-1) 4,57142857142857142857142857142857…MHz Sampling Frequency (FH - FL) × 16 384/(K-1) (FH - FL) × 4 096/(K-1) 18,2857142857142857142857142857143…MHz NOTE: The long periodic decimals have been calculated using the Calculator facility from Windows, and have been left here as resulted from copying through the clipboard, as a matter of curiosity only. Values in italics are approximate values. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 127 Table E.10 Nominal value (7 MHz Channels) Nominal value (6 MHz Channels) 8k mode 2k mode 8k mode 2k mode Occupied bandwidth 6,656250 MHz 5,70535714285714285714285714285842… MHz Frequency Spacing 976,5625 Hz 3 906,25 Hz 837,053571428571…Hz 3 348,2142857142…Hz Useful duration 1 024 μs 256 μs 1 194,666666… μs 298,666666… μs Centre channel 1st IF 4 MHz 3,42857142857142857142857142857334…MHz Sampling Frequency 16 MHz 13,7142857142857142857142857142934…MHz E.2 Selectivity See clause 9.2. DVB-T Test transmitter CW Signal Generator DVB-T Rx BER Monitor N W, X DUT Figure E.10: Selectivity E.3 AFC capture range See clause 9.3. D V B -T R x M P E G -2 T S A n a ly s e r D U T N Z D V B -T T e s t tra n s m itte r Figure E.11: AFC capture range E.4 Phase noise of Local Oscillators (LO) E.4.0 General See clause 9.4. The measurement can be done with a spectrum analyser. As the spectrum shape of the phase noise sidebands of any Local Oscillator (LO) used in the process of up/down conversion could be very different depending on factors such as the type of crystal cut, the filter of the PLL, the noise of the active devices involved, etc. it is not convenient to integrate the spectrum of a sideband to reflect a single measured number which could not have meaning at all. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 128 However, samples at certain offsets of the oscillator signal could have more meaning, as indicated in clause 9.4. In each case of Common Phase Error (CPE) and Inter-Carrier Interference (ICI), 3 frequencies at each side of the oscillator signal should be measured. In order to make the measurement as accurate in frequency as possible, the spectrum analyser should be set to the minimum resolution filter available, and should be, at least, as low as 1 kHz for the 2 k system and 300 Hz for the 8 k system. In order to average the noise, the video filter should be activated with a value of at least 100 times narrower than the resolution filter used. The measured values should be normalized to a 1 Hz bandwidth. Should the spectrum analyser used not have the 1 Hz normalization capability, it can be done manually with the following criterion: For example: carrier frequency: 36 MHz fm = 10 kHz (represents any of the required offsets fa, fb or fc) ΔB = Equivalent Noise Bandwidth (ENB) of the resolution bandwidth filter: 270 Hz video bandwidth: 10 Hz or 30 Hz NOTE 1: The spectrum analysers typically use near Gaussian filters for the resolution bandwidth with a 20 % tolerance. The Equivalent Noise Bandwidth (ENB) is equal to the bandwidth of the filter measured at -3,4 dB, (by actually measuring the filter of the spectrum analyser, the 20 % tolerance factor is eliminated). Then the following conversion to 1 Hz bandwidth can be applied: ( ) dB B dBm B in power noise Pn 5,2 log 10 _ _ _ 10 + Δ − Δ ≅ in [dBm/Hz] NOTE 2: The 2,5 dB term accounts for the correction of 1,05 dB due to narrowband envelope detection and the 1,45 dB due to the logarithmic amplifier. E.4.1 Practical information on phase noise measurements This example from the works of AC106 VALIDATE Project and taken from the DTG D book, shows a recommended mask for phase noise measurements that is valid for local oscillators and is considered to cover safe limits for both CPE and ICI phase errors in the 2k mode of DVB-T. The following values are recommended. Table E.11: Frequency offsets for phase noise measurements fa fb fc fd Frequency 10 Hz 100 Hz 3 kHz 1 MHz Limits La to Ld -55 dBc/Hz -85 dBc/Hz -85 dBc/Hz -130 dBc/Hz ETSI ETSI TR 101 290 V1.4.1 (2020-06) 129 Figure E.12: Example for phase noise mask The total phase noise in the signal is the cumulative effect of all local oscillators (L.O.) that are used in the signal path. Clause A.4 can be seen for additional information on phase noise measurements. E.5 RF/IF signal power E.5.0 General See clause 9.5. The signal power can be measured directly at the interfaces K, L, M, N or P or by using a calibrated splitter. Care should be taken at interfaces L or M not to overdrive the maximum allowed input signal for the spectrum analyser or power metre. The shoulders of the spectrum should not be accounted for in the measurement of power because they do not represent any useful power conveying information. The shoulders are unwanted results of the FFT process and also due mainly to non-linearity of the practical implementations. E.5.1 Procedure 1 (power metre) An spectrum analyser is used with an integrating routine which can measure the mean power along frequency slots covering the overall part of the spectrum to be measured (this capability is currently available in several spectrum analyser on the market). In this case the values to be supplied to such a routine or to be used if manual undertaken of the measurement is wanted are: 1) Centre frequency of the spectrum: if possible as calculated under measurement E.2; 2) Spectrum bandwidth of the signal: 7,61 MHz for an 8 MHz channel system. A B C fa fb fc -La -Lb & Lc -Ld 0 dB 0 Hz Frequency offsetts Carrier 1.- Possible mask for phase noise measurements. The axis are not to scale, see table for values. D fd A B C fa fb fc -La -Lb & Lc -Ld 0 dB 0 Hz Frequency offsetts Carrier 1.- Possible mask for phase noise measurements. The axis are not to scale, see table for values. D fd ETSI ETSI TR 101 290 V1.4.1 (2020-06) 130 E.5.2 Procedure 2 (spectrum analyser) With the above considerations in mind, it would be very difficult to use an exact square filter for the measurement with a power sensor, however a good approximation should be obtained if a filter is used which can even take in account part of the shoulders in the measurement. For measuring with a thermal power sensor such an appropriate filter should be used. Figure E.13: Test set-up for RF/IF power measurement E.6 Noise power E.6.0 General See clause 9.6. Typically all the power present in a channel which is not part of the signal can be regarded as unwanted noise. It can be produced from different origination and be of the form of random noise (thermal), pseudo-random (digitally modulated interfering carriers) or periodic (Continuous Waves CW or narrowband interference), the first two are called non-coherent and the periodic ones are termed as coherent. In this measurement, all different types of noise are measured simultaneously, and the measured result can be termed as unwanted power. For doing this measurement the signal should be switched off. The measurements can be done at interface N (RF level) or at interface P (IF level). Noise level can be measured with a spectrum analyser or any other appropriate device. The same bandwidth considerations and methodology used in clause E.6 apply to this measurement in both cases, using a power metre and a spectrum analyser. Figure E.14: Test set-up for out-of-service noise power measurement E.6.1 Procedure 1 Exactly equal to the above preferred procedure for signal power, clause E.6, but understanding that the signal for this channel under measurement has been switched off. E.6.2 Procedure 2 Using a power metre as in the alternate procedure above in clause E.6, using the same filter and with the channel signal off. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 131 E.6.3 Procedure 3 If the noise floor in all bandwidth of interest is flat, it would be possible to measure the noise power at any frequency point inside the channel bandwidth and normalize the value to the nominal bandwidth of (n-1) × fSPACING (7,61 MHz for 8 MHz channels 6,66 MHz for 7 MHz channels). If the spectrum analyser does not have normalization routine to the wanted bandwidth the following procedure can be used. In order to average the noise, the video filter should be activated with a value of at least 100 times narrower than the resolution filter used, this resolution bandwidth filter should be chosen to be as wide as possible in order to average as much spectrum of the channel as possible, but not exceeding such bandwidth (e.g. 7,61 MHz), the equivalent noise bandwidth ΔB (MHz) of the filter should be known by the specifications given by the manufacturer, or measured following manufacturer indications. The noise power measured can be normalized to the wanted bandwidth using the following formulae: Noise power (dB) = Measured level (dB) + 10 log10 (7,61/ΔB) + 2,5 dB (for 8 MHz channels) If the spectrum analyser has a routine to normalize to 1 Hz, (this use to include the 2,5 dB correction) but not able to normalize to the wanted bandwidth, the following conversion can be applied: Noise power (dB) = Measured level (dB/Hz) + 10 log10 (7,61 × 106) = Measured level (dB/Hz) + 68,8 dB (for 8 MHz channels) E.6.4 Measurement of noise with a spectrum analyser Care should be taken when the measured noise has a display level close to the display level of instrument noise, (less than 10 dB), because an additional proximity factor should be applied. This is typically done automatically in some instruments available in the market. If this is not available in the instrument, it is necessary to subtract a correction factor CF from the noise level measured, the following correction table can be used. Table E.12: Correction Factor (CF) for measured noise level D (dB) CF (dB) 0,5 8,63 1 6,87 1,5 5,35 2 4,33 3,01 3,01 4 2,2 5 1,65 6 1,26 7 0,98 8 0,75 9 0,58 10 0,46 D is the distance in display level between the instrument noise (no signal applied to the input) and measured noise level (with no change in the settings). Notice that below 2 dB of D, the reliability of the result after applying the CF is under question due to the uncertainty of the measurement and the corresponding big value of CF to be subtracted. E.7 RF and IF spectrum See clause 9.7. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 132 To be defined after some practical experience is achieved. E.8 Receiver sensitivity/dynamic range for a Gaussian channel See clause 9.8. Figure E.15: Receiver sensitivity/dynamic range for a Gaussian channel E.9 Equivalent Noise Degradation (END) E.9.0 General See clause 9.9. N, P, S W , X DVB-T Tx DVB-T Rx Noise Generator BER M onitor Figure E.16: Equivalent Noise Degradation (END) All measurements of performance parameters are carried out by using a dummy load which provides a return loss for the wanted channel which is low enough not to influence the measurement. E.9.1 Description of the measurement method for END To improve the accuracy of the measurement, two independent noise sources are used. By this, the influence of the tolerance of the first attenuator is eliminated which could well be in the same magnitude as the wanted measurement result. N W, X DUT DVB-T Test transmitter DVB-T Rx BER Monitor ETSI ETSI TR 101 290 V1.4.1 (2020-06) 133 The following steps should be carried out to arrive at an accurate ENF value: 1) Connect the real DVB-T transmitter to the DVB-T receiver and add Gaussian noise, Ncal, to the point where the BER reaches a pre-determined value (e.g. 2 x 10-4 after Viterbi decoding). Ncal does not have to be measured. No channel noise, Nch, should be added. The C/N at the input to the receiver (Interface C) is therefore C/(Ntx + Ncal). 2) Replace the real DVB-T transmitter by the ideal one (disconnect Ntx in figure E.17). The C/N at Interface C is now somewhat higher (C/Ncal), since Ntx is no longer present. The BER is therefore now lower than the predetermined value. 3) Add Gaussian channel noise, Nch, to the point where the BER has reached its predetermined value again. The C/N at interface C is now C/(Nch + Ncal). 4) Measure the value of C/Nch at Interface B. Figure E.17: ENF measurement scheme Since both C/(Ntx + Ncal) and C/(Nch + Ncal) lead to the same BER, Nch can be identified with Ntx and be regarded as an estimate of Ntx. The ENF is defined to be 10 10log(Ntx/C). The estimated ENF value is similarly 10 10log(Nch/C) As long as all distortions of a DVB-T transmitter can be well approximated by the Gaussian noise, Ntx, the ENF measurement, as described above, should be completely independent of both the DVB-T mode and the receiver characteristics. For highest measurement accuracy the measurement should however preferably be done using the (non-hierarchical) mode requiring the highest C/N, i.e. 64-QAM R=7/8. In practice, there might however be selective effects such as amplitude ripple and spurious signals within the useful bandwidth. In these cases the ENF will in principle be better (= a more negative value) when stronger code rates are used (such as R = 1/2 or 2/3) than when weaker codes are used (such as R = 5/6 or 7/8). Whether this difference is measurable or not remains to be seen. It is therefore recommendable to measure the ENF also for the other code rates. If there is negligible difference between the ENF figures for the different code rates, this will imply that there are few selective effects and/or that these effects can be well approximated by Gaussian noise. If however there is a significant difference in ENF figures this implies that the ENF (and hence END) is code rate dependent. In such a case the ENF value to be used (either by itself or for the calculated END) should preferably be the one measured with the same code rate as the DVB-T transmitter will be used with by the network operator. + + + Ideal DVB-T Tx Ncal Nch Ntx Unknown DVB-T Rx Real DVB-T Tx Fixed BER=2 x 10-4 after Viterbi or RS decoding C Interface A Interface B Interface C ETSI ETSI TR 101 290 V1.4.1 (2020-06) 134 E.9.2 Conversion method between ENF and END Let (C/N)min, theory be the minimum C/N requirement for a DVB-T mode given by ETSI EN 300 744 [i.9]. Assume an implementation loss of 3,0 dB for all modes. Let X = (C/N)min, real be the corresponding minimum required C/N for a DVB-T mode. X = (C/N)min, real = (C/N)min, theory + 3,0 dB END can be calculated from ENF by the formula: END = -10 10log(10 -X/10 -10 ENF/10) - X Example: X = 19,5 dB (64QAM, R= 2/3) ENF = -30,0 dB END = -10 10log(10 -19,5/10 -10 -30,0/10) - 19,5 dB = 0,41 dB E.10 Linearity characterization (shoulder attenuation) E.10.0 General Figure E.18: Test set-up for "linearity characterization" E.10.1 Equipment (1) OFDM signal source (interface K or L of DVB-T transmitter); (2) attenuator, possibly adjustable in 0,1 dB (max. 0,5 dB) steps. Optional, see clause E.10.2, remark (d); (3) transmitter under measurement; (4) power attenuator; (5) directive coupler or attenuator, see clause E.10.2, remark (a); ETSI ETSI TR 101 290 V1.4.1 (2020-06) 135 (6) spectrum analyser; (7) attenuator, possibly adjustable. Optional, see clause E.10.2, remark (c); (8) power metre. Optional, see clause E.10.2, remark (a). E.10.2 Remarks and precautions a) Power metre (8) can be useful to verify and monitoring the output power of the transmitter (3) and for the calibration process. If power metre (8) is not available, the directive coupler (5) can be replaced by an opportune attenuator connected to the spectrum analyser (6). b) Care should be taken in the choice of the power attenuator (4) in terms of max. admitted power. c) Care should be taken in the choice of all attenuators (and directive coupler) to prevent damage to test-set equipment. For example, the function of the optional attenuator (7) is to protect the probe of the power metre. The attenuator (7) can also be useful for other measurements and, for example, be connected in a chain to the receiver. d) Pay attention to the admitted power at the IF (or RF) input of the transmitter, in order to obtain a proper working point. Optional attenuator (2) can be used for this purpose. E.10.3 Measurement procedure (example for UHF channel 47) Step 1: Select the centre frequency of spectrum analyser in the middle of the UHF channel (i.e. 682 MHz for channel 47). Verify the output power level using an high resolution BW (3 MHz or 5 MHz) and compare with the value obtained by the power metre (if available). Step 2: Select the centre frequency of spectrum analyser at the end of the UHF channel (i.e. 686 MHz for channel 47). Step 3: Select an adequate span (for example 2 MHz). Step 4: Select the resolution BW (10 kHz is adequate for 2 k and 8 k mode) and adjust levels. Video BW is of the same order. Step 5: Measure the power level at 300 kHz and 700 kHz from upper edge of the DVB-T spectrum and proceed as indicated in clause 9.10. Last DVB-T carrier is at approximately +3,8 MHz from the centre of the UHF channel: then, for channel 47, the two measurement points are at 686,1 MHz and 686,5 MHz. Step 6: Repeat steps from 2 to 5 for the lower edge of the spectrum. Step 7: The worst case value of the upper and lower results is the "shoulder attenuation" (dB). NOTE: The value obtained should be joined up with the used mode (2 k or 8 k) of the OFDM source. If available, the "maximum-hold" function of the spectrum analyser can help to carry out the measurement. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 136 685 685,5 686 686,5 687 Power [dBm] Res. BW 10 kHz Video BW 10 kHz Span 2 MHz Shoulder attenuation +700 kHz +500 kHz +300 kHz End of UHF channel 47 Last DVB-T carrier (at approx. 685,8 MHz) DVB-T spectrum (max. value) ref. 0 kHz DVB-T spectrum Frequency [MHz] Figure E.19: Example with the upper edge of the DVB-T spectrum in UHF channel 47 E.11 Power efficiency Figure E.20: Power efficiency E.12 Coherent interferer Connect a suitable spectrum analyser to interface N. DVB-T Tx Mains Power Meter RF Power Meter A M ETSI ETSI TR 101 290 V1.4.1 (2020-06) 137 E.13 BER vs. C/N by variation of transmitter power Figure E.21: BER vs. C/N by variation of transmitter power Adjust signal level at receiver input to the same value for different Tx output power values by attenuator. The results of this measurement can be put in diagrams, such as: - BER vs. C/N for constant Pout; - BER vs. Pout for constant C/N; - BER vs. Pout for constant noise power. E.14 BER vs. C/N by variation of Gaussian noise power Figure E.22: BER vs. C/N by variation of Gaussian noise power DVB-T Tx Noise Generator RF Power Meter M PRBS Generator E, F BER Test Set DVB-T Test Receiver U, V N, P, R N ETSI ETSI TR 101 290 V1.4.1 (2020-06) 138 E.15 BER before Viterbi (inner) decoder See clause 9.15. NOTE: For the measurements described in clauses 9.15, 9.16, 9.17, 9.18 and 9.19 dedicated measurement instruments are envisaged. E.16 Overall signal delay Void. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 139 Annex F: Specification of test signals of DVB-T modulator F.1 Introduction In order to compare simulated data within a DVB-T modem it is necessary to specify test points, signal formats and a subset of modes. The present document contains the specifications of how to do this. The present document should be accurate enough to enable comparison of simulated data at different points within the modulator. F.2 Input signal Figure F.1: Input test sequence generator for DVB-T modulator The number of bits in a super-frame is depending on the actual DVB-T mode. The maximum number of Reed- Solomon/MPEG-2 packets in a super-frame is 5 292. This corresponds to 7 959 168 input bits that is shorter than a maximum length sequence of length 223-1 = 8 388 607. The input test sequence to the modulator can therefore be generated by a shift register of length 23 with suitable feedback. The generator polynomial should be 1 + x18 + x23. The PRBS data on every 188 byte is replaced by the sync byte content, 47 HEX. This means that during the sync bytes the PRBS generator should continue, but the source for the output is the sync byte generator instead of the PRBS generator. The input test sequence starts with a sync byte as the first eight bits, and the initialization word in the PRBS generator is "all ones". The PRBS generator is reset at the beginning of each super-frame. The test sequence at the beginning of each super-frame starts with: 0100 0111 0000 0000 0011 1110 0000 0000 0000 1111 1111 1100 (first byte is sync byte 47 HEX). The corresponding HEX numbers are: 47 00 3E 00 0F FC. There are up to eight possible phases of the energy dispersal with respect to the start of the super-frame. The first sync byte in the sequence, i.e. the first 8 bits should be inverted by the energy dispersal block. The length of the input signal can in principle be arbitrary. However, it is not meaningful to have a sequence shorter than one OFDM symbol. The maximum length will in practice be limited by the amount of data. Very large data files may be difficult to handle and interchange. One super-frame is therefore regarded as the longest sequence of interest. The outer interleaver will spread data across the super-frame boundaries. The ambiguity in the output sequence caused by this is circumvented by using the second super-frame in the simulated sequence as the output signal. This means that the simulator should produce one super-frame before useful data starts to appear at the output. The file format for storing data allows for variable lengths of simulated data since the length indicator is contained in the header of the file. Simulations with different lengths can therefore be compared over the length of the shortest sequence. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 140 F.3 Test modes The file header in the test file contains information about the specific DVB-T mode used for the simulation. By reading this information a complete description of the set-up is obtained. In order to ease comparison of data and to reduce the amount of simulations necessary a set of "preferred modes" are defined. The preferred test mode for non-hierarchical transmission is: Inner code rate: 2/3; Modulation method: 64 QAM; FFT size: 8 k; Guard interval: 1/32. For hierarchical transmission the preferred mode is: Inner code rate HP: 2/3; Inner code rate LP: 3/4; Modulation method: QPSK in 64 QAM, α = 2; FFT size: 8 k; Guard interval: 1/32. F.4 Test points The simulated data can be probed at different points within the modulator. Eight test points are defined, which are related to the interfaces described in figure 9.1: 1) at input (A); 2) after mux adaptation, energy dispersal (B); 3) after outer encoder (C); 4) after outer interleaver (D); 5) after inner encoder (E); 6) after inner interleaver (F); 7) after frame adaptation (H); 8) after guard interval insertion (J). F.5 File format for interchange of simulated data F.5.0 General The file header as well as simulated data from the modem are stored as ASCII characters on files with carriage return and line feed at the end of each line. In order to interchange data it is important that the same file format be used by everyone. A file containing such data should have a header which has the following information: - text string with a maximum of 80 characters (affiliation, time, place etc.); - "printf" string used to store the data in the data section of the file; ETSI ETSI TR 101 290 V1.4.1 (2020-06) 141 - test point description; - length of data buffer; - constellation; - hierarchy; - code rate (code rate for HP); - code rate LP (not applicable for non-hierarchical transmission modes); - guard interval; - transmission mode; - simulated data (HEX or floating point). The specification for each of these entries are given in tables F.1 to F.8. F.5.1 Test point number Table F.1: Test point number Test point Interface Text contained in file header 1 A at input 2 B after MUX adaptation and energy dispersal 3 C after outer coder 4 D after outer interleaver 5 E after inner coder 6 F after inner interleaver 7 H after frame adaptation 8 J after guard interval insertion F.5.2 Length of data buffer The length indicator specifies the number of lines contained in the data section of the file which has two floating points or one two digit HEX on each line. F.5.3 Bit ordering after inner interleaver The signal at test point 4 after inner interleaver should contain data from one carrier on each line. The bit ordering should be according to table F.2. Table F.2: Bit ordering in the signal representation at test point 4, after the inner interleaver Modulation method Bit ordering Representation QPSK y0q y1q 2-digit HEX (00 to 03) 16 QAM y0q y1q y2q y3q 2-digit HEX (00 to 0F) 64 QAM y0q y1q y2q y3q y4q y5q 2-digit HEX (00 to 3F) ETSI ETSI TR 101 290 V1.4.1 (2020-06) 142 F.5.4 Carrier allocation The signal contains 1 705 or 6 817 active carriers for the 2 k and 8 k modes respectively. In order to ease comparison of different data sets the allocation of these into the FFT bins should be specified. The signal is arranged such that it is centred around half the sampling frequency. Table F.3: Carrier allocation FFT bins containing zeros FFT bins containing active FFT bins containing zeros 2 k mode 0 to 171 172 (Kmin) to 1 876 (Kmax) 1 877 to 2 047 8 k mode 0 to 687 688 (Kmin) to 7 504 (Kmax) 7 505 to 8 191 F.5.5 Scaling At test point 7 (after frame adaptation) the data should be scaled such that: "Vector length of a boosted pilot" is equal to unity. The gain factor through the IFFT should be equal to unity. This gain factor is defined as: ) ( ) ( * * = N N x x z z η where x are the complex numbers representing one complete OFDM symbol at the input of the IFFT including data carriers, pilots and null-carriers. And z is the complex signal for the corresponding OFDM symbol at the IFFT output before guard interval insertion. The number N is equal to the IFFT size (2 k or 8 k). The asterisk denotes complex conjugate. This ensures correct scaling of data at test point 8 (after guard interval insertion). F.5.6 Constellation The possible constellations are listed in table F.4. The file header should contain one of them. Table F.4: Constellations QPSK 16-QAM 64-QAM F.5.7 Hierarchy The hierarchical identifier specifies if hierarchical mode is on or off and also the alpha value in case hierarchical mode is on. For non-hierarchical transmission alpha is set to one. Table F.5 contains the possible choices and the file header should contain one of them. Table F.5 Hierarchical identifier Non-hierarchical, alpha = 1 Hierarchical, alpha = 1 Hierarchical, alpha = 2 Hierarchical, alpha = 4 ETSI ETSI TR 101 290 V1.4.1 (2020-06) 143 F.5.8 Code rate LP and HP The code rate identifier specifies the code rate for the LP and HP streams. Table F.6 contains the possible choices and the file header should contain one of them. Table F.6: Code rate identifier Code rate identifier 1/2 2/3 3/4 5/6 7/8 F.5.9 Guard interval Table F.7 contains the possible choices for the guard interval and the file header should contain one of them. Table F.7: Guard interval identifier Guard interval identifier 1/32 1/16 1/8 1/4 F.5.10 Transmission mode The transmission mode can be either 2 k or 8 k. Table F.8 contains the possible choices and the file header should contain one of them. Table F.8: Transmission mode identifier Transmission mode identifier 2 048 8 192 F.5.11 Data format The data at test point 1 to 6 are written to file using 2-digit HEX numbers with "printf" string % X\n. At test point 7 and 8 each line in the file contains real and imaginary parts with at least 6 significant decimal digits each. The real and imaginary parts and separated by at least 2 spaces. The data is written to file using "printf" with % e\n. F.5.12 Example This is an example of a print-out of a file containing the data sequence at the input for the preferred mode for non- hierarchical transmission. The text in parenthesis is just for explanation and should not be contained in the file. Stockholm, May 22, 1996, example of input data. Preferred non-hierarchical mode: %X\n (Data stored in HEX format); at input (Data at test point 1 at modulator input); 758 016 (One super-frame of data); ETSI ETSI TR 101 290 V1.4.1 (2020-06) 144 64-QAM (Constellation 64 QAM); non-hierarchical, alpha = 1 (Non-hierarchical transmission); 2/3 (2/3 inner code rate); 0 (Don't care. Code rate LP); 1/32 (Guard interval = 1/32); 8 192 (8 k IFFT size); 47 (First data byte is sync byte 47 HEX); 00 (Rest of data). ETSI ETSI TR 101 290 V1.4.1 (2020-06) 145 Annex G: Theoretical background information on measurement techniques G.0 Introduction This informative annex presents a review of the theoretical background to the measurement techniques recommended in the present document. It is an attempt to gather the most relevant background information into one location, particularly for the benefit of engineers and technicians who are new to digital modulation techniques. It is hoped that it will provide a working knowledge of the theoretical and practical issues, particularly the potential sources of ambiguity and error, to help users of the present document make valid, accurate and repeatable measurements. G.1 Overview The basic purpose of a digital transmission system is to transfer data from A to B with as few errors as possible. It follows that the fundamental measure of system quality is the transmission error rate. The transmission error rate is usually measured as the Bit Error Rate (BER), however it can also be informative to consider the error rate of other transmission elements such as bytes, MPEG packets, or m-bit modulation symbols. In practice, although a certain guaranteed minimum BER performance may be a system implementation goal, the system BER alone is not a particularly informative measurement. The most important figure of merit for any digital transmission system is the BER expressed as a function of the ratio of wanted information power to unwanted interference power (C/N). This is underlined by the fact that most of the measurements in the present document are built around this central theme of BER vs. C/N (or, equivalently, BER vs. Eb/N0). There are measurements of the individual elements (power and BER measurements). There are measurements of the difference between theoretical and ideal performance (margin and degradation measurements). There are measurements intended to help identify the sources of transmission errors (interference, spectrum, jitter and I/Q measurements). There are measurements for monitoring the consequences of transmission errors at the system level (availability, error event logging). G.2 RF/IF power ("carrier") When describing the Quadrature Amplitude Modulated (QAM) signals employed by DVB-C or the Quadrature Phase Shift Keying (QPSK) signals employed by DVB-S, it is common to refer to the modulated RF/IF signal as "carrier" (C), mainly to distinguish it from "signal" (S) which is generally used to refer to the baseband demodulated signal. Strictly, it is incorrect to describe this signal as "carrier" because QAM and QPSK (which is equivalent to 4-state QAM) are suppressed carrier modulation schemes. For OFDM, with thousands of suppressed carriers and assorted pilot tones, the label "carrier" is even more inappropriate. This is why deliberately the expression "wanted information power" is used in the clause above, and why the parameter is referred to as "RF/IF power" in the present document. However, it is clear that engineers will continue to use "carrier" as a convenient shorthand for this parameter, particularly when talking about the "carrier"-to-noise ratio. It seems futile to attempt to change this, so instead it is clearly defined what is meant by "carrier" in this context. Carrier, more accurately called RF/IF power, is the total power of the modulated RF/IF signal as would be measured by a thermal power sensor in the absence of any other signals (including noise). ETSI ETSI TR 101 290 V1.4.1 (2020-06) 146 For DVB compliant systems the QAM/QPSK passband spectrum is shaped by root raised cosine filtering with a roll-off factor alpha (α) of 0,15 for DVB-C systems, or 0,35 for DVB-S systems. For an ideal QAM/QPSK system this means that all the RF/IF power will lie in the frequency band: ( ) 2 1 ) ( S C QAM OCC f f BW × + ± = α (G.1) Equation G.1 defines the occupied bandwidth of the signal, where fC is the carrier frequency, fS is the symbol rate of the modulation, and α is the filter roll-off factor. RF/IF power (or "carrier") is the total power in this "rectangular" bandwidth, that is, with no further filtering applied. For OFDM systems the definition of occupied bandwidth is expressed differently because of the radically different modulation technique, however the principle is very similar. The OFDM "shoulders" are not considered to be wanted information power, and are not included in the RF/IF power calculation, even though the power does actually come out of the transmitter: SPACING OFDM OCC f n BW × = ) ( (G.2) where n = 6 817 (8 k mode) or 1 705 (2 k mode) and fSPACING = 1 116 Hz (8 k mode) or 4 464 Hz (2 k mode). In a real multi-signal system (e.g. a live CATV network) measurement of the RF/IF power in a single channel requires a frequency selective technique. This could employ a thermal power metre preceded by a suitably calibrated channel filter, a spectrum analyser with band power measurement capability, or a measuring receiver. Depending on the measurement technique a filter may be required to exclude the "shoulders" of a single OFDM signal. G.3 Noise level The noise level is the unwanted interference power present in the system when the wanted information power is removed. This is a less bounded quantity than the RF/IF power because there is no definitively "correct" bandwidth over which to measure the noise. The choice is to some extent arbitrary, but the "top three" choices are probably: 1) Channel bandwidth: In a channel based system such as a CATV network you could choose the channel bandwidth, for example 8 MHz, as the system noise bandwidth. This is considered by the DVB-MG to be inappropriate for C/N measurements in digital TV systems. It will result in misleadingly poor C/N ratios when the modulation symbol rate is low relative to the available channel bandwidth. It unnecessarily complicates conversion between C/N measurements made "in the channel " and "in the receiver " by introducing symbol rate dependent correction factors. 2) Symbol rate: For digital modulation employing Nyquist filtering split equally between the transmitter and receiver, the noise bandwidth of the receiver equals the symbol rate. This is considered by the DVB-MG to be appropriate for "in the receiver " C/N measurements of digital TV systems since this reflects the amount of noise entering the receiver independent of symbol rate. 3) The occupied bandwidth: For digital modulation employing Nyquist filtering the occupied bandwidth of the modulated signal is (1 + α) × fS. This is considered by the DVB-MG to be appropriate for "in the channel " C/N measurements of digital TV systems since it exactly covers the transmitted spectrum, independent of symbol rate. The DVB-MG have chosen occupied bandwidth, as defined by equation G.1, as the standard definition of noise bandwidth in DVB-C and DVB-S systems. This is primarily because "in the channel " C/N is considered to be the fundamental measurement, but also because a simple correction factor can be applied to determine the equivalent "in the receiver " C/N value. The other possibility that should be mentioned is to assume that the noise power is evenly distributed across the frequency spectrum of interest and so can be described by a single noise power density value (N0) which is the noise power present in a 1 Hz bandwidth. From this, the noise power present in any given system noise power bandwidth (BWSYS) can be obtained by simple multiplication: SYS BW N N × = 0 (G.3) ETSI ETSI TR 101 290 V1.4.1 (2020-06) 147 By talking in terms of N0 there is no need to define a noise bandwidth, but an assumption is made that the noise power spectrum is flat across the bandwidth of interest. G.4 Energy-per-bit (Eb) Trying to commission a DVB system against tight deadlines, Energy-per-bit (Eb) seems to be a rather academic concept, particularly since the directly measurable quantity is RF power. However, it is useful to understand Eb, even if only to avoid confusion when it appears in technical specifications or discussions. Historically, use of Eb arises from information theory and as part of an academic desire to normalize the performance of different modulation formats and coding schemes for comparative purposes. The Energy-per-bit is the energy expended in transmitting each single bit of information. Eb is of little practical use on its own, it is most useful in the context of a graph of BER vs. the Eb/N0 ratio - the well-known "waterfall curve" (see figures G.1 and G.2). By normalizing to an Eb/N0 ratio on the X axis, the relative performance of various complexities of digital modulation and channel coding can be compared because the scaling effects of actual signal and noise powers, number of bits-per-symbol and symbol rate are removed. It is then simply a case of comparing the bit error probability for a given ratio. Energy-per-bit can be easily translated to carrier power. Power is energy-per-second. Which can be expanded to energy-per-bit, times bits-per-symbol, times symbols-per-second. Expressed algebraically it gives: ( ) S b f M E C × × = 2 log (G.4) G.5 C/N ratio and Eb/N0 ratio The parameters that can be directly measured are RF/IF or "carrier" power (C) and noise power in a certain bandwidth (N). From these measurements the C/N ratio can be computed. With the equations above, knowledge of the other parameters (e.g. fS ) and a little algebra it is also possible to arrive at an equivalent Eb/N0 ratio. G.6 Practical application of the measurements At this point it seems that C/N (or Eb/N0) is defined, and indeed it is from an algebraic perspective. However, there is scope for endless confusion in applying these simple formulae unless the user is very clear about where the C/N or Eb/N0 ratio is being measured, and what values are being used for the subordinate parameters, most particularly the system noise bandwidth. C/N (or Eb/N0) can be measured "in the channel" or "in the receiver". The meaning of "in the channel" is fairly self- evident, "in the receiver" may need further explanation. There are typically three filtering processes present in a receiver. The first (which is optional) is a relatively broadband tuneable pre-selection simply to reduce the power presented to the receiver RF front-end. The second, usually applied at an IF, is a high-order bandpass channel selection filter to extract the desired signal with (ideally) no modification of the signal spectrum. The third is the root-raised cosine Nyquist filtering, commonly implemented in the low pass filters following the I/Q demodulation. For theoretical simplicity it is assumed that the receiver's bandwidth and band shape are defined totally by the low-pass root-raised cosine filters because the intended purpose of the other RF/IF filters is only signal pre-selection. So the receiver can be modelled as a broadband receiver with a root-raised cosine passband filter followed by I/Q demodulation. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 148 With this in mind, "in the receiver" can be seen to mean "after the bandwidth and band shape modifying effects of the receiver Nyquist filters has been taken into account". Whether artificially generating a specific C/N ratio or just measuring the existing C/N ratio it is important to understand the difference between the "in the channel" and "in the receiver" nodes. On a more practical note, graphing the BER performance of a receiver versus Eb/N0 removes the ambiguity introduced by varying noise bandwidth. If the "in the channel" Eb value is used then a certain BER curve is derived, if the slightly lower "in the receiver" Eb value is used then the Eb/N0 ratio is slightly poorer for the same BER, the curve moves to the left (closer to the theoretical curve) and the implementation loss decreases because the loss due to the receive filters is not included. An example may help to explain this. G.7 Example Creation of a signal with a specific C/N ratio in order to test the performance of an Integrated Receiver Decoder (IRD), or perhaps to degrade an incoming RF/IF signal to a specific C/N ratio in order to establish the noise margin. To do this, add broadband white Gaussian noise "in the channel " to the relatively noise free RF/IF signal. Measure (or compute) the carrier power and then adjust the noise power density to give the required noise power in the selected noise power bandwidth. Taking the following QAM system parameters as an example: Symbol rate: fS = 6,875 MHz; Filter roll-off: α = 0,15; System noise bandwidth: BWNOISE = 8 MHz; Constellation size: M = 64; Carrier power (in dB): C = -25 dBm. then: 25 − = C dBm ( ) ( ) 15 , 101 log log 10 2 10 − = × × − = S b f M C E dBm If a C/N ratio of 23 dB is wanted, then: 00 , 48 − = − = dB N C C N dBm ( ) 03 , 117 log 10 10 0 − = × − = NOISE BW N N dBm So the ratio of Carrier-to-Noise applied in an 8 MHz system bandwidth at RF/IF can be described as: 00 , 23 = N C dB 88 , 15 0 = N Eb dB ETSI ETSI TR 101 290 V1.4.1 (2020-06) 149 This signal is then passed through the receiver root-raised cosine filters. The equivalent noise bandwidth of a bandpass root-raised cosine filter is equal to the symbol rate fS. The noise power originally defined in an 8 MHz system bandwidth is reduced accordingly: 66 , 48 log 10 10 − = × + = NOISE S REC BW f N N dB (G.5) The noise power density N0 is unchanged by the receive filter: N0(REC) = N0 = -117,03 dBm. The signal power is already root-raised cosine shaped by the transmitter and so its power is only modified by the factor (1-α/4): 17 , 25 4 1 log 10 10 − = − × + = α C CREC dB (G.6) The Energy-per-bit Eb is subject to this same reduction factor: Eb(REC) = -101,32 dBm. So the ratio of Carrier-to-Noise inside the receiver can be described as: 49 , 23 = REC REC N C dB 71 , 15 ) ( 0 ) ( = REC REC b N E dB It is this received C/N (or Eb/N0) ratio that, when demodulated translates directly to a Signal-to-Noise Ratio (SNR) in the I/Q domain. In the idealized case that white Gaussian noise is the only impairment present then this also determines the Modulation Error Ratio (MER). A general formula for the C/N modification due to the receive filters can be derived: − × + = NOISE S REC REC BW f N C N C 4 1 log 10 10 α dB (G.7) and another for Eb/N0: − × + = 4 1 log 10 10 0 ) ( 0 ) ( α N E N E b REC REC b dB (G.8) For the C/N case the correction factor is dependent on filter roll-off, symbol rate and the system noise bandwidth used to define the noise power. However, if the occupied bandwidth is used as the system noise bandwidth, then equation G.7 simplifies to: + − × + = α α 1 1 4 1 log 10 10 N C N C REC REC dB (G.9) and the correction factor becomes a constant dependent on the filter α only. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 150 For DVB-C with filter α = 0,15 441 ,0 + = N C N C REC REC dB; For DVB-S with filter α = 0,35 906 ,0 + = N C N C REC REC dB. For comparison, if one were to always use the channel bandwidth (e.g. 8 MHz) as the system noise bandwidth then one should use equation G.7, the correction factor becomes symbol rate dependent, and ranges from +0,441 dB for a theoretical maximum occupancy symbol rate of 6,957 MBaud, through +0,492 dB for the example symbol rate of 6,875 MBaud, to +1,285 dB for a typical lower rate of 5,728 MBaud. For the Eb/N0 case the correction for the DVB-C standard filter roll-off of α = 0,15 the correction factor is -0,166 dB, and for the DVB-S standard filter roll-off of α = 0,35 it is -0,398 dB. It is perhaps worth mentioning that using the C/N correction formula (equation G.7) gives correction factors which suggest that the C/N is actually improved by the receive filter, but this is only because the system noise bandwidth is larger than the receiver noise bandwidth. The Eb/N0 formula (equation G.8) more accurately reflects reality, the information-to-noise ratio is actually degraded by a small amount by the receive filter, because for the filter to pass the RF signal spectrum properly at the band edges it should also pass proportionately more noise power than signal power. G.8 Signal-to-Noise Ratio (SNR) and Modulation Error Ratio (MER) When a randomly modulated QAM or QPSK carrier and the associated passband noise are demodulated, approximately half the signal power and half the noise power will be delivered into each baseband component channel (I and Q). The demodulation process will have a certain gain, but this gain factor will apply equally to the signal and to the noise so the resulting SNR in each channel will be approximately the same as the CREC/NREC ratio computed above. The vector sum of the mean I and Q signal powers ratioed to the vector sum of the mean I and Q noise powers will, at least theoretically, be exactly the same as the CREC/NREC ratio computed above. This ratio of I/Q signal power to I/Q noise power expressed in dB is the definition given in the present document for both SNR and for MER. The difference between these two measurements lines in what perturbations of the received signal are included in the computation. When the only significant impairment is noise then SNR and MER are equivalent, and are numerically equal to CREC/NREC. The relationship between CREC/NREC and C/N depends on the choice of system noise bandwidth. If the symbol rate is chosen as the system noise bandwidth (as defined in the present document clause 6.7) then the relationship is a fixed offset of a fraction of 1 dB as described above. This would appear to suggest that C/N measured in the passband can be equated directly to SNR in baseband. Unfortunately other factors should also be considered in a real system. The SNR of the source modulator, the signal amplitude dependence of the noise floor of system components, and the fact that the receiver equalizer will have the effect of translating some linear impairments into noise. The exact interrelation of these parameters is the subject of further study. G.9 BER vs. C/N As was stated in the introduction, the Bit Error Rate (BER) as a function of Carrier-to-Noise ratio (C/N) is the most important figure of merit for any digital transmission system. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 151 To evaluate the performance of modulator and demodulator realizations, measured BER values are compared against the theoretical limits of the Bit Error Probability (BEP) PB. Regarding DVB satellite and cable transmission schemes the BEP is usually determined based on the following assumptions: - the only noise present is additive white Gaussian noise; - the channel itself does not introduce any linear or non-linear distortions; - modulator and demodulator are perfect devices (no timing errors, ideal band-limiting filters). Based on these assumptions it is possible to calculate fairly accurate upper limits for BEP vs. C/N. Since C/N depends on noise bandwidth it is common practice to normalize C/N by using Eb/N0 instead, where Eb is the Energy-per-bit and N0 is the noise density. The transition from one value to the other is given by: m f BW N C N E S NOISE b × × = 0 (G.10) where BWNOISE is the equivalent noise bandwidth, fS is the symbol rate, and m is the number of bits-per-symbol, m = log2(M), where M is the number of constellation points. When applying this formula it is important to be consistent in using either the "in the channel" C/N or the "in the receiver" C/N values. If Forward Error Correction (FEC) is employed, the information rate RI is increased up to the transmission rate RT by adding the FEC information. The relation: T I C R R R = (G.11) is called the FEC rate. The transmission rate of an FEC rate 1/2 system for example will be 2 times the information rate. Therefore the "Transmission Rate" Eb/N0 will be 3 dB less than the "Information Rate" Eb/N0, provided C/N stays constant. This results from the fact that half of the available signal power is spent on FEC information. To compensate for this effect Eb/N0 should be increased by 3 dB in case of "Information Rate" BEP. In general, if the BEP should be calculated based on the information rate, Eb/N0 should be increased by 10 × log10(1/RC) dB. If the performance of different FEC schemes is to be compared for power limited channels like satellite transmission, the information rate should be used because it explicitly takes into account the signal power which is used for redundancy only, and which is therefore lost for the information itself. In case of bandwidth limited channels like cable results based on the transmission rate may be more appropriate. G.10 Error probability of Quadrature Amplitude Modulation (QAM) Each state in an M state QAM constellation represents a log2(M) = m bit symbol. For example, each state in a 64 QAM constellation represents a 6-bit symbol. When the received signal is perturbed by Additive White Gaussian Noise (AWGN) there is a probability that any particular symbol will be wrongly decoded into one of the adjacent symbols. The Symbol Error Probability PS of QAM with M constellation points, arranged in a rectangular set, for m even, is given by (see "Digital Communication" [i.38]): ( ) ( ) ( ) ( ) × − × × × − × − × × − × × × − × = 0 2 0 2 0 1 2 log 3 erfc 1 1 2 1 1 1 2 log 3 erfc 1 1 2 N E M M M N E M M M N E P b b b S (G.12) ETSI ETSI TR 101 290 V1.4.1 (2020-06) 152 where erfc(x) is the complimentary error function given by: ( ) ∞ − = x t dt e x 2 2 erfc π For practical purposes equation G.12 can be simplified by omitting the, generally insignificant, joint probability term to give the approximation; ( ) ( ) × − × × × − × = 0 2 0 1 2 log 3 1 1 2 N E M M erfc M N E P b b S (G.13) This approximation introduces an error which increases with degrading Eb/N0, but is still less than 0,1 dB for 64 QAM at Eb/N0 = 10 dB. When M is not an even number (for example M = 5 (32 QAM) or M = 7 (128 QAM), then equation G.14 provides a good approximation to the upper bound on PS ("Digital Communication" [i.38]): ( ) ( ) 2 0 2 0 1 2 log 3 erfc 1 1 × − × × − − ≤ N E M M N E P b b S (G.14) As already stated, the above equations for Symbol Error Probability are based certain simplifying assumptions which can be summarized as "the system is perfect except for the presence of additive white Gaussian noise", but within this rather generous constraint the equations for PS are exact. The corresponding Bit Error Probability (BEP) is less easily determined. It is directly related to the Symbol Error Probability (SEP) but the exact relationship depends on how many bit errors are caused by each symbol error, and that in turn depends on the constellation mapping and the use of differential encoding. Two different approaches can be found in the literature. The first one makes no assumption about the constellation mapping and is based on the probability that any particular bit in a symbol of p bits is in error, given that the symbol itself is in error (see "Digital Communication" [i.38] and see also "Satellite Communications" [i.42]). This approach leads to: ( ) S p p B P P × − = − 1 2 2 1 (G.15) The other approach assumes that an erroneous symbol contains just one bit in error. This assumption is valid as long as a Gray coded mapping is used and the BER is not too high. Under these assumptions: S B P p P × = 1 (G.16) These approaches give different results for symbols of two or more bits. The second approach is generally adopted because DVB systems employ Gray code mapping. The results tabulated in annex D are based on equations G.12 and G.16. It should be mentioned that for QAM systems DVB only employs Gray coding within each quadrant, the quadrant boundaries are not Gray coded, and the mapping is partially differentially coded. Further work is required to establish the exact PB to PS relationship for this combination of mapping and coding. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 153 G.11 Error probability of QPSK QPSK can be analysed as 4 QAM. Evaluation of the general QAM equation (G.12) for M = 4 gives: × − × = 0 0 0 erfc 4 1 1 erfc N E N E N E P b b b S (G.17) Again this can be simplified by dropping the joint probability term to give: = 0 0 erfc N E N E P b b S Using the PS to PB relationship defined in equation G.16, the expression for PB for QPSK modulation becomes: × = 0 0 erfc 2 1 N E N E P b b B (G.18) G.12 Error probability after Viterbi decoding Since it is not possible to derive exact theoretical expressions for the performance of convolutional codes, only upper bounds can be presented in this annex. The upper bound: ( ) × × × × × ≤ ∞ = 0 0 erfc 2 1 1 N E d R d w k N E P b c d d b B f (G.19) provides a good approximation for infinite precision, soft decision Viterbi decoding and infinite path history, as long as Eb/N0 is not too low (see "High-Rate Punctured Convolutional Codes for Viterbi and Sequential Decoding", IEEE Trans. Commun., vol 37 [i.39] and also see "Further Results on High-Rate Punctured Convolutional Codes for Viterbi and Sequential Decoding", IEEE Trans. Commun., vol 38 [i.40]). In equation G.19, df specifies the free distance of the used code, w(d) can be derived from the transfer function of the convolutional code or determined directly by exhaustive search in the trellis diagram of the code, Rc = k/n is the rate of the convolutional code, and Eb/N0 is given for the transmission rate. Since erfc(x) converges to zero quite quickly for increasing x only very few terms of the sum should be taken into account. Values for df and w(d) can be found in table G.1 regarding convolutional codes used in DVB satellite transmissions. The performance of convolutional codes for low Eb/N0 values can only be evaluated by simulations. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 154 Table G.1: Free distance and weights w(d) for DVB convolutional codes Code Rate Rc 1/2 2/3 3/4 5/6 7/8 free distance df 10 6 5 4 3 w(df) 36 3 42 92 9 w(df+1) 0 70 201 528 500 w(df+2) 211 285 1 492 8 694 7 437 w(df+3) 0 1 276 10 469 79 453 105 707 w(df+4) 1 404 6 160 62 935 791 795 1 402 089 w(df+5) 0 27 128 379 546 7 369 828 17 888 043 w(df+6) 11 633 117 019 2 252 394 67 809 347 221 889 258 w(df+7) 0 498 835 13 064 540 609 896 348 2 699 950 506 w(df+8) 2 103 480 75 080 308 5 416 272 113 32 328 278 848 w(df+9) 8 781 268 427 474 864 47 544 404 956 382 413 392 069 G.13 Error probability after RS decoding A Reed-Solomon (RS) code is specified by the number of transmitted symbols (note) in a block N and the number of information symbols K (see "Error Control Coding Handbook" [i.41]). Such a code will be able to correct up to t = (N-K)/2 symbol errors. As for DVB transmission N = 204 and K = 188 are used. Therefore up to t = 8 erroneous symbols can be corrected. NOTE: Whereas the symbols mentioned in context with QAM and QPSK are related to the modulation the symbols mentioned here are just a group of bits. The probability PBLOCK of an undetected error for a block of N symbols as a function of the error probability of the incoming symbols PSIN is given by: ( ) + = − − × × = N t i i N SIN i SIN BLOCK P P i N P 1 1 (G.20) From this expression the probability: ( ) i N SIN i SIN N t i i S P P i N N P − + = − × × × × = 1 1 1 β (G.21) of a symbol error can be derived, where βi is the average number of symbol errors remaining in the received block given that the channel caused i symbol errors. Of course βi = 0 for i ≤ t. When i > t, βi can be bounded by considering that if more than "t" errors occur, a decoder which can correct a maximum of "t" errors will at best correct "t" of the errors and at worst add "t" errors. So: t i t i i + ≤ ≤ − β (G.22) is the possible range for βi. A good approximation is βi = i but also βi = t + i is used, which can be regarded as an upper limit. From G.21 the BEP can be calculated by using G.15 or G.16. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 155 G.14 BEP vs. C/N for DVB cable transmission For DVB transmission in cable networks, QAM-M systems with M = 16, 32 and 64 are specified. To evaluate the BEP after RS decoding, the following steps should be done: a) calculate the SEP after QAM demodulation by using (G.12) or (G.14); b) transform the SEP into a BEP by applying (G.15) or (G.16) to the SEP with p = m; c) transform the resulting BEP into a SEP with p = 8 by using (G.15) or (G.16); d) use (G.21) to calculate the SEP PS after RS decoding; e) apply (G.15) or (G.16) to PS with p = 8 to determine the final BEP; f) if the BEP should be based on the information rate, shift the curve by: - 10 × log10(204/188) = 0,35 dB to the right. If just the BEP before Reed-Solomon is needed, only the first two steps are necessary. In this case there is no difference between information rate and transmission rate. All bits are regarded as information bits. The limits before and after Reed-Solomon decoding for M = 64, βi = i and Eb, based on the transmission rate, are presented in figure G.1. Figure G.1: BER for QAM-64 DVB cable transmission G.15 BER vs. C/N for DVB satellite transmission For satellite transmission three different BEPs are possible: - BEP after QPSK demodulation; - BEP after Viterbi decoding; 64-QAM Demodulation and Reed Solomon Decoding 1E-13 1E-11 1E-09 1E-07 1E-05 1E-03 13 15 17 19 21 23 Eb/N0 [dB] Bit Error Rate (BER) QAM Demodulation Reed Solomon Decoding ETSI ETSI TR 101 290 V1.4.1 (2020-06) 156 - BEP after Reed-Solomon decoding. The BEP after QPSK can be derived from equation (G.17). There is no difference to be made between information bit rate and transmission bit rate. The BEP after Viterbi decoding is expressed by equation (G.18). The result is based on the information rate, because RC is taken explicitly into account in equation (G.18). BEP after Reed-Solomon decoding can be derived from the above result by applying the following steps to the outcome of equation (G.18): a) transform the BEP after Viterbi decoding into a SEP by using equation (G.15) or (G.16) with p = 8; b) use equation (G.17) to determine the SEP after Reed-Solomon decoding; c) apply equation (G.15) or (G.16) to PS with p = 8 to determine the final BEP; d) if the BEP should be based on the information rate, shift the curve by: 10 × log10(204/188) = 0,35 dB to the right. The results for the three different BEPs and for all the different code rates Rc are presented in figure G.2. Figure G.2: BER for DVB satellite transmission Since it is common practice in satellite transmission to refer the results to the information rates the curves for BEP after Reed-Solomon decoding have been shifted accordingly. The equation (G.19) is only valid for low error rates. Despite the fact that for decreasing Eb/N0 the BER should converge to 1/2 the results according to (G.19) will possess a singularity for Eb/N0 = 0. This behaviour is especially pronounced for Rc = 7/8, where the assumption of a low error rate is not fulfilled above a BEP of 10-4. QPSK Demodulation, Viterbi and Reed Solomon Decoding 1E-12 1E-10 1E-08 1E-06 1E-04 1E-02 0 2 4 6 8 10 12 14 Eb/N0 [dB] Bit Error Rate (BER) 1/2 2/3 3/4 7/8 5/6 QPSK-Demodulation Viterbi Reed- Solomon 1/2 2/3 3/4 7/8 5/6 ETSI ETSI TR 101 290 V1.4.1 (2020-06) 157 G.16 Adding noise to a noisy signal In a practical situation where noise is deliberately added to real signal in order to create a specific C/N ratio for measurement purposes, it is important to realize that there are two fundamental assumptions implicit in this technique. The first assumption is that the input signal has a high C/N ratio and can, for practical purposes, be regarded as carrier only. The second assumption is that the input signal has a considerably better C/N ratio than the target C/N ratio. In practice noise may be added to an already noisy signal, and in this case there are accuracy issues related to the above assumptions that should be considered. First consider how noise is typically added to a signal. Figure G.3 gives a simplified block diagram. Figure G.3: Simplified block diagram of C/N test set The input is the carrier signal to be impaired. The carrier power is measured using the power metre. A broadband Gaussian noise source is then filtered and attenuated appropriately to deliver the required noise density (N0) across the frequency band of interest. The same power metre is used to set the noise power which helps ensure good C/N0 ratio accuracy, The generated noise is added to the input signal to achieve the required C/N0 ratio in the output signal. Finally, the carrier power is monitored and the power of the noise source is adjusted accordingly to maintain the required C/N0. In automated versions of this process, the user simply selects the desired C/N0 ratio. This can be entered as C/N0, but it is more typically entered as C/N which requires that the user also enters the receiver or system noise bandwidth, or it can be input as Eb/N0 which requires that the user also enters the system bit rate. From this description it is evident that it is assumed that all the measured input power is carrier and the noise power to achieve the required C/N ratio is computed accordingly. If the input already contains some noise or other carriers then this will: a) appear at the output in addition to the generated noise; b) cause the generated noise power to be too large because it is based on the C + N power at the input, not just the C power. This error is exacerbated if the input is not band limited. A formula can be derived for the actual output C/N ratio as a sum of the theoretical C/N ratio and an error term: 4 4 4 4 3 4 4 4 4 2 1 4 4 3 4 4 2 1 term error N N N N ratio N C l theoretica N C CN n c i c c actual + + × − × = 10 10 log 10 / log 10 dB (G.23) Where Nc is the noise power added due to the carrier power, Ni is the noise power already present in the input, Nn is the noise power added due to the input noise. If further manipulation of the error term are performed then an expression can be derived in terms of the fractional input and output C/N ratios. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 158 + + × = 1 1 1 log 10 10 in out in error CN CN CN CN dB (G.24) The error becomes significant if either the 1/CNin or the CNout/CNin term in the denominator moves away from zero which will happen if either the C/Nin ratio or the C/Nout to C/Nin margin is reduced. The present document gives a minimum value of 15 dB for the C/Nin ratio and for the C/Nout to C/Nin margin as a guideline figure. To meet this condition in satellite systems it is necessary to use a sufficiently large dish to get the required C/N ratio. A received C/N ratio of 20 dB or more is desirable. Alternatively, it is possible to work with higher noise signals if it is possible to measure the carrier and noise power accurately, for example by measuring carrier plus noise then switching off the carrier and measuring noise only. Equation G.23 can then be used to compensate for the errors due to the input noise. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 159 Annex H: Void ETSI ETSI TR 101 290 V1.4.1 (2020-06) 160 Annex I: PCR related measurements Void. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 161 Annex J: Bitrate related measurements J.1 Introduction J.1.1 Purpose of bitrate measurement This annex is intended to clarify a bitrate measurement technique which will allow different vendors of equipment to display the same bitrate value on their equipment when they analyse the same transport stream. The measurement technique in the present document should be applicable to the whole transport stream as well as its individual components. This should allow displays of transport stream information such as the traditional "bouncing bars" statistical multiplex display to be shown consistently on different equipment. This display is intended to dynamically show the different allocation of bitrate between different services. The intention is that the measurement should be stand-alone and non-intrusive. The measurement technique should also be easy to implement so that cost-effective designs can be introduced to large MPTS systems. It should also be scalable so that as extra precision is required, a more expensive device can be built using the same principles. The technique is also appropriate for non Transport Stream system, but the use in such systems is outside the scope of the present document. J.1.2 User Rate versus Multiplex Rate MPEG-2 transport streams are comprised of many different elements including but not limited to multiple compressed video and audio streams, teletext, table data, conditional access streams, IP data, and other private data. Each of these individual elements and the overall transport stream have data rates associated with them. The data rates can be time varying for the individual elements and the overall stream. It is of importance to define the measurement of these rates and have a common definition for these measurements. Before the measurements can be defined, the multiplexing of all the elements into a transport stream needs to be understood with regards to rate calculations. Figure J.1 depicts a general representation of the multiplexing process. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 162 Video 1 Audio 1 Video 2 ... Audio 2 PSI/SI Tables Null PID Multiplex Switch User Rates Multiplex Rates Buffers Transport Stream Packets ... ... Rate Measurement Device Figure J.1: General representation of the multiplexing process This diagram represents a number of different elements being multiplexed into a single transport stream. Before all the streams are multiplexed together they can be considered to have User rates which are established by the user (e.g. 4 Mbits/s for Video 1). It can be modelled that each element has a User data rate entering the buffer and a Multiplex rate leaving the buffer since the data is extracted directly from the buffer and placed as a complete packet in the transport stream. Over the long term average, the User and Multiplex rates should be the same, but the creation of the transport stream through the multiplex process can either increase or decrease the User rate in the actual transport stream over a specific Time Gate. For example, the video might have a 4,1 Mbits of data over a one-second Time Gate in the transport stream, but in the next one second interval it could have 3,9 Mbits. But with respect to the PTS/DTS values in the stream, the video rate as set by the user could still be 4,0 Mbits/s. The Multiplex rates will also depend upon what is actually being multiplexed together, and the measurement of the multiplex rate in the output stream will vary if different elements are combined. If only one video is being transmitted at one time and another video is being transmitted at another time, the output Multiplex rate will be different at those two times even if the User rate has not changed. The User rate for video also needs to be better understood since a single number is often given for this rate (e.g. 4 Mbits/s). This number typically means the total number of bits in a GOP multiplied by the number of GOPs per second. The actual rate of video varies with each frame. An I frame typically receives a much higher percentage of the bits compared to the B and P frames. What generally happens is that even though the I frame has significantly more data than a B frame, it will take longer to transmit this frame and the Multiplex rate can approach the User rate. This definition of User rate for video applies to both the CBR and VBR approaches. In the CBR case, the user provides one value for the rate, while in the VBR case the user provides a minimum and maximum and typically lets compression equipment vary the rate between these parameters in order to maximize video quality based on some constraints. The rate as calculated by the compression equipment is still considered a User rate since it is before the video data is multiplexed into the transport stream. Since the rates of the elements are less than or equal to the rate of the output transport stream, the positioning of these elements in the output stream is important to consider in calculating the User rate. For example, an element that generates 10 packets per second may have these packets placed at the beginning of the second, in the middle, dispersed throughout, etc. Buffer models in general restrict the packet placement but as an extreme example, it could be assumed that the packets are placed at the beginning of a second and the transport rate is 1,5040 Mbits/s. If the Time Gate of a rate measurement of this element is 0,1 s and this Time Gate started with the transmission of these packets, the first rate measurement would be 0,1504 Mbits/s. If the next measurement also uses 0,1 s of duration and starts just after the packet is transmitted, the rate would be 0,0 Mbits/s. Neither of these numbers matches the expected User rate of 0,01504 Mbits/s. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 163 A real world example for a 256 kbit/s audio stream can easily indicate differences of 2 % in the User rate versus the Multiplex rate. This audio stream has approximately 200 packets per second with each audio frame containing about 5 packets. In a measurement interval of one second that begins in the second half of an audio frame, all 5 of the first packets can be transmitted in the second half of an audio frame, and all 5 of the last five packets can be transmitted in the first half of the last audio frame. These results in a Multiplex rate of 205 packets per second that is 2,5 % higher than the User rate of 200 packets per second. This error difference can increase with smaller measurement intervals since for a 100 ms interval the number of packets for the User rate would be 20 while the Multiplex rate could be 25 resulting in a 25 % difference. J.1.3 User rate applications The rate measurements for transport streams are computed for a variety of purposes. These include but are not limited to: - Verification/conformance/troubleshooting - the overall transport stream rate or rates of individual elements are expected to be certain values as set by a user or compression/multiplex system. The user needs to validate that the rates in the stream meet the "expected" rates. This validation can be done over time or just once and can include statistics (e.g. minimum and maximum) as well as history of any rate calculation values. The validation would include all elements including video, audio, conditional access data, PSI/SI tables, etc. - Video and audio quality - there is a strong correlation between video and audio quality and the rate at which these items are transmitted in the transport stream. There is especially a need to monitor the rate of the video since this rate often varies over time and if an video quality issue is determined by visual inspection, there would be a need to determine the rate of the video at that time. A service provider may also guarantee a minimum bit rate for video and audio for a particular program and with a contract, this provider will need to prove that those rates have been met. - Sale of bandwidth - there is a need to monitor the rate of individual elements in a stream over a longer period so that a service provider can charge a user for the bandwidth that has been used in one hour or one day or one week, etc. - Monitoring - there is a need to generate an alarm if the rate of a particular element or the whole stream goes outside some user-specified minimum and maximum range. This error could mean that an element is no longer being included in the transport stream due to a multiplexer malfunction. The accuracy of these rate measurements is not critical to the overall application. J.2 Principles of Bit rate measurement J.2.0 General This is a difficult subject as a measured bitrate depends on the time over which the bitrate is averaged. Bit rate is usually expressed in terms of bits per second, but the actual value that is measured will depend on the way the bits are counted. A bitrate measurement will depend on where in the system the bitrate is measured. For example, in a system, slightly different bitrates may be seen depending on whether the bitrate is measured before or after a large buffer. J.2.1 Gate or Window function On the assumption that Transport Stream packet based systems are being dealt with in the DVB world, there are 3 main choices when counting bytes: - packet based - count only the synchronization bytes; - byte based - count every byte when it arrives; - bit based - count every bit as it arrives. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 164 There are also 2 options for applying the window function: - "continuously" rolling window; - a jumping window (the end of each window is the start of the next window). A jumping window is very undesirable as the bitrate measured will vary depending on when the window is first applied. This rules it out very early. A rolling window is therefore more desirable, but some caution is needed in the use of the term "continuous". The most precise bitrates would be given with a bit based counting scheme. Here, each time a new bit is received, or sent, the total number of bits in the last time window (e.g. 1 s) could be counted and a value displayed. This would always give the most accurate value, but there are a number of serious technical difficulties in implementing this, particularly in offline and semi-offline systems. These difficulties include processing bandwidth and timing accuracy. A byte based system also requires large bandwidth, but both bit and byte based may be required in some special circumstances. Although the present document does not to define byte or bit based profiles, they can easily be added by counting the bytes or bits and adjusting the nomenclature appropriately. A packet based approach may be favourable in situations where cheap implementations with reasonable accuracy are required. It is likely that most DVB Tx and Rx systems would have the capability of deriving some timing information on a packet basis. J.2.2 "Continuous window" If all transport streams were of a constant bitrate, not bursty, continuously clocked and could be easily analysed as a signal with fixed and uniform temporal sampling, then bitrate measurement would be easy. In real systems (bursty ASI, Transport streams over IP, 1394b hubs, cascaded networks, etc.) the bytes and packets do not necessarily arrive on a uniform sampling grid and pragmatic measures need to be taken in defining the window function. To simplify implementation, systems have been considered where the window function is moved across the data in different ways: by byte, by packet, by fixed time interval. There are several points to note about the algorithm in the present document: 1) Strictly speaking, this measure is not continuous. 2) It is a discrete measure whose bitrate values are only valid on time slice boundaries. 3) It is easy to implement and gives a new TS bitrate value every τ (11,1 μs to 1 s). 4) It is applicable to partial transport streams where only a subset of PIDs are being inspected. 5) It can be extended to measure the bitrate of the payload of TS packets. 6) It is repeatable between equipment vendors because the time slice can be made sufficiently small to ensure aliasing is not a problem e.g. when τ = 1 / 90 kHz J.2.3 Time Gate values: 20 ms: gives the peak bitrate of a stream based on variable bitrate elements within it. 1 s: gives a longer term "smooth" average. user: could be used for elements such as subtitles which may only be present from time to time and may require windows of 1 minute or more. J.2.4 Rate measurements in a transport stream Only the Multiplex rates are available to be measured in the transport stream and not the original User rates. In general, it is the User rates that are of interest as outputs of a measurement device with some exception regarding issues of burstiness and buffer models. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 165 Depending on the customer application, the parameters that should be used in the MG bitrate equation in clause 5.3.3. will be different if the user wants to measure User rates or Multiplex rates as finding the best accuracy for the User rates is different than finding the best accuracy for the Multiplex rates. The parameters also need to take into account tracking the changes in the rate versus time. The parameters should in general be different for elements that differ either in type or in rate in order to maintain accuracy. Here are some general considerations for the parameters: - For elements that have CBR, increasing T will push the measured Multiplex rate towards the User rate. - For reasonable accuracy of the User rate, T should be large enough to include multiple elements of what is being measured. For example, if the rate of a SDT is being measured, it should include at least 10 different arrivals of the SDT. Decreasing τ will cause the Multiplex rate to be more accurately tracked but will not increase the accuracy of calculating User rates for CBR streams. For VBR streams, a smaller τ to within some limits will allow the changes to be better averaged over time. J.3 Use of the MG profiles J.3.0 General The profiles in clause 5.3.3.2 have been designed to have the properties described below. J.3.1 MGB1 Profile - the backwards compatible profile This is a backwards compatible profile where a 1 second jumping window is used to measure bitrate. In a rigidly CBR system, this will give a good indication of the bitrate, but will give aliasing and inaccuracy if the bitrate being measured is changing faster than every 1s. This makes it impractical for looking at VBR systems, or for looking at the bitrates of VBR components (e.g. stat-mux video) in a CBR transport stream. This profile is included for backwards compatibility with existing equipment. J.3.2 MGB2 Profile - the Basic bitrate profile This profile is recommended for new designs. It is intended to give a good idea of the average bitrate of a system, yet have enough resolution (due to a small τ value) to show whether the bitrate is truly static or is varying with time. The values have been chosen to allow simple implementation. J.3.3 MGB3 Profile - the precise Peak bitrate profile This profile has a time gate which is small enough to show the variable bitrate characteristics of a statistical multiplex environment. The timeSlice is small enough to ensure that only a single packet header will occur in each timeSlice for most distribution systems. The time gate is short enough so that frame by frame averaging does not take place. The timebase chosen can be locked to, or derived from the PCR in a decoder or encoder environment for ease of implementation. J.3.4 MGB4 Profile - the precise profile This profile is intended to give a "true" smoothed bitrate. The timeSlice is small enough to ensure that only a single packet header will occur in each timeSlice for most distribution systems. The time gate is a little over 1 second to give a long time constant averaging to the data. The timebase chosen can be locked to, or derived from the PCR in a decoder or encoder environment for ease of implementation. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 166 J.3.5 MGB5 Profile - the user profile This profile is intended to give extensibility to the bitrate measurement algorithm. It allows different time gates and timeSlice values to be defined. These can be applied to the whole transport stream, or to individual components of the stream. It is important when using this profile that the results are carefully documented using the nomenclature in these guidelines. This will ensure that results can be repeated at a later date. J.4 Error values in the measurements J.4.0 General It is worth noting the areas where errors can be introduced into the measurement: • clock instability in the time gate and time slice functions; • quantization due to counting elements which are too big e.g. too many or too few packet headers may fall within the time gate; • aliasing due to having a timeSlice or a time Gate which is too large for the parameter being measured. In real systems, the errors due to clock instability and quantization tend to be rather small. The biggest problem is inappropriate use of timeSlice and time gate values. This can be best demonstrated by an example. Imagine a DVB-S statistical multiplex system (38.1 Mbit/s) where a particular video PID has a bitrate limit of 3 -5 Mbit/s and the hypothetical video encoder is able to change its bitrate every 80ms. Bit rate is measured by counting packet headers of a certain PID. The average video rate is 4 Mbit/s. If the MGB4 profile is used, DVB-S ≈ 38,1 Mbit/s packet duration ≈ 40 μs packets per τ ≈ 0,25 The clock frequency error uncertainty may be as high as 500 ppm. This would lead to an error in the duration of the time gate of 500 ppm (0,05 %). This could increase the 1 second window by 500 μs which at 5 Mbit/s could allow an extra 2 packets into the gate. This would give an error of: = 2 × 188 × 8 bits/s = 0,06 % of 5 Mbit/s The uncertainty due to quantization is equal to the element size which is counted which is 1 packet per time gate in this case: = 188 × 8 bit/s = 1 504 bit/s = 0,03 % of 5 Mbit/s It can be seen that these values are all quite small. In case of the slightly contrived example of a sequence which requires the bitrate shown below: Difficult 5 Mbit/s 1 s Easy 3 Mbit/s 1 s Difficult 5 Mbit/s 1 s Easy 3 Mbit/s 1 s Difficult 5 Mbit/s 1 s Easy 3 Mbit/s 1 s • The MGB4 profile will show a smoothed version of the above bitrate with peak values of 5 Mbit/s and 3 Mbit/s. • The MGB3 profile will show much sharper edges to the bitrate changes and will report the peak values of 5 Mbit/s and 3 Mbit/s. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 167 • The MGB1 profile, however will show different values depending on the moment when the 1 second window jumps to its next starting point. If it is synchronized with the start of the 1 second sequences, then it will report the correct values of 5 Mbit/s and 3 Mbit/s. If, however it starts its measurements 50 % of the way through a 1 second sequence, it will report that the bitrate is constant at 4 Mbit/s. This is an error of 33 % at 3 Mbit/s or 20 % at 5 Mbit/s. Real errors are less than in this contrived example, but this source of error is the most significant in real systems. Note that in some monitoring applications errors of a few percent may be tolerable, whereas in other applications a precision of 1ppm or better may be required. J.4.1 Very Precise measurements In very accurate measurements, it may be necessary to count individual bytes, or individual bits to obtain the required precision. The same algorithm, nomenclature and synchronization as described in clause 5.3.3 may still be used and the results will be repeatable. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 168 Annex K: DVB-T channel characteristics K.0 Introduction This annex provides some information on terrestrial channel profiles which can be used for off-line computer simulations and realtime simulations based on dedicated equipment. The properties of these profiles reflect realistic reception conditions and/or worst-case scenarios and were used to verify specific features of the DVB-T standard. K.1 Theoretical channel profiles for simulations without Doppler shift (quoted from ETSI EN 300 744 [i.9]) The performance of the DVB-T system has been simulated during the development of the standard ETSI EN 300 744 [i.9] with two channel models for fixed reception - F1 and portable reception - P1, respectively. The channel models have been generated from the following equations where x(t) and y(t) are input and output signals respectively: a) Fixed reception F1: = = ⋅ ⋅ − − ⋅ ⋅ + ⋅ = N i i N i i j i t x e t x t y i 0 2 1 2 0 ) ( ) ( ) ( ρ τ ρ ρ θ π where: - the first term before the sum represents the line of sight ray; - N is the number of echoes equals to 20; - θi is the phase shift from scattering of the i'th path - listed in table K.1; - ρi is the attenuation of the i'th path - listed in table K.1; - τi is the relative delay of the i'th path - listed in table K.1. The Ricean factor K (the ratio of the power of the direct path (the line of sight ray) to the reflected paths) is given as: = = N 1 i 2 i 2 0 r r K In the simulations a Ricean factor K = 10 dB has been used. In this case: = ⋅ = N i i o 1 2 10 ρ ρ ETSI ETSI TR 101 290 V1.4.1 (2020-06) 169 b) Portable reception, Rayleigh fading (P1): = ⋅ ⋅ − − ⋅ ⋅ ⋅ = N i i j i t x e k t y i 1 2 ) ( ) ( τ ρ θ π where = = N i i k 1 2 1 ρ θi, ρi and τi are given in table K.1. Table K.1: Attenuation, phase and delay values for F1 and P1 i ρi τi [μs] θi [rad] 1 0,057 662 1,003 019 4,855 121 2 0,176 809 5,422 091 3,419 109 3 0,407 163 0,518 650 5,864 470 4 0,303 585 2,751 772 2,215 894 5 0,258 782 0,602 895 3,758 058 6 0,061 831 1,016 585 5,430 202 7 0,150 340 0,143 556 3,952 093 8 0,051 534 0,153 832 1,093 586 9 0,185 074 3,324 866 5,775 198 10 0,400 967 1,935 570 0,154 459 11 0,295 723 0,429 948 5,928 383 12 0,350 825 3,228 872 3,053 023 13 0,262 909 0,848 831 0,628 578 14 0,225 894 0,073 883 2,128 544 15 0,170 996 0,203 952 1,099 463 16 0,149 723 0,194 207 3,462 951 17 0,240 140 0,924 450 3,664 773 18 0,116 587 1,381 320 2,833 799 19 0,221 155 0,640 512 3,334 290 20 0,259 730 1,368 671 0,393 889 NOTE: Figures in italics are approximate values. NOTE: For practical implementations profiles with reduced complexity have been used successfully. In many cases it seems sufficient to use e. g. only the six paths with the highest amplitude. K.2 Profiles for realtime simulations without Doppler shift The following profiles were used in laboratory tests in a research project with satisfactory results. NOTE: AC106 Validate (1995-1998). Table K.2: Echo Profiles Path fixed delay [µs] C/I [dB] Portable delay [µs] C/I [dB] dense SFN delay [µs] C/I [dB] #1 (main) 0 0 - - 0 0 #2 0,5 17,8 0,5 7,8 7,8 9,3 #3 1,95 17,9 1,95 7,9 11,6 5,5 #4 3,25 19,1 3,25 9,1 17,5 16,1 #5 2,75 20,4 2,75 10,4 20,0 14,5 #6 0,45 20,6 0,45 10,6 23,4 23,4 #7 - - 0,85 11,6 - - ETSI ETSI TR 101 290 V1.4.1 (2020-06) 170 K.3 Profiles for realtime simulation with Doppler shift (mobile channel simulation) In the course of a research project (see note), three channel profiles were selected to reproduce the DVB-T service delivery situation in a mobile environment. Two of them reproduce the characteristics of the terrestrial channel propagation with a single transmitter, the third one reproduces the situation coming from an SFN operation of the DVB-T network. NOTE: AC318 Motivate (1998-2000). The following tables describe the composition of the chosen profiles. • Typical Urban reception (TU6) This profile reproduces the terrestrial propagation in an urban area. It was originally defined by COST 207 [i.44] as a Typical Urban (TU6) profile and is made of 6 paths having wide dispersion in delay and relatively strong power. This channel profile has also been used for GSM and DAB tests. Table K.3: Echo profile for Typical Urban reception (TU6) Tap number Delay (us) Power (dB) Doppler spectrum 1 0,0 -3 Rayleigh 2 0,2 0 Rayleigh 3 0,5 -2 Rayleigh 4 1,6 -6 Rayleigh 5 2,3 -8 Rayleigh 6 5,0 -10 Rayleigh • Typical Rural Area reception (RA6) This profile reproduces the terrestrial propagation in an rural area. It has been defined by COST 207 [i.44] as a Typical Rural Area (RA6) profile and is made of 6 paths having relatively short delay and small power. This channel profile has been used for GSM and DAB tests. Table K.4: Echo profile for Typical Rural Area reception (RA6) Tap number Delay (us) Power (dB) Doppler spectrum 1 0,0 0 Rice 2 0,1 -4 Rayleigh 3 0,2 -8 Rayleigh 4 0,3 -12 Rayleigh 5 0,4 -16 Rayleigh 6 0,5 -20 Rayleigh • 0 dB Echo profile This profile has been defined by Motivate partners. Its composition has been largely influenced by the specific nature of the DVB-T signal, especially its spread spectrum technique (introducing an Inter Carrier Interference sensitivity to Doppler spread) and its use of a Guard Interval (introducing an Inter Symbol sensitivity to the echoes delays). Moreover, its definition has been driven by the analysis of the profiles encountered during the various field trials performed during the Motivate project. This profile is made of two rays having the same power, delayed by half the Guard Interval value and presenting a pure Doppler characteristic. Table K.5: Profile for 0 dB echo reception Tap number Delay (us) Power (dB) Doppler spectrum Frequency ratio 1 0 0 Pure Doppler -1 2 1/2 Tg 0 Pure Doppler +1 ETSI ETSI TR 101 290 V1.4.1 (2020-06) 171 Annex L: The measurement of MER under ACE When ACE method means Active Constellation Extension, it uses outer point constellation points of the useful carriers to reduce the Peak to Average value of the DVB-T2 signals. It is an option of the DVB-T2 standard and it is used for example in the L1 signalling parameters since ETSI EN 302 755 (V1.3.1) [i.27]. When ACE is applied on a COFDM signal, the professional receiver is not capable of referring to the theoretical reference point in order to measure the modulation error distance as in the formula. The problem is to find a solution that allows MER measurement on carriers when Active Constellation Extension is used on a DVB-T2 signal: 1) Enable Active constellation point. 2) Disable ACE on some carriers based on a circular mask pattern. The number of useful carriers is less than 1% in order not to impact the Peak to Average Ratio of the final signal. 3) This method is a test mode located in the modulation part implementing ACE algorithm. The receiver can recover the MER based on the mask pattern definition. The ACE method is defined in clause 9.6.1 of ETSI EN 302 755 [i.27]. Figure L.1: Implementation of the ACE method (extracted from ETSI EN 302 755 [i.27]) In the figure L.1, "extendable" signification is modified from the DVB-T2 specification ETSI EN 302 755 [i.27] clause 9.6.1, in order to add a circular mask pattern exclusion. Standard definition of "extendable": A component is defined as extendable if it is an active cell (i.e. an OFDM cell carrying a constellation point for L1 signalling or a PLP), and if its absolute amplitude is greater than or equal to the maximal component value associated to the modulation constellation used for that cell; a component is also defined as extendable if it is a dummy cell, a bias balancing cell or an unmodulated cell in the Frame Closing Symbol. ETSI ETSI TR 101 290 V1.4.1 (2020-06) 172 For the test mode, the text is adjusted: A component is defined as extendable if it is an active cell (i.e. an OFDM cell carrying a constellation point for L1 signalling or a PLP), and if its absolute amplitude is greater than or equal to the maximal component value associated to the modulation constellation used for that cell; a component is also defined as extendable if it is a dummy cell, a bias balancing cell or an unmodulated cell in the Frame Closing Symbol. In test mode, active cells corresponding to the mask pattern index position list are not extended. The circular mask pattern is changed for each new T2 frame symbol in order to measure MER for all the carriers. The mask pattern M[] for ACE is defined as follows: Idx=Frame Idx mod 100; i=0; While (idx<Cuseful) { Idx=mod(Idx+div(Cuseful /100), Cuseful) M[i]=idx; i++; } For a given frame index, where Idx provides the carrier position and M[] is the mask pattern containing index position is ascendant form defined for a Frame number FrameIdx. Where Cuseful Cuseful = Cdata for data symbols Cuseful= CP2 for P2 symbols Cuseful = CFC for Frame closing symbols ETSI ETSI TR 101 290 V1.4.1 (2020-06) 173 Annex M: Bibliography • ETSI TS 102 154: "Digital Video Broadcasting (DVB); Implementation guidelines for the use of Video and Audio Coding in Contribution and Primary Distribution Applications based on the MPEG-2 Transport Stream". • Recommendation ITU-T G.826: "Error performance parameters and objectives for international, constant bit rate digital paths at or above the primary rate". • ETSI TR 101 290 (V1.2.1) 2001-05: "Digital Video Broadcasting (DVB); Measurement guidelines for DVB systems". ETSI ETSI TR 101 290 V1.4.1 (2020-06) 174 History Document history Edition 1 May 1997 Publication as ETSI ETR 290 V1.2.1 May 2001 Publication V1.3.1 July 2014 Publication V1.4.1 June 2020 Publication |
17276d18a3f94eabc0ca9dab26c56b19 | 101 310 | 1 Scope | The present document describes the traffic capacity and the spectrum requirements for multi-system and multi-service Digital Enhanced Cordless Telecommunications (DECT) applications coexisting on a common frequency band. Configurations for typical DECT applications, and relevant mixes of these, including residential, office, public and Radio in the Local Loop (RLL) applications, are defined and the traffic capacity is analysed, mainly by advanced simulations. These results are used together with relevant deployment scenarios to estimate spectrum requirements for reliable services, specifically for a public multi-operator licensing regime. Recommendations are given on conflict solving rules that conserve the high spectrum efficiency gain of shared spectrum while maintaining control of the service quality in one's own system. These recommendations cover synchronization, directional gain antennas, traffic limits per DECT Radio Fixed Part (RFP), use of Wireless Relay Stations (WRSs), different rules for private and public operators and procedures needed for timely local adjustments where and when the local traffic increases. Results of studies on compatibility with other relevant radio technologies using spectrum adjacent to the DECT band, are also included. |
17276d18a3f94eabc0ca9dab26c56b19 | 101 310 | 2 References | For the purposes of this Technical Report (TR) the following references apply:. [1] ETSI EN 300 175-1: "Digital Enhanced Cordless Telecommunications (DECT); Common Interface (CI); Part 1: Overview". [2] ETSI EN 300 175-2: "Digital Enhanced Cordless Telecommunications (DECT); Common Interface (CI); Part 2: Physical Layer (PHL)". [3] ETSI EN 300 175-3: "Digital Enhanced Cordless Telecommunications (DECT); Common Interface (CI); Part 3: Medium Access Control (MAC) layer". [4] ETSI EN 300 175-4: "Digital Enhanced Cordless Telecommunications (DECT); Common Interface (CI); Part 4: Data Link Control (DLC) layer". [5] ETSI EN 300 175-5: "Digital Enhanced Cordless Telecommunications (DECT); Common Interface (CI); Part 5: Network (NWK) layer". [6] ETSI EN 300 175-6: "Digital Enhanced Cordless Telecommunications (DECT); Common Interface (CI); Part 6: Identities and addressing". [7] ETSI EN 300 175-7: "Digital Enhanced Cordless Telecommunications (DECT); Common Interface (CI); Part 7: Security features". [8] ETSI EN 300 175-8: "Digital Enhanced Cordless Telecommunications (DECT); Common Interface (CI); Part 8: Speech coding and transmission". [9] ETSI EN 300 444: "Digital Enhanced Cordless Telecommunications (DECT); Generic Access Profile (GAP)". [10] ETSI TBR 006: "Digital Enhanced Cordless Telecommunications (DECT); General terminal attachment requirements". [11] ETSI EN 300 765-1: "Digital Enhanced Cordless Telecommunications (DECT); Radio in the Local Loop (RLL) Access Profile (RAP); Part 1: Basic telephony services". [12] ETSI EN 300 765-2: "Digital Enhanced Cordless Telecommunications (DECT); Radio in the Local Loop (RLL) Access Profile (RAP); Part 2: Advanced telephony services". [13] ETSI TR 101 178: "Digital Enhanced Cordless Telecommunications (DECT); A high level guide to the DECT standardization". [14] ETSI ETR 246: "Digital Enhanced Cordless Telecommunications (DECT); Application of DECT Wireless Relay Stations (WRS)". ETSI ETSI TR 101 310 V1.2.1 (2004-04) 8 [15] ETSI EN 300 700: "Digital Enhanced Cordless Telecommunications (DECT); Wireless Relay Station (WRS)". [16] ETSI ETR 308: " Digital Enhanced Cordless Telecommunications (DECT); Services, facilities and configurations for DECT in the local loop". [17] Akerberg, Brouwer, van de Berg, Jager: "DECT technology for radio in the local loop", Proceedings of the IEEE 44th Vehicular Technology Conference, (Stockholm June 8-10 1994). [18] TIA/T1 JTC(AIR)/95.02.02-012R1: "TAG 3 (PACS) Radio Channel System Report". [19] TIA/EIA-662: "Personal Wireless Telecommunication Standard (PWT)". [20] TIA/EIA-696: "Personal Wireless Telecommunications - Enhanced Interoperability Standard". [21] ETSI ETR 042: " Digital Enhanced Cordless Telecommunications (DECT); A Guide to DECT features that influence the traffic capacity and the maintenance of high radio link transmission quality, including the results of simulations". NOTE: ETR 042 has been given historical status, which means that this report will not be maintained by ETSI any longer. The historical status indicates that parts of this report may no longer be considered valid. [22] ETSI ETR 139: "Radio Equipment and Systems (RES); Radio in the Local Loop (RLL)". [23] Directive 1999/5/EC of the European Parliament and of the Council of 9 March 1999 on radio equipment and telecommunications terminal equipment and the mutual recognition of their conformity (R&TTE Directive). [24] Council Directive 91/287/EEC of 3 June 1991 on the frequency band to be designated for the coordinated introduction of digital European cordless telecommunications (DECT) into the Community. [25] Council Recommendation 91/288/EEC of 3 June 1991 on the coordinated introduction of digital European cordless telecommunications (DECT) into the Community. [26] ETSI EN 301 406: "Digital Enhanced Cordless Telecommunications (DECT); Harmonized EN for Digital Enhanced Cordless Telecommunications (DECT) covering essential requirements under article 3.2 of the R&TTE Directive; Generic radio". [27] ETSI EN 301 908-10: "Electromagnetic compatibility and Radio spectrum Matters (ERM); Base Stations (BS), Repeaters and User Equipment (UE) for IMT-2000 Third-Generation cellular networks; Part 10: Harmonized EN for IMT-2000, FDMA/TDMA (DECT) covering essential requirements of article 3.2 of the R&TTE Directive". [28] CEPT ERO Report 065: "Adjacent band compatibility between UMTS and other services in the 2 GHz band", November 1999. [29] CEPT ERC Report 100: "Compatibility between certain radiocommunications systems operating in adjacent bands; Evaluation of DECT/GSM 1800 compatibility", February 2000. [30] CITEL: "Guide on results of the CITEL study to quantify issues of incompatibility between FWA and PCS on the 1850-1990 MHz band", OEA/Ser.L/XVII.6.1, February 2000. [31] ETSI TR 101 370: "Digital Enhanced Cordless Telecommunications (DECT); Implementing DECT Fixed Wireless Access (FWA) in an arbitrary spectrum allocation". [32] FCC 02-151: "Federal Communications Commission, Second Report and Order, Amendment of Part 15 of the Commission's Rules Regarding Spread Spectrum Devices". [33] ERC/DEC/(94)03 ERC Decision of 24th October 1994 on the frequency band to be designated for the coordinated introduction of the Digital European Cordless Telecommunications system [34] European Commission Decision of 6 April 2000 establishing the initial classification of radio equipment and Telecommunications terminal equipment. ETSI ETSI TR 101 310 V1.2.1 (2004-04) 9 [35] ERC Decision of 23 November 1998 on Exemption from Individual Licensing of DECT equipment, except fixed parts which provide for public access and associated identifiers. [36] CEPT/ERC Recommendation 21-13 E: "Licensing regime for digital European Cordless Telecommunications (DECT) Equipment", (Brussels 1994). |
17276d18a3f94eabc0ca9dab26c56b19 | 101 310 | 3 Definitions and abbreviations | |
17276d18a3f94eabc0ca9dab26c56b19 | 101 310 | 3.1 Definitions | For the purposes of the present document, the following terms and definitions apply: antenna diversity: implies that the RFP for each bearer independently can select different antenna properties such as gain, polarization, coverage patterns, and other features that may affect the practical coverage NOTE: A typical example is space diversity, provided by two vertically polarized antennas separated by 10 cm to 20 cm. broadcast: simplex point-to-multipoint mode of transmission NOTE: The transmitter may disregard the presence or absence of receivers. call: all of the Network (NWK) layer processes involved in one NWK layer peer-to-peer association NOTE: Call may sometimes be used to refer to processes of all layers, since lower layer processes are implicitly required. cell: domain served by a single antenna(e) system (including a leaky feeder) of one Fixed Part (FP) NOTE: A cell may include more than one source of radiated Radio Frequency (RF) energy (i.e. more than one radio end point). centrex: implementation of a private telecommunication network exchange that is not located on the premises of the private network operator NOTE: It may be co-located with, or physically a part of a public exchange. cluster: logical grouping of one or more cells between which bearer handover is possible NOTE 1: A Cluster Control Function (CCF) controls one cluster. NOTE 2: Internal handover to a cell which is not part of the same cluster can only be done by connection handover. Cordless Radio Fixed Part (CRFP): WRS that provides independent bearer control to a PT and FT for relayed connections coverage area: area over which reliable communication can be established and maintained double-simplex bearer: use of two simplex bearers operating in the same direction on two physical channels NOTE 1: These pairs of channels always use the same RF carrier and always use evenly spaced slots (i.e. separated by 0,5 Time Division Multiple Access (TDMA) frame). NOTE 2: A double-simplex bearer only exists as part of a multibearer MAC connection. down-link: transmission in the direction FT to PT duplex bearer: use of two simplex bearers operating in opposite directions on two physical channels NOTE: These pairs of channels always use the same RF carrier and always use evenly spaced slots (i.e. separated by 0,5 TDMA frame). ETSI ETSI TR 101 310 V1.2.1 (2004-04) 10 End System (ES): logical grouping that contains application processes and supports telecommunication services NOTE: From the OSI point of view, end systems are considered as sources and sinks of information. external handover: process of switching a call in progress from one FP to another FP Fixed Part (DECT Fixed Part) (FP): physical grouping that contains all of the elements in the DECT network between the local network and the DECT air interface NOTE: A DECT FP contains the logical elements of at least one FT, plus additional implementation specific elements. Fixed radio Termination (FT): logical group of functions that contains all of the DECT processes and procedures on the fixed side of the DECT air interface NOTE: A FT only includes elements that are defined in EN 300 175, parts 1 [1] to 8 [8]. This includes radio transmission elements (layer 1) together with a selection of layer 2 and layer 3 elements. full slot (slot): one 24th of a TDMA frame which is used to support one physical channel guard space: nominal interval between the end of a radio transmission in a given slot, and the start of a radio transmission in the next successive slot NOTE: This interval is included at the end of every slot, in order to prevent adjacent transmissions from overlapping even when they originate with slightly different timing references (e.g. from different radio end points). half slot: one 48th of a TDMA frame which is used to support one physical channel handover: process of switching a call in progress from one physical channel to another physical channel NOTE 1: These processes can be internal (see internal handover) or external (see external handover). NOTE 2: There are two physical forms of handover, intracell handover and inter-cell handover. Intracell handover is always internal, inter-cell handover can be internal or external. incoming call: call received at a Portable Part (PP) inter-cell handover: switching of a call in progress from one cell to another cell internal handover: handover processes that are completely internal to one FT. NOTE 1: Internal handover reconnects the call at the lower layers, while maintaining the call at the NWK layer. NOTE 2: The lower layer reconnection can either be at the DLC layer (see connection handover) or at the MAC layer (see bearer handover). interoperability: capability of FPs and PPs, that enable a PP to obtain access to teleservices in more than one location area and/or from more than one operator (more than one service provider) InterWorking Unit (IWU): unit that is used to interconnect subnetworks NOTE: The IWU will contain the InterWorking Functions (IWF) necessary to support the required subnetwork interworking. intracell handover: switching of a call in progress from one physical channel of one cell to another physical channel of the same cell multiframe: repeating sequence of 16 successive TDMA frames, that allows low rate or sporadic information to be multiplexed (e.g. basic system information or paging) network (telecommunication network): the means of providing telecommunication services between a number of locations where the services are accessed via equipment attached to the network operator (DECT operator): individual or entity who or which is responsible for operation of one or more DECT FPs NOTE: The term does not imply any legal or regulatory conditions, nor does it imply any aspects of ownership. ETSI ETSI TR 101 310 V1.2.1 (2004-04) 11 outgoing call: call originating from a PP paging: process of broadcasting a message from a DECT FP to one or more DECT PPs NOTE: Different types of paging message are possible. For example, the {Request paging} message orders the recipient to respond with a call set-up attempt. paging area: domain in which the PP will be paged as a part of incoming call establishment NOTE: In general, the paging area will be equal to the Temporary Portable User Identity (TPUI) domain, since the TPUI is used for paging. Portable Part (DECT Portable Part) (PP): physical grouping that contains all elements between the user and the DECT air interface. NOTE 1: PP is a generic term that may describe one or several physical pieces. NOTE 2: A DECT PP is logically divided into one PT plus one or more portable applications. Portable radio Termination (PT): logical group of functions that contains all of the DECT processes and procedures on the portable side of the DECT air interface NOTE: A PT only includes elements that are defined in EN 300 175, parts 1 [1] to 8 [8]. This includes radio transmission elements (layer 1) together with a selection of layer 2 and layer 3 elements. private: attribute indicating that the application of the so qualified term, e.g. a network, an equipment, a service, is offered to, or is in the interest of, a determined set of users NOTE: The term does not include any legal or regulatory aspects, nor does it indicate any aspects of ownership. public: attribute indicating that the application of the so qualified term, e.g. a network, an equipment, a service, is offered to, or is in the interest of, the general public NOTE: The term does not include any legal or regulatory aspects, nor does it indicate any aspects of ownership. public access service: service that provides access to a public network for the general public NOTE: The term does not imply any legal or regulatory aspect, nor does it imply any aspects of ownership. radio channel: See RF channel. Radio Fixed Part (RFP): one physical sub-group of a FP that contains all the REpeater Parts (REPs) (one or more) that are connected to a single system of antennas REpeater Part (REP): WRS that relays information within the half frame time interval RF carrier (carrier): centre frequency occupied by one DECT transmission RF channel: nominal range of frequencies (RF spectrum) allocated to the DECT transmissions of a single RF carrier service provider (telecommunications service provider): individual, or entity, who, or which, interfaces to the customer in providing telecommunications service NOTE 1: The term does not imply any legal or regulatory conditions, nor does it indicate whether public service or private service is provided. NOTE 2: The term service provider is also used with a different meaning in the ISO/OSI layered model. simplex bearer: simplex bearer is the MAC layer service that is created using one physical channel NOTE: See also duplex bearer and double simplex bearer. subscriber (customer): natural person, or the juristic person who has subscribed to telecommunication services, and is, therefore, responsible for payment TDMA frame: time-division multiplex of 10 ms duration containing 24 successive full slots NOTE: A TDMA frame starts with the first bit period of full slot 0 and ends with the last bit period of full slot 23. ETSI ETSI TR 101 310 V1.2.1 (2004-04) 12 telecommunication: any transmission and/or emission and/or reception of signals representing signs, writing, images, and sounds or intelligence of any nature by wire, radio, optical or other electromagnetic systems teleservice: type of telecommunication service that provides the complete capability, including terminal equipment functions, for communication between users, according to protocols that are established by agreement up link: transmission in the direction PT to FT user (of a telecommunication network): person or machine delegated by a subscriber (by a customer) to use the services, and/or facilities, of a telecommunication network Wireless Relay Station (WRS): physical grouping that combines elements of both PTs and FTs to relay information on a physical channel from one DECT termination to a physical channel to another DECT termination NOTE: The DECT termination can be a PT or an FT or another WRS. |
17276d18a3f94eabc0ca9dab26c56b19 | 101 310 | 3.2 Abbreviations | For the purposes of this the present document, the following abbreviations apply: C/I Carrier to Interference ratio CDCS Continuous Dynamic Channel Selection CRFP Cordless Radio Fixed Part CTA Cordless Terminal Adaptor DAS DECT Access Site DCS Dynamic Channel Selection DECT Digital Enhanced Cordless Telecommunications E Erlangs ES End System FCA Fixed Channel Allocation FDD Frequency Division Duplex FDMA Frequency Division Multiple Access FP Fixed Part GAP Generic Access Profile GFSK Gaussian Frequency Shift Keying GoS Grade of Service GPS Global Positioning System IS Intermediate System LOS Line Of Sight NLOS Near Line Of Sight O&M Operations and Maintenance PABX Private Automatic Branch Exchange PCS Personal Communications Systems PP Portable Part REP REpeater Part RFP Radio Fixed Part RLL Radio in the Local Loop TDD Time Division Duplex TDMA Time Division Multiple Access UMTS Universal Mobile Telephone System WPBX Wireless PABX WRS Wireless Relay Station ETSI ETSI TR 101 310 V1.2.1 (2004-04) 13 |
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